WO2016006066A1 - Contactless power supply device - Google Patents

Abstract

This contactless power supply device is equipped with: a power transmission coil for transmitting power in a contactless manner to a power-receiving coil which is electrically connected to a load; an inverter for converting power from a power source into AC power, and outputting the AC power to the power transmission coil; and a control means for setting a plurality of inverter drive frequencies, and driving the inverter using a varying control for varying the frequency when driving the inverter. The inverter is connected between the power transmission coil and the power source. The drive frequencies are a plurality of discrete frequencies. In addition, during the frequency-varying control, the control means varies the frequency when driving the inverter by repeating the plurality of drive frequencies.

Description

Non-contact power feeding device

The present invention relates to a non-contact power feeding device.

In a non-contact power supply device that supplies high-frequency power to a load in a non-contact manner, the high-frequency power generated by the high-frequency power supply unit is linked to the high-frequency magnetic flux converted by the primary conduction and the high-frequency magnetic flux converted by the primary conduction. A receiving coil that generates an induced electromotive force and supplies the induced electromotive force to a load, a resistance detection circuit that detects a resistance component of the load, and a frequency control unit that controls the frequency of the AC wave power. By calculating the equivalent resistance component of the load and controlling the drive frequency of the inverter so that the impedance seen from the load side is almost a pure resistance component, the power factor of the power supplied to the load side is controlled to be close to 1. Is disclosed (for example, Patent Document 1).

JP 2002-272134 A

However, since the above non-contact power feeding device drives the inverter at a single frequency, the ampere-turn composite value is determined to be a unique value corresponding to the single frequency. When the ampere turn composite value is determined at a high value, there is a problem that the ampere turn composite value cannot be controlled and the leakage magnetic field becomes large.

The problem to be solved by the present invention is to provide a non-contact power feeding device that can control the ampere turn composite value and reduce the leakage magnetic field.

In the present invention, the drive frequency of the inverter is set as a plurality of discrete frequencies. Then, during frequency variable control, the frequency at the time of driving the inverter is repeatedly varied at the plurality of driving frequencies to drive the inverter, and the power is supplied from the power transmission coil to the power reception coil in a non-contact manner, thereby solving the above problem. To do.

According to the present invention, since the energy is dispersed by driving the inverter while changing the drive frequency, the ampere-turn composite value is lowered, and the leakage magnetic flux can be suppressed.

It is a block diagram of the non-contact electric power feeding system concerning the embodiment of the present invention. It is a block diagram of the inverter controller of FIG. In the non-contact electric power feeding system which concerns on this embodiment, it is a graph which shows the frequency characteristic of an output current, an ampere turn synthetic | combination value, and a coil current. In the non-contact electric power feeding system which concerns on a comparative example, the circuit diagram of a power transmission coil and a receiving coil, the vector diagram of an ampere turn synthetic value, and the graph of the electric field strength with respect to an ampere turn synthetic value are shown. In the non-contact electric power feeding system concerning this embodiment, it is a graph which shows the frequency characteristic of an output current, an ampere turn synthetic value, a coil current, and input impedance. In the non-contact power supply system according to the present embodiment is a vector diagram for explaining the ampere-turns synthesis value when the carrier frequency is set to f ₁ and f _2. In the non-contact power supply system according to the present embodiment, the magnitude of the coil current (I ₁ , I ₂ ) when the inverter 18 is driven with a single carrier frequency (f ₁ , f ₂ ) and the ampere-turn composite value are expressed as follows: It is a graph to show. It is a circuit diagram of the inverter of FIG. It is a flowchart which shows a control flow in the non-contact electric power feeding system which concerns on this embodiment. In the non-contact electric power feeding system which concerns on this embodiment, it is a graph which shows the characteristic of the carrier signal at the time of PAM control, and the characteristic of the carrier signal at the time of PFM control. It is a graph which shows the characteristic of a carrier frequency in the non-contact electric power feeding system which concerns on this embodiment. It is the graph which showed the coil loss and the ampere turn synthetic | combination value about the non-contact electric power feeding system which concerns on this embodiment, and a comparative example. It is the graph which showed the EMI level and the EMI level maximum value about the non-contact electric power feeding system which concerns on this embodiment, and a comparative example. It is a flowchart which shows a control flow in the non-contact electric power feeding system which concerns on other embodiment. It is a graph which shows the frequency characteristic of an input impedance, an output current, and an ampere turn synthetic value in the non-contact electric supply system concerning other embodiments. In the non-contact power supply system according to another embodiment, each carrier frequency _{(f 1, f 2, f} 3, f 4), is a table showing the phase, the relationship between the ampere-turns combined value. In the non-contact power supply system according to another embodiment, a graph showing the magnitude of the size and ampere turn combined value of the coil current _{(I 1,} I _2). In the non-contact electric power feeding system which concerns on other embodiment, it is a graph which shows the characteristic of the carrier signal at the time of PAM control, and the characteristic of the carrier signal at the time of PFM control. It is a graph which shows the characteristic of a carrier frequency in the non-contact electric power feeding system which concerns on other embodiment. It is the graph which showed the coil loss and the ampere turn synthetic | combination value about the non-contact electric power feeding system which concerns on other embodiment, and a comparative example. It is the graph which showed the EMI level and the EMI level maximum value about the non-contact electric power feeding system which concerns on other embodiment, and a comparative example. It is a graph which shows the frequency characteristic of an input impedance, an output current, and an ampere turn synthetic value in the non-contact electric supply system concerning other embodiments. In the non-contact power supply system according to another embodiment, each carrier frequency _{(f 1, f 2, f} 3, f 4), is a table showing the phase, the relationship between the ampere-turns combined value. It is a graph which shows the characteristic of a carrier frequency in the non-contact electric power feeding system which concerns on a modification. It is a graph which shows the frequency characteristic of an input impedance, an output current, and an ampere turn synthetic value in the non-contact electric supply system concerning a modification. In the non-contact power supply system according to a modification, and each carrier frequency _{(f 1, f 2, f} 3, f 4), is a table showing the phase, the relationship between the ampere-turns combined value.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

<< First Embodiment >>
FIG. 1 is a block diagram of a non-contact power feeding system according to an embodiment of the present invention. The contactless power supply system of this example is a system for supplying contactless power from the ground side toward the vehicle battery when charging a battery of a vehicle such as an electric vehicle. The non-contact power feeding system according to the present invention is not limited to a vehicle including a battery, and can be applied to other devices.

The non-contact power supply system includes a primary-side non-contact power supply device provided on the ground side and a secondary-side power supply device provided on the vehicle side. The non-contact power feeding device on the primary side includes an AC power source 11, a rectifier circuit 12, a power factor correction circuit (PFC) 15, a capacitor 17, an inverter 18, a primary side resonance circuit 19, voltage sensors 13, 16 and a current sensor. 14 and a ground side controller 100. The secondary-side non-contact power feeding device includes a secondary-side resonance circuit 21, a rectifier circuit 22, an LC filter 23, a voltage sensor 24, a current sensor 25, a battery 26, and a vehicle-side controller 200.

AC power supply 11 is a commercial power supply for outputting AC power of a commercial frequency (for example, 50 Hz or 60 Hz). The AC power supply has a pair of power lines. The rectifier circuit 12 is a circuit that rectifies AC power output from the AC power supply 11 into DC. The rectifier circuit 12 is connected between the AC power supply 11 and the power factor correction circuit 15. The rectifier circuit 12 includes a plurality of diodes connected in a bridge shape.

The voltage sensor 13 and the current sensor 14 are sensors that respectively detect the voltage and current input from the rectifier circuit 12 to the power factor correction circuit 15. Detection values of the voltage sensor 13 and the current sensor 14 are output to the PFC controller 110.

A power factor correction circuit (PFC (Power Factor Correction) circuit) 15 is a step-up chopper circuit in which a coil, a diode, and a transistor are connected, and is a circuit that improves an input power factor to the inverter 18. The power factor correction circuit 15 has a series circuit in which one end of a coil and the anode of a diode are connected, and a parallel circuit of a transistor and a diode. The series circuit is connected to the positive power supply line. The parallel circuit is connected to a connection point between the coil and the diode between the pair of power supply lines.

The operation of the power factor correction circuit 15 will be described.
The target amplitude value of the input current to the power factor correction circuit 15 is calculated based on the target value of the output voltage of the power factor correction circuit 15 and the actual output voltage value (output voltage control). The actual output voltage is a detection voltage of the voltage sensor 16. Based on the calculated amplitude target value of the input current and the actual input voltage to the power factor correction circuit 15, the waveform of the target value of the input current is calculated (input current control). The actual input voltage is a detection voltage of the voltage sensor 13. The actual input current is detected by the current sensor 14, and the duty of the transistor is calculated so that the actual input current matches the target value of the input current. Then, the power factor correction circuit 15 operates by switching on and off of the transistor according to the duty.

The voltage sensor 16 is a sensor that detects the voltage output from the power factor correction circuit 15 to the inverter 18. The detection voltage of the voltage sensor 16 is output to the PFC controller 110.

The smoothing capacitor 17 is a capacitor for smoothing the voltage input from the power factor correction circuit 15 to the inverter 18. The smoothing capacitor 17 is connected to the input side of the inverter 18.

The inverter 18 converts power input from the AC power supply 11 via the rectifier circuit 12 and the like into high-frequency AC power, and outputs AC power to the power transmission coil 19 a included in the primary side resonance circuit 19. Circuit. The inverter 18 is a circuit in which switching elements S ₁ to S ₄ such as IGBTs and free-wheeling diodes D ₁ to D ₄ are connected in parallel, and a parallel circuit of the switching elements and the diodes is connected in series by each arm. The diodes D ₁ to D ₄ are connected so as to be opposite to the direction of the current flowing through the switching elements S ₁ to S ₄ . The input side of the inverter 18 is connected to the power factor correction circuit 15, and the output side is connected to the primary side resonance circuit 19, so that the inverter 18 is connected between the AC power source 11 and the power transmission coil 19a. Yes.

The primary side resonance circuit 19 is a circuit for resonating AC power on the power transmission side. The primary side resonance circuit 19 includes a power transmission coil 19a and a capacitor 19b. The power transmission coil 19a and the capacitor 19b form a series LC resonance circuit. The power transmission coil 19a is a coil for supplying electric power to the power reception coil 21a in a non-contact manner. The power transmission coil 19a is constituted by, for example, a loop-shaped coil having a coil surface along the ground surface of a parking space where a vehicle is parked. When a vehicle including the power receiving coil 21a is parked in the parking space, the power transmitting coil 19a faces the power receiving coil 21a while leaving a predetermined gap. Thereby, between the power transmission coil 19a and the receiving coil 21a is magnetically coupled. Further, due to the resonance action between the primary side resonance circuit 19 and the secondary side resonance circuit 21, power is supplied from the power transmission coil 19a to the power reception coil 21a in a non-contact manner. The primary side resonance circuit 19 is not limited to the above, and may be another resonance circuit.

The ground side controller 100 controls the power supply device on the ground side. The ground side controller 100 includes a PFC controller 110, a wireless communication device 120, and an inverter controller 130. The PFC controller 110 controls the power factor improvement circuit 15 so as to improve the power factor of the output power with respect to the input power to the power factor improvement circuit 15 based on the detection values of the voltage sensor 13 and the current sensor 14. A drive signal is transmitted to the transistor. The power factor improvement circuit 15 operates by the transistor of the power factor improvement circuit 15 being switched on and off according to the drive signal, and the power factor is improved.

The wireless communication device 120 is a communication device for wireless communication with the vehicle-side controller 200. The wireless communication device 120 receives a signal including information such as a command signal for starting charging of the battery 26, an output voltage to the battery 26, and an output current, from the wireless communication device 220 of the vehicle-side controller 200.

The inverter controller 130 controls the inverter 18 based on the charging command and output power received by the wireless communication device 120. The charging command includes a command to start charging and a command indicating required power required for charging the battery 26. The output power is power output from the LC filter 23 to the battery 26.

The detailed configuration of the inverter controller 130 will be described with reference to FIG. FIG. 2 is a block diagram of the inverter controller 130. The inverter controller 130 includes a power feedback control calculation unit 131, a carrier frequency variable unit 132, a carrier signal generation unit 133, a duty calculation unit 134, a carrier comparison value conversion unit 135, and a drive signal generation unit 136. The inverter controller 130 controls the inverter 18 while switching between PAM control for controlling at a single drive frequency and PFM control for controlling the frequency at the time of driving the inverter 18 at a plurality of drive frequencies. .

The power feedback control calculation unit 131 calculates a drive frequency (f _PWM ) at which the output power to the battery 26 becomes the required power from the required power and the actual output power to the battery 26. The drive frequency corresponds to the carrier frequency when controlling the inverter by PAM control or PFM control. The power feedback control calculation unit 131 can set a plurality of frequencies that are required power. Specific control when setting the drive frequency will be described later.

The carrier frequency variable unit 132 selects a drive frequency (hereinafter also referred to as a carrier frequency) when the inverter 18 is actually driven from a plurality of drive frequencies set by the power feedback control calculation unit 131. When controlling the inverter 18 by PAM control, the carrier frequency variable unit 132 selects one frequency as the carrier frequency of the inverter 18 from among a plurality of drive frequencies set by the power feedback control calculation unit 131, The selected carrier frequency is not changed during PAM control.

On the other hand, when the inverter 18 is controlled by PFM control, the carrier frequency variable unit 132 selects one frequency as the carrier frequency of the inverter 18 from among a plurality of drive frequencies set by the power feedback control calculation unit 131. However, the frequency to be selected is varied periodically. Further, the carrier frequency variable unit 132 repeatedly varies the frequency to be selected during PFM control. That is, the carrier frequency variable unit 132 repeatedly varies the carrier frequency at the time of driving the inverter 18 at a plurality of frequencies set by the power feedback control calculation unit 131 during the PFM control. Then, the carrier frequency variable unit 132 outputs the selected frequency (f _D ) to the carrier signal generation unit 133.

The carrier signal generation unit 133 generates a carrier signal having a frequency (f _D ) selected by the carrier variable frequency 132 and outputs the carrier signal to the drive signal generation unit 136.

The duty calculation unit 134 calculates the duty (D) so that the output power to the battery 26 matches the required power indicated by the charge command. The duty is a duty in the PWM control of the inverter 18 and represents the ON period of the switching elements (S ₁ to S ₄ ). The carrier comparison value conversion unit 135 calculates a comparison value (determination voltage) for comparison with the carrier signal from the duty (D), and outputs it to the drive signal generation unit 136. The drive signal generation unit 136 generates a drive signal by comparing the comparison value with the carrier signal. The drive signal is a switching signal used when switching the switching elements S ₁ to S ₄ of the inverter 18 on and off.

When the inverter 18 is controlled by PAM control, the drive signal generation unit 136 compares the carrier signal having the frequency (f _D ) with the comparison value. The frequency (f _D ) is a frequency set by the carrier frequency variable unit 132. Then, the drive signal generation unit 136 outputs a drive signal based on the comparison result to the inverter 18. Thereby, the inverter 18 is driven by PWM control. On the other hand, when the inverter 18 is controlled by PFM control, the drive signal generation unit 136 generates a drive signal while comparing the variable frequency (f _D ) with the carrier frequency. Further, during PAM control, the inverter controller 106 controls the inverter 18 while changing the input voltage to the inverter 18 in conjunction with the PFC controller 110, thereby increasing the output voltage to the battery 26. I have control.

Referring back to FIG. 1, the secondary side configuration will be described. The secondary side resonance circuit 21 is a circuit for resonating the AC power on the power receiving side. The secondary resonance circuit 21 includes a power receiving coil 21a and capacitors 21b and 21c. The power receiving coil 21a and the capacitor 21b are connected in parallel, and a parallel circuit of the power receiving coil 21a and the capacitor 21b is connected in series to the capacitor 21c. The power receiving coil 21a is constituted by, for example, a loop-like coil similar to the power transmitting coil 19a, and is attached so that the coil surface follows the vehicle chassis. The secondary resonance circuit 21 is not limited to the above, and may be another resonance circuit.

The rectifier circuit 22 is a circuit in which a plurality of diodes are connected in a bridge shape, and is connected to the secondary resonance circuit 21 and the LC filter 23. The LC filter 23 is configured by connecting a capacitor between a pair of power supply lines while being connected to the positive electrode side of the power receiving coil 21a and connecting the coil to the power supply line. The voltage sensor 24 is connected to the capacitor of the LC filter 23 and detects the output voltage from the LC filter 23 to the battery 26. The current sensor 25 is connected between the LC filter 23 and the battery 26 and detects an output current from the LC filter 23 to the battery 26. The battery 26 is a power source for supplying electric power to the motor of the vehicle, and includes a plurality of secondary batteries. The battery 26 is electrically connected to the power receiving coil 21a via the LC filter 23 and the like.

The vehicle-side controller 200 controls the vehicle-side power supply device. The vehicle-side controller 200 has a data processing unit 210 and a wireless communication device 220. When the data processing unit 210 acquires a command (charging command) for starting charging of the battery 26 input from the outside, the data processing unit 210 converts the data into data for wireless communication and outputs the charging command to the wireless communication device 220. The charge command includes power suitable for charging the battery 26 as required power. The battery controller that transmits a charging command to the data processing unit 210 is mounted on the vehicle, and calculates the power suitable for charging the battery 26 as the required power while managing the state of the battery 26.

Further, the data processing unit 210 outputs detection values (output voltage and output current to the battery 26) detected by the voltage sensor 24 and the current sensor 25 to the wireless communication device 220 while the battery 26 is being charged. The wireless communication device 220 is a communication device for wirelessly communicating with the ground controller 100.

Next, the leakage magnetic flux and noise when the inverter 18 is controlled at a single frequency will be described as a comparative example. In the comparative example, the controller on the ground side changes the carrier frequency from the high frequency side to the low frequency side, and drives the inverter 18 at the frequency when the output current to the battery 26 matches the current value corresponding to the required power. Set to frequency. Then, after setting the carrier frequency, the controller drives the inverter 18 at a constant single frequency without changing the carrier frequency. Each configuration of the comparative example other than the control of the inverter 18 is the same as the configuration shown in FIG.

FIG. 3 shows the frequency characteristic (a) of the output current (I _out ), the frequency characteristic (b) of the ampere-turn composite value (AT), and the frequency characteristic (c) of the coil current in the comparative example. The current threshold value (I _th ) of the output current is a current value corresponding to the required power with the output voltage to the battery 26 being a constant value. If the output current to the battery 26 is a current threshold value (I _th ), the required power is input to the battery 26.

As shown in FIG. 3, when the carrier frequency reaches f ₁ , the output current to the battery 26 becomes the current threshold value (I _th ), and the controller actually sets the carrier frequency (f ₁ ) to the inverter 18 The inverter 18 is controlled by setting the drive frequency.

For example, if the battery 26 is a lithium ion battery, the SOC of the battery 26 is 50%, and the voltage of the battery 26 is 300 V, the threshold value (I _th ) of the output current is 10 A when the required power is 3 kW. The frequency (f ₁ ) when the current threshold value (I _th ) is reached becomes the carrier frequency when controlling the inverter 18 in the comparative example.

In the comparative example, the coil current (I ₁ , I ₂ ), the ampere turn composite value, and the electric field strength when the inverter 18 is driven at the carrier frequency (f ₁ ) will be described with reference to FIG. 4A is a circuit diagram of the power transmission coil 19a and the power reception coil 21a, FIG. 4B is a diagram showing a vector of the coil current, and FIG. 4C is a graph showing the relationship between the ampere turn composite value and the electric field strength. It is. Further, the phase (θ ₁₂ ) in FIG. 4B is a phase difference of the coil current (I ₂ ) with respect to the coil current (I ₁ ).

As shown in FIG. 4 (a), the current flowing through the transmitting coil 19a and _{I 1,} the current flowing through the power receiving coil 21a and _{I 2,} the number of turns of the transmission coil 19a and receiving coil 21a and the _{N 1} and _{N 2} . The coil current in Figure 4 _(I 1, _{I 2)} is the same as the coil current _(I 1, _{I 2)} shown in FIG.

The ampere turn composite value corresponds to the leakage magnetic field (or leakage electric field) between the coils, and is a value uniquely determined from the coil current. The ampere-turn composite value is represented by the vector represented by the product of the coil current (I ₁ ) and the winding (N ₁ ), and the product of the coil current (I ₂ ) and the winding (N ₂ ). It is determined by the synthesized value of the vector synthesized with the vector (see FIG. 4B). And as shown in FIG.4 (c), a leakage magnetic field (equivalent to the electric field strength shown on the vertical axis | shaft of FIG.4 (c)) becomes large, so that an ampere turn synthetic | combination value is large. Therefore, as in the comparative example, when the coil current (I ₁ , I ₂ ) is large, the leakage magnetic flux increases. Moreover, since the loss of a coil also becomes large, coil efficiency (power transmission efficiency) deteriorates.

Furthermore, since the inverter 18 is driven at a single carrier frequency in the comparative example, the level of EMI noise associated with the switching operation of the switching elements S ₁ to S ₄ increases. This is a case where a communication device that communicates with the outside, such as a radio timepiece, is mounted on the vehicle. When the EMI noise when the inverter 18 is driven based on the carrier frequency (f ₁ ) interferes with the frequency band used in the communication device, the EMI noise is large in the comparative example, so the influence on the communication device is large.

Next, the leakage magnetic flux and noise when the inverter 18 is driven while the carrier frequency of the inverter 18 is varied at a plurality of frequencies as in the present invention will be described with reference to FIG. FIG. 5 shows the frequency characteristic (a) of the output current (I _out ), the frequency characteristic (b) of the ampere-turn composite value (AT), the frequency characteristic (c) of the coil current, and the characteristic (d _in ) of the input impedance (Z _in ). ). The input impedance (Z _in ) is an impedance when the primary side resonance circuit 19 is viewed from the output side of the inverter 18, and is an impedance obtained by combining the primary side resonance circuit 19 and the secondary side resonance circuit 21.

In the present invention, the controller 100 on the ground side sets a plurality of frequencies when the output current to the battery 26 matches the current value corresponding to the required power while changing the carrier frequency from the high frequency side to the low frequency side.

As shown in FIG. 5, when the carrier frequency is f ₁ when the frequency is varied from the high frequency side to the low frequency side, the output current (I _out ) of the battery 26 becomes the current threshold value (I _th ). If the carrier frequency is further varied to the low frequency side, the output current (I _out ) of the battery 26 becomes the current threshold value (I _th ) when the carrier frequency reaches f ₂ . Thereby, the carrier frequency (f ₁ , f ₂ ) becomes a discrete frequency.

Using FIG. 6, the ampere turn composite value when the inverter 18 is driven with the carrier frequency set to f ₁ and the ampere turn when the inverter 18 is driven with the carrier frequency set to f _2. The composite value will be described. 6 (a) shows a vector diagram of ampere-turns synthesis value when the carrier frequency is f _1, it shows a vector diagram of (b) is ampere-turns synthesis value when the carrier frequency is f _2. The phase difference (θ ₁₂ ) in FIG. 6A represents the phase (θ ₁₂ ) with respect to the frequency (f ₁ ) in the phase characteristics in FIG. 5D, and the phase difference ( θ ₁₂ ) represents the phase (θ ₁₂ ) with respect to the frequency (f ₂ ) in the phase characteristics of FIG.

When the carrier frequency is f ₁ , the ampere-turn composite value when the carrier frequency is f ₂ is shown in FIGS. 6 (a) and 6 (b). When the carrier frequency is f ₂ , the phase difference (θ ₁₂ ) of the coil current is almost the same as the phase difference (θ ₁₂ ) when the carrier frequency is f ₁ when viewed in absolute value. . On the other hand, the coil currents (I ₁ , I ₂ ) are smaller than the coil currents (I ₁ , I ₂ ) when the carrier frequency is f ₁ . For this reason, the ampere-turn composite value when the carrier frequency is f ₂ is smaller than the ampere-turn composite value when the carrier frequency is f ₁ .

The magnitude of the coil current (I ₁ , I ₂ ) when the inverter 18 is driven with a single carrier frequency (f ₁ , f ₂ ) while setting the carrier frequency to the frequency (f ₁ , f ₂ ), respectively. FIG. 7A and FIG. 7B show the magnitude of the ampere turn composite value (AT), respectively.

The carrier frequency in comparison with the case of the f _1, when the carrier frequency is f _2, the coil current (I _{1, I 2)} is reduced, ampere turns composite value (AT) is also reduced. From this point, by setting the carrier frequency to the frequency (f ₂ ) and driving the inverter 18 with a single carrier frequency (f ₂ ), the ampere-turn composite value (AT) should be smaller than that of the comparative example. Can do.

However, when the carrier frequency is set to the frequency (f ₂ ), the phase of the coil current (I ₁ , I ₂ ) is advanced (see FIG. 5 (d)), and therefore the diode D _{1 of the} inverter 18. calorific value of ~ _{D 4} increases.

The relationship between the phase of the coil current (I ₁ , I ₂ ) and the amount of heat of the diodes D ₁ to D ₄ will be described with reference to FIG. FIG. 8 is a circuit diagram of the inverter 18. In FIG. 8, the load represents a load (including the primary side resonance circuit 19, the secondary side resonance circuit, the battery 26, etc.) connected to the output side of the inverter 18.

When the phase of the coil current output from the inverter 18 is advanced, the load on the output side of the inverter 18 appears to be a capacitive load. Therefore, the load on the output side of the inverter 18 accumulates energy. Immediately after turn-off of the switching elements _S 1 ~ _{S 4} of the inverter 18, the recovery current is generated by the stored energy in the load, the recovery current flows as a return current to the diodes _D 1 ~ _{D 4,} diodes _D 1 ~ _{D 4} Generates heat.

In the diodes D ₁ to D ₄ , allowable power values are set in advance according to the characteristics of the elements. Then, the recovery current flows through the diodes D _{1 ~} D _4, when the power value of the diode D _{1 ~} D ₄ becomes higher than the allowable value, the heating value of the diode D _{1 ~} D ₄ exceeds the allowable value As a result, there is a high possibility that abnormality occurs in the diodes D ₁ to D ₄ . Thus, in the state of phase advance phase of the coil current, when continued to drive the inverter 18, the recovery current flowing through the diode D _{1 ~} D _4, the heating value of the diode D _{1 ~} D ₄ is increased . Therefore, it is not preferable to drive the inverter 18 for a long time with a single carrier frequency having a frequency (f ₂ ).

Hereinafter, the control of the non-contact power feeding system will be described using the flowchart shown in FIG. FIG. 9 is a flowchart showing the control procedure of the controller of the non-contact power feeding system. A specific control flow will be described with an example in which the characteristics such as the output current to the battery 26 have the characteristics shown in FIG. The flowchart shown in FIG. 9 is executed while looping from the start to the end of charging of the battery.

In step S1, when the ground-side controller 100 receives the charging start command and the required power for charging the battery 26 from the vehicle-side controller 200 by wireless communication, the ground-side controller 100 drives the inverter 18 by PAM control. While driving the inverter 18, the vehicle-side controller 200 transmits the detected voltage and the detected current to the ground-side controller 100 by wireless communication while detecting the voltage and current output to the battery 26 by the voltage sensor 24 and the current sensor 25. . In addition, the vehicle-side controller 200 transmits a charging command including the output voltage to the battery 26, the output current, and the required power to the ground-side controller 100. The ground-side controller 100 controls the power factor correction circuit 15 and the inverter 18 so that the output power to the battery 26 matches the required power.

In step S2, the ground-side controller 100 determines whether to switch from PAM control to PFM control. The time zone for performing the PFM control is determined in advance, and is set according to the time zone for receiving radio waves in order to adjust the radio clock, for example. If the current time falls within the PFM control time zone, the ground controller 100 switches the control by the inverter controller 130 from PAM control to PFM control, and proceeds to step S3. On the other hand, when the current time is not the PFM control time zone, the ground-side controller 100 continues the PAM control.

In step S3, inverter controller 130 acquires output power from vehicle-side controller 200 to battery 26 while changing the carrier frequency from the high frequency side to the low frequency side. The inverter controller 130 identifies the carrier frequency that is the required power while comparing the output power to the battery 26 and the required power while the carrier frequency is variable. At this time, the specified frequency is a discrete frequency. In step S3, the inverter controller 130 controls the inverter 18 to energize at a low output before charging the battery 26. Then, the inverter controller 130 varies the carrier frequency and sets a plurality of carrier frequencies in a state where the output from the inverter 18 is lower than the output when the battery 26 is charged. In the example of FIG. 5, the power feedback control calculation unit 131 sets a plurality of frequencies (f ₁ , f ₂ ) as carrier frequencies.

In step S4, the power feedback control calculation unit 131 calculates the phase of the coil current when the inverter 18 is driven at the set carrier frequency. The phase of the coil current is calculated from the phase characteristic of the input impedance (Z _in ) with respect to the output of the inverter 18. Then, the power feedback control calculation unit 131 identifies whether the phase corresponding to the set carrier frequency is a fast phase or a slow phase. In the example of FIG. 5, the power feedback control calculation unit 131 specifies the phase corresponding to the carrier frequency (f ₁ ) as a slow phase and specifies the phase corresponding to the carrier frequency (f ₂ ) as a fast phase.

In step S <b> 5, the carrier frequency variable unit 132 of the inverter controller 130 selects one frequency as the carrier frequency of the inverter 18 from among the plurality of drive frequencies set by the power feedback control calculation unit 131. Then, the inverter controller 130 drives the inverter 18 with the selected carrier frequency. The output of the inverter 18 is larger than the output when the carrier frequency is set in step S3. Further, the carrier frequency varying unit 132 varies the carrier frequency at a predetermined period. The predetermined period is determined by the reciprocal of a plurality of set drive frequencies.

FIG. 10 shows the time characteristics of the carrier signal. Graph a shows the characteristics of the carrier frequency for PFM control, and graph b shows the characteristics of the carrier frequency for PAM control. In PAM control, the carrier frequency is fixed at the frequency (f ₁ ). On the other hand, in the PFM control, the carrier frequency varying unit 132 varies the carrier frequency from f ₁ to f ₂ when the period (1 / f ₁ ) elapses, and the period (1 / f ₂ ) elapses from the frequency variation time. the carrier frequency at the time of the variable from f ₂ to f _1. Then, the carrier frequency variable unit 132 repeatedly varies the carrier frequency at the time of the period (1 / f ₁ ) and the period (1 / f ₂ ). By repeatedly changing the carrier frequency, the inverter 18 is driven a plurality of times at the carrier frequency (f ₁ ) and is driven a plurality of times at the carrier frequency (f ₂ ) during PFM control.

In step S6, the inverter controller 130 calculates the current value of each of the diodes D ₁ to D ₄ from the input current to the inverter 18 and the switching waveform of the switching elements S ₁ to S ₄ , and diodes D ₁ to current value of D ₄ and calculates the power value of the diode _D 1 ~ _{D 4} of the diode of the on resistance _D 1 ~ _{D 4.} Then, the inverter controller 130 determines whether or not the power values of the diodes D ₁ to D ₄ are equal to or less than an allowable value. If the power value is greater than the allowable value, the process proceeds to step S7, and if the power value is less than the allowable value, the process proceeds to step S8. The allowable value is a value determined by the specification values (heat generation upper limit value) of the elements of the diodes D1 to D4, and is set to, for example, 80% of the specification value.

When the power values of the diodes D ₁ to D ₄ are larger than the allowable values in step S 7, the inverter controller 130 causes the carrier frequency variable unit 132 to change only the carrier frequency at the slow phase to the carrier at the time of driving the inverter 18. The inverter 18 is controlled while selecting the frequency. Then, the process proceeds to step S8. In the example of FIG. 5, the inverter controller 130 controls the inverter 18 only with the carrier frequency (f ₁ ). When there are a plurality of slow-phase frequencies, the carrier frequency varying unit 132 varies the carrier frequency only with the plurality of slow-phase frequencies.

In step S8, the ground-side controller 100 determines whether to switch from PFM control to PAM control. If the current time is in the PFM control time zone, the ground-side controller 100 returns to step S5 and continues PFM control. On the other hand, when the current time is out of the time zone of the PFM control, the ground-side controller 100 returns the control of the inverter 18 from the PFM control to the PAM control, and the control flow of FIG.

As described above, in the present embodiment, the non-contact power feeding apparatus controls the inverter 18 by repeatedly varying the carrier frequency with a plurality of seed wave numbers. The time transition of the carrier frequency of the inverter 18 is represented by the graph of FIG. In FIG. 11, PAM control is performed in the time zone from time (0) to time (t ₁ ) and the time zone after time (t ₂ ), and the time from time (t ₁ ) to time (t ₂ ). The band performs PFM control. In FIG. 11, the carrier frequency during PAM control is set to f _A , but the carrier frequency during PAM control may be a frequency other than f _A or f ₁ .

From time (0) to time (t ₁ ) and after time (t ₂ ), since the inverter 18 is controlled by PAM control, the carrier frequency is a fixed value of f _A. Between time (t ₁ ) and time (t ₂ ), since the inverter 18 is controlled by PFM control, the carrier frequency is a timing determined by the period (1 / f ₁ ) and the period (1 / f ₂ ). alternately switched at f ₁ and _{f 2.} That is, the carrier frequency is continuously changed. When the carrier frequency is varied by f ₁ and f ₂ , the modulation frequency (1 / Tm: Tm = 1 / f ₁ + 1 / f ₂ ) is set to 1 kHz or more. Thereby, it is possible to avoid a radio reception frequency (audible frequency band).

Next, the coil loss, the ampere turn composite value, and the EMI noise in the contactless power supply system of the present invention will be described in comparison with the contactless power supply system of the comparative example. FIG. 12A is a graph showing the coil loss, and FIG. 12B is a graph showing the ampere turn composite value. FIG. 13A is a graph showing the frequency characteristics of the EMI noise of the comparative example, and FIG. 13B is a graph showing the frequency characteristics of the EMI noise of the present invention. FIG. 13C is a graph showing the maximum value of the EMI level. The non-contact power feeding system of the comparative example shows characteristics when the inverter 18 is driven with a single carrier frequency (f ₁ ).

As shown in FIG. 12 (a), in a non-contact power supply system of the present invention, the carrier frequency (f ₁₎ and the carrier frequency (f ₂₎ by varying, it is possible to suppress the coil loss than the comparative example. Further, as described above, when the inverter 18 is driven at the carrier frequency (f ₂ ), the ampere turn composite value is smaller than the ampere turn composite value at the carrier frequency (f ₁ ) (see FIG. 6). Since the contactless power feeding system of the present invention includes not only the carrier frequency (f ₁ ) but also the carrier frequency (f ₂ ) in the selectable frequency, the ampere-turn composite value can be made smaller than that of the comparative example.

Further, as shown in FIG. 13A, in the non-contact power feeding system of the comparative example, noise corresponding to the harmonic of the carrier frequency (f ₁ ) is generated at a high value. On the other hand, the contactless power feeding system of the present invention drives the inverter 18 at the carrier frequency (f ₁ , f ₂ ), so that energy can be dispersed. Therefore, as shown in FIG. 13B, the noise level corresponding to the harmonics of the carrier frequency (f ₁ ) and the carrier frequency (f ₂ ) becomes small. And as shown in FIG.13 (c), the maximum value of the EMI noise level of the non-contact electric power feeding system of this invention can be made smaller than a comparative example.

As described above, in the present embodiment, a plurality of drive frequencies of the inverter 18 are set, and the inverter 18 is driven by repeatedly varying the drive frequency of the inverter 18 at the set drive frequencies. Thereby, compared with the case where the inverter 18 is driven only by a single drive frequency, the ampere-turn composite value can be controlled, so that the leakage magnetic field (leakage electric field) can be reduced. Moreover, since coil loss is reduced, the efficiency at the time of power transmission can be improved. Moreover, EMI noise accompanying switching of the inverter 18 can be reduced.

In the present embodiment, the drive frequency at which the power supplied from the power receiving coil 21 a to the battery 26 becomes the required power to the battery 26 is set to the frequency when the inverter 18 is driven. Thereby, the leakage magnetic field can be reduced while increasing the output power to the battery 26 to the required power.

In the present embodiment, PFM control for controlling the inverter 18 while repeatedly varying at a plurality of drive frequencies and PAM control for controlling the inverter at a single drive frequency are switched. As a result, the control mode can be switched in accordance with a time zone in which noise is not desired to be interfered, such as a time zone in which a radio wave for adjusting a radio timepiece is received.

In the present embodiment, the drive frequency at which the phase of the output current of the inverter 18 is advanced is set to the frequency at the time of driving the inverter 18 according to the power values of the diodes D ₁ to D ₄ . Thereby, even when the carrier frequency satisfying the required power is a frequency at the time of phase advance, the carrier frequency can be used for driving the inverter 18, so that the leakage magnetic flux can be reduced.

In the first embodiment, when the power value of the diode D _{1 ~} D ₄ is equal to or less than the allowable value of power allowed to the diode D _{1 ~} D ₄ is driven phase of the output current of the inverter 18 becomes the leading phase The frequency is set to the frequency when the inverter 18 is driven. On the other hand, when the power value is larger than the allowable value, only the drive frequency at which the phase of the output current of the inverter 18 is delayed is set as the frequency at the time of driving the inverter. As a result, the leakage magnetic flux can be reduced while suppressing the heat generation of the diodes D ₁ to D ₄ .

In the embodiment, the inverter 18 is driven while periodically changing a plurality of drive frequencies. Thereby, the energy of noise generation can be evenly distributed.

In the present embodiment, two frequencies (f ₁ , f ₂ ) are set as carrier frequencies in the PFM control, but the number of carrier frequencies may be three or more. In addition, for example, when the power feedback control calculation unit 131 sets three frequencies (f _x , f _y , f _z ) as carrier frequencies, the carrier frequencies may be varied randomly. For example, in the PFM control, the carrier frequency _{is, f x, f y, f} z, f x, f z, in the order of _{f x,} may be variable. Thereby, the carrier frequency at the time of driving the inverter 18 is repeatedly varied.

The above-mentioned ground side controller 100 corresponds to the “control means” of the present invention.

<< Second Embodiment >>
A non-contact power feeding system according to another embodiment of the invention will be described. In this example, in contrast to the first embodiment described above, when the inverter 18 is driven by PFM control, only the drive frequency at which the phase of the output current of the inverter 18 is delayed is set as the frequency at the time of driving the inverter. The point is different. Other configurations are the same as those in the first embodiment described above, and the description thereof is incorporated.

Hereinafter, the control of the non-contact power feeding system will be described using the flowchart shown in FIG. The control in steps S11 and S12 is the same as that in steps S1 and S2 according to the first embodiment, and the control in steps S18 and S19 is the same as that in steps S8 and S9 according to the first embodiment. Is omitted.

In step S13, the inverter controller 130 causes the carrier frequency variable unit 132 to drive the inverter 18 while varying the carrier frequency from the high frequency side to the low frequency side, and obtains output power from the vehicle-side controller 200 to the battery 26. To do. When changing the carrier frequency, the upper limit frequency and the lower limit frequency are determined in advance, and the inverter controller 130 changes the carrier frequency from the upper limit frequency to the lower limit frequency. Then, the power feedback control calculation unit 131 of the inverter controller 130 specifies all carrier frequencies at which the output power to the battery 26 matches the required power when the carrier frequency is varied from the upper limit frequency to the lower limit frequency. .

The above control will be described with a specific example. FIG. 15A is a graph showing the frequency characteristics of the input impedance (Z _in ) and the phase, FIG. 15B is a graph showing the frequency characteristics of the output current of the inverter 18, and FIG. It is a graph which shows a frequency characteristic. The solid line in FIG. 15A indicates the input impedance characteristic, and the dotted line indicates the phase characteristic. The characteristics shown in FIG. 15 are merely examples, and differ depending on the battery capacity, the circuit parameters of the resonance circuit, and the like.

In the example of FIG. 15, the inverter controller 130 varies the carrier frequency from the high frequency side to the low frequency side, and specifies _four frequencies (f ₁ , f ₂ , f ₃ , f ₄ ) that match the required power. .

In step S14, the power feedback control calculation unit 131 calculates a phase corresponding to the specified frequency from the phase characteristic of the input impedance (Z _in ), and determines whether the phase is a leading phase or a lagging phase. To do. As shown in FIG. 15, the phases corresponding to the carrier frequencies (f ₁ , f ₂ , f ₃ ) are delayed, and the phases corresponding to the carrier frequency (f ₄ ) are advanced.

In step S15, the power feedback control calculation unit 131 calculates an ampere-turn combined value corresponding to each carrier frequency (f ₁ , f ₂ , f ₃ , f ₄ ). The ampere-turn composite value is calculated from the phase difference (θ ₁₂ ) between the coil current (I ₁ , I ₂ ) and the coil current when driven at each carrier frequency (f ₁ , f ₂ , f ₃ , f ₄ ). . The inverter controller 130 is preset with a regulation value (upper limit value) of the ampere-turn composite value. The regulation value defines the upper limit value of the EMI noise level, and is set according to the upper limit of the electric field strength regulated by the law. For example, in Japan, since the electric field intensity at a position 30 m away from the specimen is regulated by the Radio Law, the regulation value is determined according to the value regulated by the Radio Law.

In addition, the power feedback control calculation unit 131 compares the ampere-turn composite value corresponding to each carrier frequency (f ₁ , f ₂ , f ₃ , f ₄ ) with the regulation value, and identifies the carrier frequency that is equal to or less than the regulation value. To do. As shown in FIG. 15C, when the regulation value is set, the carrier frequencies that are equal to or less than the regulation value are f ₁ , f ₂ , and f ₃ . The relationship between each carrier frequency (f ₁ , f ₂ , f ₃ , f ₄ ), the phase, and the ampere turn composite value is expressed as a table in FIG. Note that “OK” in FIG. 16 indicates that the ampere-turn composite value is less than or equal to the regulation value, and “NO” indicates that the ampere-turn synthesis value is greater than the regulation value.

Here, the ampere-turn composite value for each carrier frequency (f ₁ , f ₂ , f ₃ ) will be described with reference to FIG. FIG. 17A shows coil currents (I ₁ , I ₂ ) when the inverter 18 is driven at each carrier frequency (f ₁ , f ₂ , f ₃ ), and FIG. Show. The power output to the battery 26, since the same in each carrier frequency _{(f 1, f 2, f} 3), the coil current _{(I 2)} flowing through the power receiving coil 21a are the same. On the other hand, the coil current (I ₁ ) flowing through the power transmission coil 19a has different magnitudes depending on the phase difference between the currents. The ampere-turn composite value for each carrier frequency (f ₁ , f ₂ , f ₃ ) becomes a different value due to the difference in phase and the magnitude of the coil current (I ₁ ).

Returning to FIG. 14, in step S < _b > 16, the power feedback control calculation unit 131 selects a frequency satisfying the condition among the carrier frequencies (f ₁ , f ₂ , f ₃ , and f ₄ ) that match the required power by the inverter 18. Set to the driving frequency. The conditions are that the phase corresponding to the carrier frequency is slow, and that the ampere-turn composite value corresponding to the carrier frequency is equal to or less than the regulation value. In the example of FIG. 15, a plurality of carrier frequencies (f ₁ , f ₂ , f ₃ ) are set as frequencies when the inverter 18 is driven. The carrier frequency variable unit 132 of the inverter controller 130 drives the inverter 18 while being varied at the set carrier frequencies (f ₁ , f ₂ , f ₃ ). Then, the process proceeds to step S17.

The time characteristics of the carrier signal when the inverter 18 is driven are shown in FIG. In FIG. 18, the graph a shows the carrier frequency characteristic of PFM control, and the graph b shows the carrier frequency characteristic of PAM control. In PAM control, the carrier frequency is fixed at the frequency (f ₁ ). On the other hand, in the PFM control, the carrier frequency varying unit 132 varies the carrier frequency from f ₁ to f ₂ when the period (1 / f ₁ ) elapses, and the period (1 / f ₂ ) elapses from the frequency variation time. the carrier frequency at which the variably from f ₂ to f _3, to vary the carrier frequency at the time when elapsed period (1 / f ₃₎ from the variable time frequency from f ₃ to f _1. The carrier frequency variable unit 132 repeatedly varies the carrier frequency at the period (1 / f ₁ ), the period (1 / f ₂ ), and the period (1 / f ₃ ).

FIG. 19 shows the time characteristics of the carrier frequency when the inverter 18 is driven. As shown in FIG. 19, the carrier frequency is a periodic function of a period (Tm), and is varied so as to make a transition from a low frequency to a high frequency in order from a high frequency to a low frequency. As a result, frequency transition time can be shortened, and energy and EMI noise can be dispersed.

Next, the coil loss, the ampere turn composite value, and the EMI noise in the contactless power supply system of the present embodiment will be described in comparison with the contactless power supply system of the comparative example. FIG. 20A is a graph showing the coil loss, and FIG. 20B is a graph showing the ampere turn composite value. FIG. 21A is a graph showing the frequency characteristics of the EMI noise of the comparative example, and FIG. 21B is a graph showing the frequency characteristics of the EMI noise of the present invention. FIG. 21C is a graph showing the maximum value of the EMI level. The non-contact power feeding system of the comparative example shows characteristics when the inverter 18 is driven with a single carrier frequency (f ₁ ).

As shown in FIG. 20A, in the non-contact power feeding system of the present embodiment, the coil loss can be suppressed more than in the comparative example by making the frequency variable by the carrier frequency (f ₁ , f ₂ , f ₃ ). The amperage composite value of the present invention is higher than that of the comparative example, but is kept below the regulation value.

Further, as shown in FIG. 21A, in the non-contact power feeding system of the comparative example, noise corresponding to the harmonic of the carrier frequency (f ₁ ) is generated at a high value. On the other hand, since the contactless power feeding system of the present embodiment drives the inverter 18 at the carrier frequency (f ₁ , f ₂ , f ₃ ), as shown in FIG. 21 (b), the carrier frequency (f ₁ , The noise level corresponding to the harmonics of f ₂ , f ₃ ) is reduced. And as shown in FIG.21 (c), the maximum value of the EMI noise level of the non-contact electric power feeder of this embodiment can be made smaller than a comparative example.

As described above, in the present embodiment, only the drive frequency (carrier frequency) at which the phase of the output current of the inverter 18 is delayed is set as the frequency when the inverter 18 is driven. Thereby, in this embodiment, the heat generation of the diodes D ₁ to D ₄ is suppressed, and the leakage magnetic flux is reduced.

As another specific example of the present embodiment, when the inverter 18 is driven by PFM control, the phase of the output current of the inverter 18 advances even if the ampere-turn composite value corresponding to the carrier frequency is equal to or less than the regulation value. If it is a phase, the above conditions are not satisfied. Therefore, the phase advance carrier frequency is excluded from the frequency set at the time of driving. Hereinafter, another specific example will be described with reference to FIGS.

22A to 22C are graphs showing the frequency characteristics of the input impedance (Z _in ) and the phase, the frequency characteristics of the output current of the inverter 18, and the frequency characteristics of the ampere-turn composite value. The solid line in FIG. 22A indicates the input impedance characteristic, and the dotted line indicates the phase characteristic.

As shown in FIG. 22, when four frequencies (f ₁ , f ₂ , f ₃ , f ₄ ) that match the required power are specified, the phases corresponding to the carrier frequencies (f ₁ , f ₃ ) are respectively The phase is delayed, and the phase corresponding to the carrier frequency (f ₂ , f ₄ ) is advanced. In addition, the ampere turn composite value corresponding to the carrier frequency (f ₁ , f ₂ , f ₃ ) is equal to or lower than the regulation value, and the ampere turn synthesis value corresponding to the carrier frequency (f ₄ ) is higher than the regulation value. The relationship between each carrier frequency (f ₁ , f ₂ , f ₃ , f ₄ ), the phase, and the ampere turn composite value is expressed as shown in the table of FIG.

The ampere-turn composite value corresponding to the carrier frequency (f ₂ ) is equal to or less than the regulation value, and the ampere-turn composite value condition is satisfied. However, since the phase corresponding to the carrier frequency (f ₂ ) is a leading phase, the phase condition is not satisfied. Therefore, the power feedback control calculation unit 131 sets the carrier frequency (f ₁ , f ₃ ) to the frequency at the time of driving the inverter 18 while excluding the carrier frequency (f ₂ ) from the carrier frequency at the time of driving the inverter 18. . Then, the inverter controller 130 drives the inverter 18 while varying the set carrier frequency (f ₁ , f ₃ ).

As a modification of the present embodiment, when the carrier frequency varying unit 132 varies the carrier frequency that satisfies the condition (the carrier frequency set in step 16 above), as shown in FIG. f ₁ , 1 / f ₂ , 1 / f ₃ ), each of which is an integer multiple (however, 2 or more), at a timing for each of a plurality of periods, from a high frequency to a low frequency, or from a low frequency to a high frequency, You may make a transition. As a result, frequency transition time can be shortened, and energy and EMI noise can be dispersed.

<< Third Embodiment >>
A non-contact power feeding system according to another embodiment of the invention will be described. In this example, when the inverter 18 is driven by PFM control with respect to the second embodiment described above, if the ampere-turn composite value corresponding to the carrier frequency is less than or equal to the regulation value, the phase of the output current of the inverter 18 is Even in the advanced phase, the carrier frequency is set to the frequency at the time of driving the inverter. Other configurations are the same as those of the second embodiment described above, and the description thereof is incorporated.

As an example, FIGS. 25A to 25C are graphs showing the frequency characteristics of the input impedance (Z _in ) and the phase, the frequency characteristics of the output current of the inverter 18, and the frequency characteristics of the ampere-turn composite value. The solid line in FIG. 25A indicates the input impedance characteristic, and the dotted line indicates the phase characteristic.

As shown in FIG. 25, when four frequencies (f ₁ , f ₂ , f ₃ , f ₄ ) that match the required power are specified, the phases corresponding to the carrier frequencies (f ₁ , f ₃ ) are respectively The phase is delayed, and the phase corresponding to the carrier frequency (f ₂ , f ₄ ) is advanced. In addition, the ampere turn composite value corresponding to the carrier frequency (f ₁ , f ₂ , f ₃ ) is equal to or lower than the regulation value, and the ampere turn synthesis value corresponding to the carrier frequency (f ₄ ) is higher than the regulation value. The relationship between each carrier frequency (f ₁ , f ₂ , f ₃ , f ₄ ), the phase, and the ampere turn composite value is expressed as a table in FIG.

As in the second embodiment, the power feedback control calculation unit 131 selects a frequency satisfying a condition among carrier frequencies (f ₁ , f ₂ , f ₃ , f ₄ ) that matches the required power when the inverter 18 is driven. Set the frequency to. The condition is that the ampere-turn composite value corresponding to the carrier frequency is less than or equal to the regulation value, and no phase condition is imposed.

Therefore, the power feedback control calculation unit 131, of the carrier frequency _{(f 1, f 2, f} 3, f 4), the carrier frequency _{(f 1, f 2, f} 3), the frequency at the time of driving of the inverter 18 Set. Then, the inverter controller 130 drives the inverter 18 while being varied at a plurality of carrier frequencies (f ₁ , f ₂ , f ₃ ).

The inverter controller 130 compares the power value and the allowable value while calculating the power values of the diodes D ₁ to D ₄ while the inverter 18 is being driven. When the power values of the diodes D ₁ to D ₄ are larger than the allowable value, the inverter controller 130 controls the inverter 18 while selecting only the carrier phase of the slow phase as the carrier frequency when the inverter 18 is driven.

On the other hand, when the power values of the diodes D ₁ to D ₄ are equal to or less than the allowable value, the inverter controller 130 selects the carrier frequency at the time of driving the inverter 18 including the advanced carrier frequency, and the inverter 18 To control. Thereby, the present invention can reduce the leakage magnetic flux while suppressing the heat generation of the diodes D ₁ to D ₄ .

DESCRIPTION OF SYMBOLS 11 ... AC power supply 19 ... Primary side resonance circuit 19a ... Power transmission coil 19b ... Capacitor 21 ... Secondary side resonance circuit 21a ... Power reception coil 21b, c ... Capacitor 26 ... Battery 100 ... Ground side controller 130 ... Inverter controller 200 ... Car Side controller

Claims (7)

A power transmission coil that transmits power in a contactless manner to a power reception coil electrically connected to a load;
An inverter that is connected between the power transmission coil and a power source, converts the power of the power source into AC power, and outputs the AC power to the power transmission coil;
Control means for driving the inverter by frequency variable control for setting a plurality of drive frequencies of the inverter and varying the frequency at the time of driving the inverter;
The plurality of drive frequencies are discrete frequencies,
The non-contact power feeding apparatus, wherein the control means repeatedly varies the frequency at the time of driving the inverter at the plurality of drive frequencies during the frequency variable control. In the non-contact electric power feeder of Claim 1,
The non-contact power feeding apparatus, wherein the control means sets the drive frequency at which the power supplied from the power receiving coil to the load becomes the required power to the load, as a frequency at the time of driving the inverter. In the non-contact electric power feeder of Claim 1 or 2,
The control means includes
A contactless power feeding device that switches between control for driving the inverter and frequency variable control while setting a frequency at the time of driving the inverter to the single drive frequency. The contactless power feeding device according to any one of claims 1 to 3,
The inverter is
A plurality of switching elements provided between a positive electrode and a negative electrode of the power source;
A plurality of diodes connected in parallel to each of the plurality of switching elements in a direction opposite to the direction of the current flowing through the switching elements;
The control means includes
The non-contact power feeding apparatus according to claim 1, wherein the driving frequency at which the phase of the output of the inverter is advanced is set to a frequency at the time of driving the inverter according to a power value of the diode. In the non-contact electric power feeder of Claim 4,
The control means includes
When the power value is less than the allowable power value allowed for the diode, the driving frequency at which the phase of the output of the inverter is advanced is set to the frequency at the time of driving the inverter,
When the power value is equal to or greater than the allowable value, only the drive frequency at which the phase of the output of the inverter is delayed is set as a frequency at the time of driving the inverter. The contactless power feeding device according to any one of claims 1 to 3,
The non-contact power feeding apparatus, wherein the control means sets only the driving frequency at which the phase of the output of the inverter is delayed to the frequency at the time of driving the inverter. The contactless power feeding device according to any one of claims 1 to 6,
The non-contact power feeding apparatus, wherein the control means drives the inverter by periodically varying the plurality of driving frequencies.

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