RU2027195C1 - Satellite system for determining location of ships and airplanes being wrecked - Google Patents

Abstract

FIELD: determination of location of ships and airplanes being damaged. SUBSTANCE: system has radio buoy, transport facility radio transmitters, artificial Earth satellite, board receiving devices, on-board memorizing devices, on-board transmitter, data reception station, receiver, data processing unit, device for matching with communication lines, control unit, search-rescue organizations control communication unit. EFFECT: improved precision; improved reliability. 3 cl, 8 dwg

Description

 The satellite system for determining the location of ships and aircraft in an accident, COSPAS - SARSAT is designed to determine the location of emergency beacons (ARB) transmitting radio signals at a frequency of 121.5 MHz and in the frequency range 406 - 406.1 MHz.

 The INMARSAT satellite system provides various types of services for use in the Global Maritime Communications System, including distress alerts and telephony, letterpress, data and facsimile communications.

 The INTELSAT VI satellite communication system consists of ten independent repeaters - one for each beam of the communication antenna.

 GLOMAR is a promising satellite communications system with mobile objects in the frequency range 1.5-1.6 MHz.

 The LOXSTAR system is designed for the location of moving objects and relaying of radio messages.

 Of the well-known satellite communication systems, the COSPAS-SARSAT system, which is a joint international satellite search and rescue system developed and currently managed by organizations in Canada, France, the USA and Russia, was chosen as the base.

 However, this system can also be used to detect vehicles stolen by intruders or only being hijacked. This problem has now become very relevant. So, every year in the United States with 140 million cars more than 1820 thousand thefts and thefts are committed. In Russia, with 17 million passenger cars, 122 thousand thefts and thefts are committed.

 Known anti-theft devices for vehicles are based on locking the steering shaft of a vehicle (ed. St. USSR NN 1232528, 1484289; UK patents NN 2180208, 2177664; French patent N 2180533; Japan patents NN 62-11691, 62-5095; US patent N 4678068 and others on locking the brake system (ed. St. USSR N 1437270; UK patent N 2182184; US patent N 4678068; German patent N 3622347 and others), on the engine lock (ed. St. USSR NN 1355521, 1337294; US patent N 4668874; Germany patent N 3607784 and others); on blocking the power circuit and the sound signal (ed. St. USSR NN 600010, 893633, 937248, 1404387; US patents NN 4619603, 4710745 and others), on the overlap of the fuel system (ed. St. USSR N 1355521, FRG patents NN 3622793, 3605229; French patent N 2591165 and others), on the use of a combination lock (ed. . USSR N 1342785; French patent N 2591973; UK patent N 2179482; US patent N 4710745 and others).

 The aim of the invention is to expand the functionality of the system by searching, detecting and locating vehicles stolen by intruders or just being stolen.

 The goal is achieved by the fact that the vehicle’s radio sensors and a third on-board receiver and a second on-board memory device are connected in series, the output of which is connected to the fourth input of the on-board transmitter, the fifth input of which is connected to the output of the third on-board receiver, and also connected to the output of the receiver the device is a third information processing device, the output of which is connected to the third communication device.

 The COSPAR-SARSAT satellite system has a block diagram shown in FIG. 1; the structural diagram of the vehicle radio sensor is shown in figure 2; a frequency diagram explaining the formation of additional (mirror and Raman) reception channels is shown in FIG. 3; the principle of direction finding of a vehicle’s radio sensor in one plane by the phase method is shown in FIG. 4; the truth table corresponding to the elimination of the ambiguity of direction finding to the deviation of the radio sensor from the equal-signal direction, is presented in figure 5; the structural diagram of the third on-board receiving device is shown in Fig.6, diagrams explaining the operation of the system are presented in Fig.7; The circuit of the information receiving point is shown in FIG.

The system contains the first 1 and second 2 emergency beacons, vehicle radio sensor 3, artificial earth satellite (AES) 4, first 5, second 6 and third 7 on-board receivers, first on-board storage device 8, on-board storage device 9, second storage device 10 , on-board transmitter 11, information reception (FDI) point 12, receiving device 13, first 14, second 15 and third 16 information processing devices, communication network interface device 17, system monitoring and control device 18, communication device 19 search and rescue organizations, power supply 20, resistor 21, LED indicator 22, remote switch 23, first 24 and second 25 windings of remote switch 23, reed switch 26, ignition key 27, modulating code generator 28, relay 29, master oscillator 30, phase a manipulator 31, a transmitter 32 with an antenna, first 33 and second 34 mixers, first 35 and second 36 local oscillators, first 37 and second 38 amplifiers of intermediate frequency, a frequency search unit 39, a first multiplier 40, a first narrow-band filter 41, a detector 42, doubler 43 frequencies, the first 44 and second 45 meters of the signal spectrum width, the first comparison unit 46, the first threshold unit 47, the first delay line 48, the first key 49, the second key 50, the first 51 and second 52 multiplexers, the subtraction unit 53, the adder 54 module two, driver 55, multi-channel correlator 56, multi-tap delay line 57i, multi-channel multiplier 58i, multi-channel low-pass filter 59i, second threshold block 60, multi-channel comparison unit 61 _i , clock generator 62, shift register 63 _i (i = 1, 2, ..., n), AND element 64, delay element 65 , counter 66, first storage register 67, third key 68, digital comparator 69, second storage register 70, fourth 71 and fifth 72 keys, second multiplier 73, second delay line 74, first phase detector 75, second narrow-band filter 76, second phase detector 77, voltage-code converter 78, frequency standard 79, time code generator 80, antenna control unit 81, rotary device 82, antenna system 83, receiver 84, phase demodulator 85, analog recording device 86, analog-to-digital converter 87, block 88 synchronization tion, the first 89 and second 90 signal processing apparatus and a device 91 for interfacing with communication lines.

An on-board memory device 8 and an on-board transmitter 11 are connected to the output of the on-board receiver 6, the second input of which is connected to the output of the receiver 6, the third input is connected to the output of the on-board receiver 5. An on-board memory 10 is connected to the output of the on-board receiver 7 which is connected to the fourth input of the transmitter 11, the fifth input of which is connected to the output of the receiving device 7. To the output of the receiving device 13 are connected in series to the processing device 15 information, a device 17 for interfacing with a communication system, a control device 18 and a communication device 19 for search and rescue organizations. In the receiving device 7, a mixer 33 is connected in series to the antenna A, the second input of which is connected through the local oscillator 35 to the output of the frequency signal search unit 39, the intermediate frequency amplifier 37, the multiplexer 51, the second input of which is connected to the output of the intermediate frequency amplifier 38, multi-tap line 57 _i delay, multiplier 58 _i , low-pass filter 59 _i , comparison unit 61 _i , shift register 63i, the second input of which is connected to the output of the clock generator 62, element And 64, the second input of which is connected to the output of the generator and 62 clock pulses, a counter 66, the second input of which through the delay element 65 is connected to the output of the shift register 63 _i , the storage register 67, the second input of which is connected to the output of the shift register 63 _i , the key 68, the second input of which is connected to the output of the threshold block 60 , a digital comparator 69, the second input of which is connected to the output of the storage register 70, and the storage register 70, the second input of which is connected to the output of the key 68, and the output is the second output of the receiving device 7. A mixer 34 is connected in series to the output of the antenna B, the second input which through the local oscillator 36 is connected to the output of the search unit 39, an intermediate frequency amplifier 38, a multiplexer 52, the second input of which is connected to the output of the intermediate frequency amplifier 37, and a key 50, the second input of which is connected to the output of the threshold unit 47, and the output is connected to the second input multiplier 58 _i . To the output of the intermediate frequency amplifier 37, a frequency doubler 43, a signal spectrum width meter 45, a comparison unit 46 are connected in series, the second input of which is connected to the output of the intermediate frequency amplifier 37 through a signal spectrum meter 44, a threshold unit 47, the second input of which is connected to the line output 48 delay, key 49, the second input of which is connected to the output of the intermediate frequency amplifier 37, key 72, the second input of which is connected to the output of the threshold unit 60, delay line 74 and phase detector 75, the second input of which It is connected to the output of the key 72, and the output is the third output of the receiving device 7. To the output of the intermediate frequency amplifier 38, a key 71 is connected in series, the second input of which is connected to the output of the threshold unit 60, the multiplier 73, the second input of which is connected to the output of the intermediate frequency amplifier 37 , a narrow-band filter 76, a phase detector 77, the second input of which through series-connected multiplier 40 and a narrow-band filter 41 is connected to the second outputs of the local oscillators 35 and 36, and the output is the fourth output many devices 7. A subtraction unit 53 is connected in series to the output of the intermediate frequency amplifier 37, the second input of which is connected to the output of the intermediate frequency amplifier 38, and the adder 54 is modulo two, the second input of which is connected to the output of the first channel 61 ₁ of the comparison unit 61 _i , and the output is connected to the third inputs of the multiplexers 51 and 52. A shaper 55 is connected to the output of the adder 54 modulo two, the output of which is the first output of the receiving device 7. A code generator 80 is connected in series to the output of the frequency standard 79 time, antenna control unit 81, rotary device 82, antenna system 83, receiver 84, the second input of which is connected to the output of the frequency standard 79, a phase demodulator 85, an analog recording device 86, an analog-to-digital converter 87, the second input of which is connected to the first the output of the phase demodulator 85, the signal processing device 89, the second input of which is connected to the second output of the antenna control unit 81, and the interface device 91. A synchronization unit 88 is connected to the second output of the second phase demodulator 85, the second input of which is connected to the second output of the analog recording device 86, and a signal processing device 90, the output of which is connected to the second input of the communication device 91.

 The system operates as follows. Currently, there are three types of ARBs: aviation, marine, and portable (for use on land) that emit signals detected and received by satellites of the COSPAS-SARSAT system for the purpose of subsequent relay to coast earth stations (PPI) for processing and determining the location of beacons. The service area of the COSPAR-SARSAT system in real time is determined by the number and geographical location of the PPI. Each PPI serves an area with a radius of approximately 2,500 km. The COSPAS-SARSAT system includes 15 PPS deployed in seven countries. In Russia, PPI are located in Moscow, Arkhangelsk Vladivostok and Novosibirsk.

 Distress messages and the coordinates of the emergency facility are transmitted through the system control center (CCC) either to the national rescue coordination center, or to another CCC or to the corresponding search and rescue service in order to deploy a search and rescue operation.

 The coordinates of the ARB are determined on the basis of measurement using the satellite Doppler frequency offset received from the ARB signal. The carrier frequency of the ARB transmitter is quite stable during the time of mutual radio visibility of the ARB-AES. The COSPAS-SARSAT system currently uses ARB 1 operating at a frequency of 121.5 MHz - the international aviation emergency frequency - and in the frequency range 406-406.1 MHz. ARB 2 operating in the range 406-406.1 MHz are technically more complex than ARB 1 due to the inclusion of an identification code and other information in the message corresponding to the search and rescue operation. The use of low-altitude near-polar satellites in the system makes it possible to optimize the use of the Doppler effect, reduce the requirements for ARB radiation power, obtain relatively short time intervals between successive satellites passes over observation areas, and ensure global sequential coverage of the Earth.

 The solution to the problem of determining the coordinates of the ARB for one passage of the satellite from Doppler measurements gives two pairs of coordinates on both sides of the satellite path - the true and false (mirror) coordinates of the ARB. The elimination of this ambiguity is solved by mathematical methods, which are based on the fact that the symmetry of Doppler readings is violated due to the rotation of the Earth. With a sufficiently high stability of the radiation frequency of the ARB, which is observed in the case of using ARB 2, the true coordinates of the ARB are determined in one passage of the satellite. When receiving signals from ARB 1, ambiguity is resolved during the second passage of the satellite, if this cannot be done during the first passage.

 The system (nominal configuration) includes four satellites, two of which are presented and supported by the COSPAS side and two by the SARSAT side.

 In the COSPAS - SARSAT system, two modes of operation are used to detect ARB signals and determine their location: a mode for receiving and transmitting information in real time and a mode for receiving and storing information on board an artificial satellite and its subsequent transmission to the information receiving point when the satellite is in the radio visibility zone PPI. ARB 1 can only be used in direct transmission mode, while ARB 2 can be used in both operating modes.

 An onboard satellite repeater at a frequency of 121.5 MHz provides the relay of ARB 1 signals directly to the PPI. If at the moment of receiving the signal on the satellite the PPI is also in its visibility, the ARB signal can be received and processed by the equipment of the ground-based PPI complex.

 After receiving signals from the ARB 2 to the satellite, the on-board processor measures the Doppler frequency of the signal, as well as processes and sortes the digital information in the ARB message. During processing, the ARB message is attached to timestamps, converted to digital form and transmitted in real time to any PPI located in the satellite visibility zone. At the same time, this information is recorded in the storage device 8 for subsequent transmission to the PPI 12, when the latter is in the visibility range of the satellite. This mode ensures the receipt of an alarm message by all PPI systems that are in operation.

 An important feature of the new generation of ARBs is the inclusion in its radiation of a digital message that carries information on the membership of the ARB (country), the identification number of the vessel or aircraft and the type of distress. The ARB message installed on ships may also include information on the location of the ship, entered manually or automatically from ship's radio navigation devices. The structure of ARB 2 may also include a transmitter emitting signals to drive search and rescue equipment on the ARB. Information about the type of drive radio equipment used is also included in the alarm message. The inclusion of the ARB can be done manually or automatically, depending on its modification (marine, aviation or portable).

 The radio transmitter 3 of the vehicle operates as follows. The vehicle itself can be in two modes: in normal operation, when the radio is off, and in security mode, when the radio is on. In the first mode, the vehicle is transferred by bringing the permanent magnet, made, for example, in the form of a keychain, to the reed switch 26 installed behind the skin of the vehicle in a place known only to the owner. In this case, the winding 24 of the remote switch 23 through the closed contacts 24.1 and the reed switch 26 is connected to a power source 20. The remote switch 23 is transferred to the first stable state, in which the contacts 24.2 are closed and the contacts 24.1 are opened. Contacts 25.1 and 25.2 are in open state. When the ignition is turned on, the supply voltage is supplied to the ignition coil and the engine is operating in normal mode, there is no malfunction in the ignition circuit.

 To put the vehicle into guard mode, i.e. to turn on the anti-theft device, the owner again brings the permanent magnet to the reed switch 26. In this case, the winding 25 is activated and the remote switch 23 is transferred to the second stable state, in which contacts 24.1, 25.1 and 25.2 are closed and contacts 24.2 are opened. In this case, the supply voltage through the resistor 21 and the closed contacts 25.1 is supplied to the LED indicator 22, which is activated and signals that the anti-theft device is turned on.

 When the ignition is switched on via closed contacts 25.2, the vehicle body is connected to a modulating code generator 28, a master oscillator 30, a phase manipulator 31, and a transmitter 32. The generator 28 starts generating a modulating code M (t) (Fig. 7a), periodically opening and closing contacts 29.1 relay 29. In this case, the engine is started during the closed state of contacts 29.1, but theft is not possible, because after a while the generator gives a positive impulse, contacts 29.1 open, the ignition system and the engine tklyuchayutsya.

 A person trying to hijack begins to consistently look for the cause of engine failure. At the same time, it is assumed that most of the faults occur in the ignition system. Usually they start checking the ignition system, since it is most simple to verify its serviceability (by the presence of a spark on high-voltage wires suitable for candles).

 Suppose a person trying to hijack has brought a high-voltage wire to the ground and cranks the engine. If there is a spark (the period when the generator 28 gives a negative impulse), then the hijacker switches to troubleshooting the power supply system and begins to sequentially check the power supply sections, i.e., it moves away from the correct search path. If during the test there is no spark (the period when the generator 28 gives a positive impulse), then the hijacker examines the electrical circuit and searches for the damaged area before the generator 28 gives a negative impulse and the fault disappears. This serves as a pointer to replace the allegedly faulty section of the circuit, i.e. misleading again. Troubleshooting is getting complicated.

 Consequently, the absence of an audible alarm does not cause concern and allows an attacker to engage in their criminal activities for a long time. At the same time, the hijacker, having made repeated attempts to start the engine, still has a real opportunity to detect the presence of an anti-theft device, to reveal the principle of its operation and to hijack a vehicle. To prevent theft of the vehicle, a radio channel is used, through which alarm information is transmitted through the satellite to the reception center, where measures are taken to detain the hijacker. When the contacts 25.2 are closed, the supply voltage is supplied to the master oscillator 30, the phase manipulator 31 and the transmitter 32 through the closed ignition key 27.

Harmonic tension
U _c (t) = U _c ˙cos (2πf _c t + φ _c ), where U _c , f _c and φ _c are the amplitude, carrier frequency and initial phase of the voltage, from the output of the master oscillator 30 goes to the first input of the phase manipulator 31 , to the second input of which a modulating code M (t) is supplied (Fig. 7a) from the output of the generator 28. At the output of the phase manipulator 31, a phase-shifted signal is generated
U _c (t) = U _c ˙cos [2πf _c t + φ _k (t) + φ _c ], 0≅ t ≅ T _s , where φ _k (t) = {0, π} is the manipulating component of the phase, which displays the law of phase manipulation in accordance with the modulating code M (t), and φ _k (t) = const at kτ _p <t <(k + 1) τ _p and can change stepwise at t = k τ _p , i.e. at the borders between elementary premises (K = 0,1,2, ... N-1);
τ _p and N is the duration and number of chips that make up a signal of duration T _s (T _c = τ _n N), which, after amplification in the transmitter 32, is emitted by the antenna.

 The onboard radio satellite complex of the COSPAR-SARSAT system operates in real-time relaying information about ARB 1, relaying information about ARB 2 in real time with preprocessing on board a satellite, storing information about ARB 2 for the purpose of transmitting PPI, relaying information about the vehicle’s radio sensor 3 in real time with preliminary processing on board the satellite, storing information about the vehicle’s radio sensor 3 with a view to subsequent transmission to the PPI.

- ширина спектра ФМн-сигнала радиодатчика;
Δ f₃ - ширина защитного частотного интервала;
l - число радиодатчиков, подлежащих контролю;
τ_п - длительность элементарной посылки ФМн-сигнала.The on-board complex 4 consists of the following main elements: a receiving device 5 operating at a frequency of 121.5 MHz, a device 6 for receiving and processing information about the ARB 2, a unit 7 for receiving and processing information about the radio sensor of the vehicle, on-board storage devices 8 and 10, on-board a storage device 9, a transmitting device 11 at a frequency of 1544.5 MHz. The receiving device 5 at a frequency of 121.5 MHz has a bandwidth of 25 kHz. A constant output level is provided by the automatic gain control (AGC) device. Block 6 receiving and processing information from ARB 2 performs the following functions: demodulating digital messages received from ARB, measuring the received frequency, linking time stamps to the measurements. Block 7 of the reception and processing of information from the radio sensor 3 of the vehicle (Fig.6) performs the following functions: detection and selection of phase-shift (QPSK) signals in a given frequency range, suppression of additional (mirror and combination) reception channels, autocorrelation detection of QPSK signals, accurate and unambiguous direction finding of the vehicle radio sensor 3, linking the results of the measurements to time stamps. The width of the search range D _{f the} signals of the radio sensors is selected from the condition of ensuring the frequency selection of signals from an individual sensor with the required quality according to the expression
D _f = Δ f _c l + Δ f ₃ (l-1), where Δf _c =

- the spectrum width of the FMN signal of the radio sensor;
Δ f ₃ - the width of the protective frequency interval;
l is the number of radio sensors to be monitored;
τ _p - the duration of the elementary sending FMN signal.

-f

= 2f_пр на выборе частот f_г1иf_г2 cимметричными относительно несущей частоты f_cпринимаемого сигнала
f_c-f

= f

-f_c= f_пр и на корреляционной обработке канальных ФМн-сигналов. Отмеченные условия приводят к удвоению числа дополнительных каналов приема (фиг.3).Suppression of additional (mirror and combination) reception channels is based on the use of two local oscillators 35 and 36, the frequencies of which are tuned synchronously and spaced by twice the intermediate frequency
f

-f

= 2f _etc. on selection frequency f _r1 and f _r2 Symmetric about the carrier frequency f _c of the received signal
f _c -f

= f

-f _c = f _CR and the correlation processing of channel PSK signals. The marked conditions lead to a doubling of the number of additional receiving channels (figure 3).

sinβ где d - измерительная база (расстояние между антеннами А и В);
λ - длина волны;
β- угол прихода радиоволны относительно нормали к плоскости установки антенн.The elimination of the ambiguity in reading the angular coordinate β inherent in the phase direction finding method is based on the correlation processing of channel PSK signals. In this case, the phase difference of the high-frequency oscillations received by antennas A and B (Fig.6) is determined by the ratio
Δφ = 2π

sinβ where d is the measuring base (distance between antennas A and B);
λ is the wavelength;
β is the angle of arrival of the radio wave relative to the normal to the plane of installation of the antennas.

 The phase direction finding method is characterized by a contradiction between the requirements for measurement accuracy and the uniqueness of the angle β reading. Indeed, according to the above expression, the phase system is the more sensitive to a change in angle, the larger the relative size of the base d / λ. However, with increasing d / λ, the value of the angular coordinate β decreases, at which the phase difference Δφ exceeds 2π, i.e. ambiguity of counting occurs.

- время запаздывания сигнала, приходящего на одну из антенн, по отношению к сигналу, приходящему на другую антенну;
ΔR - разность расстояний от радиодатчика транспортного средства до антенн А и В;
с - скорость распространения радиоволн.On the other hand, the indicated phase difference is determined as follows:
Δφ = 2πf _c (t + τ) -2πf _c t = 2πf _c τ, where τ =

- the delay time of the signal arriving at one of the antennas with respect to the signal arriving at the other antenna;
ΔR is the difference between the distances from the vehicle radio sensor to antennas A and B;
C is the propagation velocity of radio waves.

sinβ = 2πf_c

sinβ
τ =

sinβ
Таким образом, измерив величину задержки τ и зная измерительную базу d, можно однозначно определить значение истинного пеленга
sinβ =

τ
Минимальное (нулевое) значение τ (τ_мин=0) cоответствует значению β = 0. Максимальное значение τ (τ_макс) соответствует углу β= 90^o;
τ_макс=

sinβ =

sin90^°=

Следовательно, sinβ =

Измерив τ с помощью корреляционной обработки принимаемых ФМн-сигналов, можно определить истинный пеленг β. При этом устраняются зависимость результатов измерения от несущей частоты f_c принимаемых ФМн-сигналов и неоднозначность измерения, присущая фазовому методу пеленгации. Предлагаемая система обеспечивает измерение τ, используя известное свойство корреляционной функции ФМн-сигналов, несущей "кнопкообразную" форму с максимумом в области нулевых задержек.Therefore, equating these ratios, get
2πf _c τ = 2π

sinβ = 2πf _c

sinβ
τ =

sinβ
Thus, by measuring the delay value τ and knowing the measuring base d, we can uniquely determine the value of the true bearing
sinβ =

τ
The minimum (zero) value of τ (τ _min = 0) corresponds to the value β = 0. The maximum value of τ (τ _max ) corresponds to the angle β = 90 ^o ;
τ _max =

sinβ =

sin90 ^° =

Therefore, sinβ =

By measuring τ using the correlation processing of the received PSK signals, one can determine the true bearing β. In this case, the dependence of the measurement results on the carrier frequency f _{c of the} received PSK signals and the ambiguity of the measurement inherent in the phase direction finding method are eliminated. The proposed system provides a measurement of τ using the well-known property of the correlation function of the PSK signals, bearing a "button-like" shape with a maximum in the region of zero delays.

- время запаздывания сигнала, приходящего на антенну В, по отношению к сигналу, приходящему на антенну А (фиг.4а);
t₁ и t₂ - время прохождения сигналом расстояний от радиодатчика транспортного средства до антенн А и В;
Δφ - разность фаз сигналов, определяющая направление на источник излучения.Viewing predetermined frequency band D _f and search PSK-signals are performed by using the search unit 39 which periodically with a period T _n sawtooth and synchronously change the frequency f _r1 and f _r2 oscillators 35 and 36. The generator may be used as a search block of the ramp 39 voltage. Keys 49, 50, 68, 71 and 72 in the initial state are closed. The first inputs of the mixers 33 and 34 from the outputs of the antennas A and B (Fig.6) receive FMN signals
U ₁ (t) = U _c ˙cos [2πf _c t + φ _к (t) + φ _c], 0 ≅ t ≅ T _s ;
U ₂ (t) = U _c ˙cos [2πf _c (t) + φ _k (t) +
+ Δφ + φ _c ] = U _c ˙cos [2πf _c (t + τ) +
+ φ _k (t + τ) + φ _c ], 0≅t≅T _c ; where τ = t ₁ -t ₂ =

- the delay time of the signal arriving at the antenna B, in relation to the signal arriving at the antenna A (figa);
t ₁ and t ₂ - the time the signal travels distances from the vehicle’s radio sensor to antennas A and B;
Δφ is the phase difference of the signals, which determines the direction to the radiation source.

(t) = U

cos(2πf

t + πγt²+

) , 0≅ t≅ T_п
U

(t) = U

cos(2πf

t + πγt²+

) , 0≅ t≅ T_п где U_г1, U_г2, f_г1, f_г2, φ_г1, φ_г2 и Т_п - амплитуды, начальные частоты, начальные фазы и период перестройки напряжений гетеродинов;
γ =

- скорость изменения частот гетеродинов (скорость перестройки).The following voltages are applied to the second inputs of the mixers 33 and 34 from the outputs of the local oscillators 35 and 36, respectively
U

(t) = U

cos (2πf

t + πγt ² +

), 0≅ t≅ T _p
U

(t) = U

cos (2πf

t + πγt ² +

), 0≅ t≅ T _p where U _g1 , U _g2 , f _g1 , f _g2 , φ _g1 , φ _g2 and T _p are the amplitudes, initial frequencies, initial phases and the period of tuning of the local oscillator voltages;
γ =

- rate of change of local oscillator frequencies (tuning rate).

(t) = U

cos[2πf_прt + φ_к(t) - πγt²+

], 0≅ t≅ T_с, T_c< T_п
U

(t) = U

cos[2πf_прt - φ_к(t) + πγt²+

- Δφ] =
= U

cos[2πf_пр(t+τ)-φ_к(t+τ) + πγ(t+τ)²+

], 0≅ t≅ T_c, T_c< T_п где U

=

K₁U_cU

U

=

K₁U_cU

K₁ - коэффициент передачи смесителей;
f_пр= f_c-f

= f

-f_c - промежуточная частота;

= φ_c-

,

=

-φ_c
Напряжение U

(t) с выхода усилителя 37 промежуточной частоты поступает на вход обнаружителя 42, состоящего из удвоителя 43 частоты, первого 44 и второго 45 измерителей ширины спектра, блока 46 сравнения, порогового блока 47 и линии 48 задержки. На выходе удвоителя 43 частоты образуется напряжение
U₃(t) = U

cos(4πf_прt-2πγt²+2

), 0≅ t≅ T_c
Так как 2 φ_k(t) = {0, 2 π}, то в указанном напряжении манипуляция фазы уже отсутствует.At the outputs of the mixers 33 and 34, voltages of combination frequencies are generated. Amplifiers 37 and 38 of the intermediate frequency are allocated only the voltage of the intermediate (difference) frequency:
U

(t) = U

cos [2πf _pr t + φ _k (t) - πγt ² +

], 0≅ t≅ T _s , T _c <T _p
U

(t) = U

cos [2πf _pr t - φ _k (t) + πγt ² +

- Δφ] =
= U

cos [2πf _pr (t + τ) -φ _to (t + τ) + πγ (t + τ) ² +

], 0≅ t≅ T _c , T _c <T _p where U

=

K ₁ U _c U

U

=

K ₁ U _c U

K ₁ - gear ratio of the mixers;
f _ol = f _c -f

= f

-f _c is the intermediate frequency;

= φ _c -

,

=

-φ _c
Voltage U

(t) from the output of the intermediate frequency amplifier 37, is fed to the input of the detector 42, consisting of a frequency doubler 43, a first 44 and a second 45 spectrum width meters, a comparison unit 46, a threshold unit 47 and a delay line 48. A voltage is generated at the output of the frequency doubler 43
U ₃ (t) = U

cos (4πf _pr t-2πγt ² +2

), 0≅ t≅ T _c
Since 2 φ _k (t) = {0, 2 π}, phase manipulation is already absent in the indicated voltage.

The width of the spectrum Δ f _{2 of the} second harmonic of the signal is determined by the signal duration T _s (Δf ₂ = 1 / T _s ), while the spectrum width of the QPSK signal is determined by the duration τ _{p of} its elementary premises (Δ f _c = 1 / τ _p ), t. e. the spectrum width of the second harmonic of the signal is N times smaller than the spectrum width of the input signal:
Δ f _c / Δ f ₂ = N.

Therefore, when the frequency of the QPSK signal is doubled, its spectrum collapses N times. This circumstance makes it possible to detect the QPSK signal by filtering in a narrow frequency band even when its power at the system input is less than the noise power. The width of the spectrum Δ f _{c of the} input signal is measured using a meter 44, and the width of the spectrum Δ f _{2 of the} second harmonic is measured using a meter 45. The voltages U ₁ and U ₂ proportional to Δ f _c and Δ f ₂ from the outputs of the meters 44 and 45 the width of the spectrum of the signals is fed to two inputs of block 46 comparison. Since U ₁ >> U ₂ , a positive pulse is generated at the output of the comparison unit 46, which exceeds the threshold level U _pore1 in the threshold unit 47, which is selected so that it is not exceeded by random noise. The indicated level is exceeded only when the QPSK signal is detected. When the threshold voltage U _pore1 is exceeded, a constant voltage is generated in the threshold block 47, which is fed to the input of the delay line 48, to the control input of the key 50, opening it, and to the control input of the search unit 39, putting it into stop mode. From this point in time, viewing the specified frequency range D _f and searching for PSK signals stop for the processing time of the detected signal, which is determined by the delay time τ _{3 of} the delay line 48.

(t) = U

cos[2πf_прt+φ_к(t)+

], 0≅ t≅ T_c
U

(t) = U

cos[2πf_прt-φ_к(t)+

-Δφ] =
= U

cos[2πf_пр(t+τ)-φ_к(t+τ)+

] , 0≅ t≅ T_c которые через мультиплексоры 51, 52 и открытый ключ 50 поступают на два входа коррелятора 56, состоящего из многоотводной линии 57_i задержки, перемножителя 58_i и фильтра 59_i нижних частот (i = 1,2,...,n). На выходе перемножителя 58i образуются напряжения суммарной и разностной частот. На выходе i-го элемента перемножителя 58i образуется напряжение, которое имеет максимальное значение при условии τ_i= τ_o, где τ_i - время задержки i-го элемента многоотводной линии 57_i задержки. Фильтр 59_iнижних частот выделяет пропорциональные корреляционной функции R( τ) напряжения разностной частоты. Причем эти напряжения максимальны только при задержке τ_i = τ_o, для которой β=β_o, где β_o - истинный пеленг, и при приеме ФМн-сигналов по основному каналу на частоте f_с.When the tuning of the local oscillators 35 and 36 ceases, the following voltages are allocated by the amplifiers 37 and 38 of the intermediate frequency:
U

(t) = U

cos [2πf _pr t + φ _k (t) +

], 0≅ t≅ T _c
U

(t) = U

cos [2πf _pr t-φ _k (t) +

-Δφ] =
= U

cos [2πf _pr (t + τ) -φ _to (t + τ) +

], 0≅ t≅ T _c which through the multiplexers 51, 52 and the public key 50 go to the two inputs of the correlator 56, consisting of a multi-tap delay line 57 _i , a multiplier 58 _i and a low-pass filter 59 _i (i = 1,2 ,. .., n). At the output of the multiplier 58i, voltages of the sum and difference frequencies are generated. At the output of the i-th element of the multiplier 58i, a voltage is generated that has a maximum value under the condition τ _i = τ _o , where τ _i is the delay time of the i-th element of the multi-tap delay line 57 _i . The low-pass filter 59 _i extracts the voltage of the difference frequency proportional to the correlation function R (τ). Moreover, these voltages are maximum only with a delay of τ _i = τ _o , for which β = β _o , where β _o is the true bearing, and when receiving PSK signals through the main channel at a frequency f _s .

From the outputs of the correlator 56, the voltages are supplied to the inputs of the threshold block 60, where they are compared with the threshold voltage U _pore2 . Moreover, the threshold level U _pore2 in the threshold block 60 is exceeded only at the maximum value of the correlation function R (τ _o ), it is not exceeded at values of τ corresponding to the side lobes of the correlation function R (τ). When the threshold voltage U _pore2 is exceeded, a constant voltage is generated in the threshold block 60, which is supplied to the control inputs of the keys 68, 71, and 72 and opens them.

From the outputs of the correlator 56, the voltages simultaneously arrive at the inputs of the multi-channel comparison unit 61i (i = 1,2, ..., n), which is n analog comparators. Each comparator is an analog comparison element, which compares two voltages - U input _Rin and the reference U _op. If the input voltage exceeds the reference voltage (U _I > U _op ), a voltage corresponding to a logical “1” is generated at the output of the comparator 61i. It should be noted that the voltages from the outputs of the correlator 56 are supplied to the comparators 61 _i (i = 1,2, ..., n) so that the same voltage is applied to two neighboring comparators, and one of the comparators as an input voltage voltage U _I , and on the other - reference U _op . Thus, a parallel binary code is formed at the outputs of the comparators, in which "1" corresponds to the excess voltage in the (i + 1) -th channel of the correlator 56 over the voltage in the i-th channel. The sequence of units of the binary code corresponds to an increase in the correlation function R (τ), and the sequence of zeros corresponds to the decline of the correlation function R (τ). Therefore, the last unit in a block of several units of the unit code corresponds to the peak of the correlation function R (τ _o ). By counting the number of units of the binary code (L), one can determine the number of channel i, in which τ _i = τ _o , and therefore the value of τ _o .

The parallel binary code from the outputs of the comparators 61 _i goes to the shift register 63i, where it is converted to a serial binary code. The shift of the parallel binary code in the register 63i is carried out by applying to its control input (synchronization input) clock pulses from the generator 62 clock pulses. The counting pulses are generated using the And 64 element, one of the outputs of which receive clock pulses from the output of the generator 62, and the other is a serial binary code from the output of the shift register 63 _i . A sequence of counting pulses, the number L of which is equal to the number "1" of the binary code, is fed to the input of the counter 66, where the number "1" is counted. The count is terminated at the end of the sequence of the block of units of the binary code from the output of the shift register 63 _i , i.e. upon transition from the logical level "1" to the logical level "0". At the end of the count, its result must be recorded in the storage register 67, and then transfer the counter 66 to the zero state. Writing to the storage register 67 is carried out simultaneously with the end of the count by the control signal from the output of the shift register 63 _i . To translate the counter 66 to the zero state, it is after writing the count result to the storage register 67 that the control signal from the output of the shift register 63 _i is delayed by the delay element 65 and is fed to the reset input of the counter 66.

The value of the binary code recorded in the storage register 67, through the public key 68 is supplied to the first input of the digital comparator 69, where it is compared with the value of the binary code available in the storage register 70 and fed to the second input of the digital comparator 69. This is done to avoid re-recording one and the same binary code value corresponding to the same true bearing β _o .

If the binary codes being compared are not equal to each other, a control signal corresponding to the logic level “1” is generated in the digital comparator 69, which is fed to the control input of the storage register 70, allowing the writing of a new binary code value. If the binary codes being compared are equal, then there is no rewriting to the storage register 70. Therefore, at the output of the storage register 70, a binary code is generated equal to the number of units L in the serial binary code coming from the output of the shift register 63i. The specified code corresponds to τ _i = τ _o , i.e. the delay value at which the correlation function R (τ _o ) has a maximum value
τ _o = Lτ ₁ , where τ ₁ is the delay value of one element of the multi-tap delay line 57 _i .

 The solution to the problem of determining the coordinates of the vehicle’s radio sensor 3 in one pass of the satellite by the phase method gives two pairs of coordinates on both sides of the satellite path — the true and false (mirror) coordinates of the radio sensor.

(t) ≠ U

(t)(фиг. 4 а,в). Если U

(t) ≈ U

(t) (фиг. 4б), то на выходе блока 53 вычитания формируется логический "0". Если радиодатчик 3 транспортного средства находится в правой полуплоскости от проекции на поверхность Земли трассы прохождения спутника (фиг. 4а), то на выходе первого канала 61₁ блока 61_i сравнения формируется логическая "1", потому что при сравнении напряжений первого и второго каналов коррелятора 56 напряжение первого канала имеет большую задержку, чем напряжение второго канала, т.е. ближе расположено к максимальному значению корреляционной функции R(τ_o) и, следовательно, имеет большую величину. При этом выход усилителя 37 промежуточной частоты оказывается подключенным через открытый ключ 50 непосредственно к перемножителю 58_i, а выход усилителя 38 промежуточной частоты - к многоотводной линии 57_i задержки. При этом на выходе сумматора 54 по модулю два формируется логический "0".To eliminate this ambiguity and the correct operation of the multi-channel correlator 56, a subtraction block 53, an adder 54 modulo two, multiplexers 51 and 52 are used. A logical “1” is formed at the output of the subtraction block 53, when U

(t) ≠ U

(t) (Fig. 4 a, c). If U

(t) ≈ U

(t) (Fig. 4b), a logical “0” is generated at the output of the subtraction block 53. If the vehicle’s radio sensor 3 is located in the right half-plane from the projection of the satellite path on the Earth’s surface (Fig. 4a), then a logical “1” is formed at the output of the first channel 61 ₁ of the comparison unit 61 _i , because when comparing the voltages of the first and second channels of the correlator 56, the voltage of the first channel has a greater delay than the voltage of the second channel, i.e. closer to the maximum value of the correlation function R (τ _o ) and, therefore, has a large value. In this case, the output of the intermediate frequency amplifier 37 is connected via the public key 50 directly to the multiplier 58 _i , and the output of the intermediate frequency amplifier 38 is connected to the multi-tap delay line 57 _i . At the same time, at the output of the adder 54 modulo two, a logical "0" is formed.

If the vehicle’s radio sensor 3 is in the left half-plane (Fig. 4c), then the voltage of the second channel of the correlator 56 is greater than the voltage of the first channel and a logical “0” is generated at the output of the first channel 61 ₁ of the comparison unit 61 _i . In this case, at the output of the adder 54 modulo two, a control signal is generated corresponding to the logic level “1”. The multiplexers 51 and 52, under the influence of a control signal corresponding to a logical “1”, switch the receiving channels, in which the intermediate frequency amplifier 37 is connected to the multi-tap delay line 57i and the intermediate frequency amplifier 38 is connected via the public key 50 to the multiplier 58. The control signal, corresponding to the logical level “1”, is fed to the input of the shaper 55, where a digital code is generated, which is a sign that the vehicle’s radio sensor 3 is located to the left of the satellite path.

 If the vehicle’s radio transmitter 3 is in the same direction, i.e. on the satellite path (Fig. 4b), then switching of the receiving channels does not occur, switching of the receiving channels is carried out according to the truth table (Fig. 5).

(t) и U

(t) c вторых выходов гетеродинов 35 и 36 поступают на два входа перемножителя 40, на выходе которого образуется напряжение
U_г(t) = U_г˙cos(4πf_прt+φ_пр), где U_г =

K₂U

U

К₂ - коэффициент передачи перемножителя;
f

-f

=2f_пр; φ_пр=

-

Это напряжение выделяется узкополосным фильтром 41 и поступает на первый вход фазового детектора 77.Stress U

(t) and U

(t) from the second outputs of the local oscillators 35 and 36 are fed to two inputs of the multiplier 40, at the output of which a voltage is generated
U _g (t) = U _g ˙cos (4πf _pr t + φ _pr), where U _g =

K ₂ U

U

K ₂ is the transmission coefficient of the multiplier;
f

-f

= 2f _straight; φ _CR =

-

This voltage is allocated by the narrow-band filter 41 and is supplied to the first input of the phase detector 77.

K₂U

U

которое выделяется узкополосным фильтром 76 и поступает на второй вход фазового детектора 77. На выходе последнего образуется постоянное напряжение
U

(β) = U

cosΔφ где U

=

K₃U_гU_пр
К₃ - коэффициент передачи фазового детектора; Δφ = 2π

sinβ-пропорциональное фазовому сдвигу Δφ определяющему направление на радиодатчик 3 транспортного средства. Это напряжение поступает на вход преобразователя 78 напряжение -код, где преобразуется в цифровой код.The voltage U _CR3 (t) from the output of the intermediate frequency amplifier 37 is supplied to the first input of the multiplier 73. The voltage U _CR4 (t) from the output of the intermediate frequency amplifier 38 is supplied through the public key 71 to the second input of the multiplier 73, at the output of which a voltage is generated
U _pr (t) = U _pr ˙cos (4πf _pr t + φ _pr -Δφ), where U _pr =

K ₂ U

U

which is allocated by a narrow-band filter 76 and enters the second input of the phase detector 77. At the output of the latter, a constant voltage is generated
U

(β) = U

cosΔφ where U

=

K ₃ U _g U _ol
K ₃ is the transfer coefficient of the phase detector; Δφ = 2π

sinβ-proportional to the phase shift Δφ determining the direction to the radio sensor 3 of the vehicle. This voltage is fed to the input of the voltage-converter 78, where it is converted into a digital code.

(t) (фиг. 7б) с выхода усилителя 37 промежуточной частоты через открытые ключи 49 и 72 поступает на первый вход фазового детектора 75 и на вход линии 74 задержки, на выходе которой образуется напряжение (фиг.7в).Voltage U

(t) (Fig. 7b) from the output of the intermediate frequency amplifier 37 through the public keys 49 and 72 is fed to the first input of the phase detector 75 and to the input of the delay line 74, at the output of which a voltage is generated (Fig. 7c).

(t) = U

cos[2πf_пр(t-τ_п)+φ_к(t-τ_п)+φ_пр]
Это напряжение поступает на второй вход фазового детектора 75, на выходе которого образуется низкочастотное напряжение (фиг.7г)
U

(t) = U

cosφ_к(t) где U

=

K₃U

, пропорциональное модулирующему коду М(t) (фиг.7а).U

(t) = U

cos [2πf _pr (t-τ _p ) + φ _k (t-τ _p ) + φ _pr ]
This voltage is supplied to the second input of the phase detector 75, at the output of which a low-frequency voltage is generated (Fig.7g)
U

(t) = U

cosφ _to (t) where U

=

K ₃ U

proportional to the modulating code M (t) (figa).

The delay time τ _{3 of} the delay line 48 is selected so that it is possible to process the detected PSK signal. After this time, the voltage from the output of the delay line 48 is supplied to the reset input of the threshold unit 47 and resets it to its initial (zero) state. In this case, the search unit 39 is transferred to the tuning mode, and the key 50 is closed, i.e. is reset. From this point in time, viewing a given frequency range D _f and searching for PSK signals continue. If the next QPSK signal is detected at a different carrier frequency emitted by the radio sensor of another vehicle, the system operates similarly to that described.

The operation of the system described above corresponds to the case of receiving FMK signals on the main channel at a frequency f _c (Fig. 3).

-f₃₁= f_пр
f₁₂= f

-f₃₁= 3f_пр где первый индекс обозначает канал, по которому принимается ложный сигнал (помеха);
второй индекс обозначает номер гетеродина, частота которого участвует в преобразовании несущей частоты принимаемого сигнала.If a false signal (interference) is received through the first mirror channel at a frequency of f ₃₁ , then in mixers 33 and 34 it is converted to the voltage of the following frequencies:
f ₁₁ = f

-f ₃₁ = f _pr
f ₁₂ = f

-f ₃₁ = 3f _pr where the first index denotes the channel through which a false signal (interference) is received;
the second index denotes the number of the local oscillator, the frequency of which is involved in the conversion of the carrier frequency of the received signal.

However, only voltage with a frequency f ₁₁ falls into the passband Δ f _{p of the} intermediate frequency amplifier 37 and to the first input of the correlator 56. The output voltage of the correlator 56 is zero, since there is no voltage at the output of the intermediate frequency amplifier 38. The keys 68, 71 and 72 do not open and the false signal (interference) received on the first mirror channel at a frequency f ₃₁ is suppressed.

= f_пр
f₂₁= f₃₂-f

= 3f_пр
Однако только напpяжение с частотой f₂₂ попадает в полосу пpопускания Δ f_п усилителя 38 промежуточной частоты и на второй вход коррелятора 56. В этом случае выходное напряжение коррелятора 56 также равно нулю, так как напряжение на выходе усилителя 37 промежуточной частоты отсутствует. Ключи 68, 71 и 72 не открываются и ложный сигнал (помеха), принимаемый по второму зеркальному каналу на частоте f₃₂, подавляется.If a false signal (interference) is received through the second mirror channel at a frequency of f ₃₂ , then in mixers 37 and 38 it is converted to voltages of the following frequencies:
f ₂₂ = f ₃₂ -f

= f _ol
f ₂₁ = f ₃₂ -f

= 3f _pr
However, only the voltage with a frequency f ₂₂ falls into the passband Δ f _{p of the} intermediate frequency amplifier 38 and the second input of the correlator 56. In this case, the output voltage of the correlator 56 is also zero, since there is no voltage at the output of the intermediate frequency amplifier 37. The keys 68, 71 and 72 do not open and the false signal (interference) received on the second mirror channel at a frequency of f ₃₂ is suppressed.

-f₃₁= f_пр
f₁₂= f

-f₃₁= 3f_пр
f₂₂= f₃₂-f

= f_пр
f₂₁= f₃₂-f

= 3f_пр
При этом напряжения с частотами f₁₁ и f₂₂ попадают в полосу пропускания Δ f_п усилителей 37 и 38 промежуточной частоты. Однако ключи 68, 71 и 72 не открываются. Это объясняется тем, что ложные сигналы (помехи) принимаются на разных зеркальных частотах f₃₁ и f₃₂, поэтому между канальными напряжениями, выделяемыми усилителями 37 и 38 промежуточной частоты, существует слабая корреляционная связь. Кроме того, следует отметить, что корреляционная функция помех не имеет ярко выраженного максимума, как это наблюдается в ФМн-сигналах. Выходное напряжение коррелятора 56 не превышает порогового уровня U_пор2 в пороговом блоке 60. Последний не срабатывает, ключи 68, 71 и 72 не открываются и ложные сигналы (помехи), принимаемые одновременно по первому и второму зеркальным каналам на частотах f₃₁ и f₃₂, подавляются.If false signals (interference) are simultaneously received on the first and second mirror channels at frequencies f ₃₁ and f ₃₂ , then in mixers 37 and 38 they are converted to voltages of the following frequencies:
f ₁₁ = f

-f ₃₁ = f _pr
f ₁₂ = f

-f ₃₁ = 3f _pr
f ₂₂ = f ₃₂ -f

= f _ol
f ₂₁ = f ₃₂ -f

= 3f _pr
When this voltage with frequencies f ₁₁ and f ₂₂ fall into the passband Δ f _p amplifiers 37 and 38 of an intermediate frequency. However, keys 68, 71 and 72 do not open. This is because false signals (interference) are received at different mirror frequencies f ₃₁ and f ₃₂ , so there is a weak correlation between the channel voltages emitted by the amplifiers 37 and 38 of the intermediate frequency. In addition, it should be noted that the correlation function of interference does not have a pronounced maximum, as is observed in PSK signals. The output voltage of the correlator 56 does not exceed the threshold level U _pore2 in the threshold block 60. The latter does not work, the keys 68, 71 and 72 do not open and false signals (interference) received simultaneously on the first and second mirror channels at frequencies f ₃₁ and f ₃₂ , are suppressed.

 For a similar reason, false signals (interference) received via other additional (combinational) channels are also suppressed. All information received onboard the satellite from the ARB and radio sensors is included in the format of the digital message transmitted to PPI 12. The generated digital message is transmitted at a speed of 2400 bit / s in real time after preprocessing and is simultaneously recorded in storage devices 8 and 10. Transmission information from storage devices 8 and 10 is produced in the same format and at the same speed as in real time, as a result of which PPI 12 receives stored in on-board storage devices 8 and 10 messages from the ARB 2 and the radio sensor 3 of the vehicle, accumulated during the full revolution of the satellite around the Earth. If at the time of the transmission of information from the storage devices 8 and 10 to the input of the receivers 7 or 7 of the satellite receives a signal from the ARB 2 or from the radio sensor 3 of the vehicle, the transmission is interrupted for signal processing, information about which is included in the message format for transmission to the PPI after processing 12. The corresponding binary number is included in the message, showing the type of transmission mode: the real time scale or from the storage devices, in addition, the transmission time of the last message from the storage is identified stroystv.

 At the input of the airborne transmitter 11 is fed information from receiving devices 5,6,7 and storage devices 8,10. The radiation power of the transmitter 11 can be adjusted from the ground control system. The transmitter also carries out phase modulation of the carrier frequency with a composite signal in stages of its formation (before multiplication). Then the oscillation is transferred to a frequency of 1544.5 MHz, amplified to the required level and fed to the input of the transmitting antenna.

 A block diagram of a typical PPI is shown in FIG. The received signal is amplified and, after conversion (lowering) of the frequency, is supplied to a linear demodulator 85 for composing a composite spectrum including all informational components of the stream. Those parts of the compositional spectrum that contain useful information are highlighted and further processed. Processing fragments of the stream is carried out in accordance with the capabilities and software of each PPI. If analog storage devices are installed on the PPI, they can be used as backup equipment in case of a malfunction of the main processor.

-f

= 2f_пр , выбраны симметричными относительно несущей частоты f_c принимаемых сигналов
f_c-f

= f

-f_c = f_пр и корреляционной обработкой канальных сигналов.Thus, the proposed system in comparison with the base provides search, detection and location of vehicles stolen by intruders or only being stolen. Wherein the search, detection, temporary and partial selection PSK signals emitted by transmitters vehicles performed in a predetermined frequency range by a synchronous serial tuning frequency f _r1 and f _r2 oscillators and a convolution of the spectrum received PSK-signals. In this case suppression of additional (and mirror combination) receiving channels inherent onboard receiver device, and thus increase noise immunity achieved by using two local oscillators, the frequency f _r1 and f _r2 which are spaced by twice the value of the intermediate frequency f

-f

= 2f, _etc., selected symmetric about the carrier frequency f _c of the received signals
f _c -f

= f

-f _c = f _ol and correlation processing of channel signals.

 Accurate and unambiguous direction finding of vehicle radio sensors is achieved by correlation processing of received PSK signals and using two scales: a phase scale for measuring the phase difference between received PSK signals (an accurate but ambiguous scale), a time scale for measuring the delay time of a signal received by one antenna, in relation to the signal received by another antenna (rough, but unambiguous scale). In addition, the phase scale operates at twice the intermediate frequency. This is achieved by multiplying the channel QPSK signals converted in frequency, as a result of which the convolution of their spectrum is carried out, and the selection of the harmonic voltage using a narrow-band filter. Therefore, by convolving the spectrum of the received QPSK signals and narrow-band filtering, a significant part of the interference is filtered out and the sensitivity of the on-board receiver is increased.

 In addition, the proposed system eliminates the ambiguity of direction finding due to invariance to the side of the deviation of the vehicle radio sensors from the equal direction. Digital presentation of direction finding results provides an opportunity for their long-term storage and registration, long-distance transmission via communication channels and interfacing with computer technology.

 To select a digital message, which contains information about the vehicle’s ownership (country), its identification number and owner, an autocorrelation method for detecting PSK signals is used, which is devoid of the phenomenon of “reverse” operation. Thus, the functionality of the system is expanded.

Claims (3)

 1. SATELLITE SYSTEM FOR DETERMINING THE LOCATION OF VESSELS AND AIRCRAFT AFTER AN ACCIDENT, containing emergency beacons, a second on-board receiving device in series, a first on-board storage device and an on-board transmitter, the second input of which is connected to the output of the first onboard receiving device, and the third input is the second on-board receiving device, the receiving device in series, the first information processing device, the interface device with communication networks, the second input to through a second information processing device, a monitoring and control device and a communication device of search and rescue organizations are connected to the output of the receiving device, characterized in that the vehicle radio sensors and the third on-board receiving device and the second on-board storage device, the output of which is connected in series, are inserted into it with a fourth input of the airborne transmitter, the fifth input of which is connected to the output of the third airborne receiving device, a third processing device and information, the output of which is connected to the third input of the device for interfacing with communication networks, and the input to the output of the receiving device.  2. The system according to claim 1, characterized in that each vehicle’s radio sensor contains a power source, a resistor, an LED indicator, a remote switch with two antiphase windings, a reed switch, a modulating code generator, a relay, a master oscillator, a phase manipulator and a transmitter with an antenna, a resistor is connected in series to the positive bus of the power supply, the first contacts of the second winding of the remote switch and an LED indicator, the first one is connected to the positive bus of the power supply remote switch winding, the first contacts of the first winding of the remote switch and the reed switch, the input of which is connected through a second winding of the remote switch and the second contacts of its first winding to the positive bus, the ignition key, relay contacts and ignition coil are connected in series to the positive bus of the power source, to the output of the master oscillator is connected in series to a phase manipulator, the second input of which is connected to the output of the generator of the modulating code, to the output to A relay coil and a transmitter with an antenna are connected, the power source is connected to a modulating code generator, a master oscillator, a phase manipulator and a transmitter through the ignition key and the second contacts of the second winding of the remote switch.  3. The system according to claim 1, characterized in that the third on-board receiving device comprises two antennas, two mixers, two local oscillators, a frequency search unit, two intermediate frequency amplifiers, two multipliers, two narrow-band filters, a frequency doubler, two spectrum width meters , two comparison blocks, two threshold blocks, four keys, two multiplexers, a subtraction block, an adder modulo two, a correlator, a shift register, a clock generator, an And element, a delay element, a counter, two storage registers, a digital comparator, two lines and delays, two phase detectors and a voltage-to-code converter, the first mixer is connected in series to the output of the first antenna, the second input of which is connected through the first local oscillator to the output of the frequency search unit, the first intermediate frequency amplifier, frequency doubler, the second spectral width meter, the first block comparison, the second input of which through the first meter of the spectral width is connected to the output of the first intermediate frequency amplifier, the first threshold block, the second input of which is connected via the first delay line nen with its output, the first key, the second input of which is connected to the output of the first intermediate frequency amplifier, the fifth key, the second input of which is connected through the second threshold block to the output of the correlator, the second delay line and the first phase detector, the second input of which is connected to the output of the fifth key , and the output is the third output of the on-board receiver, the second mixer is connected in series to the second antenna, the second input of which is connected through the second local oscillator to the output of the frequency search unit, the second amplifier an intermediate frequency amplifier, a first multiplexer, the second input of which is connected to the output of the first intermediate frequency amplifier, a correlator, a second comparison unit, a shift register, the second input of which is connected to the output of the clock generator, element And, the second input of which is connected to the output of the clock generator, a counter, the second input of which through the delay element is connected to the output of the shift register, the first storage register, the second input of which is connected to the output of the shift register, the third key, the second input of which connected to the output of the second threshold unit, a digital comparator, the second input of which is connected to the output of the second storage register, and a second storage register, the second input of which is connected to the output of the third key, and the output is the second output of the on-board receiver, to the output of the second intermediate frequency amplifier in series a second multiplexer is connected, the second input of which is connected to the output of the first intermediate frequency amplifier, and a second switch, the second input of which is connected to the output of the first threshold block, and the output is connected to the second input of the correlator, the subtraction unit is connected in series to the output of the first intermediate frequency amplifier, the second input of which is connected to the output of the second intermediate frequency amplifier, the adder is modulo two, the second input of which is connected to the output of the first channel of the second comparison unit, and the shaper, the output of which is the first output of the on-board receiver, the input of the frequency search unit is connected to the output of the first threshold unit, the third inputs of the multiplexers are connected to the output of the sum there are two modules, the fourth key is connected in series to the output of the second intermediate frequency amplifier, the second input of which is connected to the output of the second threshold unit, the second multiplier, the second input of which is connected to the output of the first intermediate frequency amplifier, the second narrow-band filter, the second phase detector, the second input which through series-connected the first multiplier and the first narrow-band filter is connected to the second output of the local oscillators, and the voltage converter is a code whose output is fourth output board receiver.

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