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WO2025106102A1 - Dispositif radar à impulsions et procédé associé - Google Patents

Dispositif radar à impulsions et procédé associé Download PDF

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Publication number
WO2025106102A1
WO2025106102A1 PCT/US2024/022583 US2024022583W WO2025106102A1 WO 2025106102 A1 WO2025106102 A1 WO 2025106102A1 US 2024022583 W US2024022583 W US 2024022583W WO 2025106102 A1 WO2025106102 A1 WO 2025106102A1
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Prior art keywords
output
pulse
frequency
radar
impulse
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Inventor
Matthew IAN BURNS
Tyler DOTSON
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Smartauger Technologies LLC
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Smartauger Technologies LLC
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Classifications

    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/03—Details of HF subsystems specially adapted therefor, e.g. common to transmitter and receiver
    • G01S7/032—Constructional details for solid-state radar subsystems
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G01S13/10—Systems for measuring distance only using transmission of interrupted, pulse modulated waves
    • G01S13/12—Systems for measuring distance only using transmission of interrupted, pulse modulated waves wherein the pulse-recurrence frequency is varied to provide a desired time relationship between the transmission of a pulse and the receipt of the echo of a preceding pulse
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/28—Details of pulse systems
    • G01S7/282—Transmitters
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/28—Details of pulse systems
    • G01S7/285—Receivers
    • G01S7/292—Extracting wanted echo-signals
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/28—Details of pulse systems
    • G01S7/285—Receivers
    • G01S7/292—Extracting wanted echo-signals
    • G01S7/2923—Extracting wanted echo-signals based on data belonging to a number of consecutive radar periods
    • G01S7/2925—Extracting wanted echo-signals based on data belonging to a number of consecutive radar periods by using shape of radiation pattern

Definitions

  • the invention generally relates to the field of impulse radar detection circuits.
  • Modem commercial radars can generally be characterized as one of two broad classes.
  • One class is the low-cost integrated circuit (IC) radar that outputs data through standard bus architectures such as SPI, I 2 C, and so on.
  • IC radars operate at high frequencies around 70 to 80 GHz. Consequently this ty pe of radar is operable over very' short ranges in air on the order of a few meters, and no more than a few centimeters in soil.
  • IC radars can be used in connection with directional antennas, and find application in devices such as vehicle collision avoidance.
  • SoC system-on-a-chip
  • DDS direct digital synthesis
  • impulse radars work by sending out a series of pulses and sampling the corresponding reflected signals after a fixed number of picoseconds. The data can be used to reconstruct the reflected signal as one pulse without requiring the more costly equipment used in frequency domain radars. The tradeoff tends to be an extended data acquisition time.
  • a microcontroller's onboard ADC might be used to quantize analog signals, but the ADC is controlled by the microcontroller’s CPU. There are several problems with this architecture that make it untenable for impulse radar applications. [0011] First, the CPU processing time is significant, which vastly reduces the ADC’s sampling rate.
  • Embodiments comprise multipurpose miniaturized impulse radars.
  • Such radars include circuitry for producing an RF pulse and a LO pulse that are substantially the same but differ enough in frequency to permit a period-to-period phase shift.
  • the periods of the RF and LO pulse can be arbitrarily large while the pulse width is very' short.
  • the short pulse width provides the desired level of range resolution while the arbitrarily large period permits the use of low-cost ADCs to digitize the RF return.
  • Each period of corresponding RF and LO pulses phase shift by a constant amount, the magnitude of which is selected to ensure that each RF return pulse is sampled at enough points to satisfy Nyquist’s Theorem.
  • FIG. 2 is a stacked plot showing full frames of higher frequency and lower frequency pulses
  • FIG. 3A is a plot of a single RF return pulse in isolation
  • FIG. 4B is a plot showing the effect of phase shift over many pulses
  • FIG. 5 is a pair of plots showing the pulse train before and after amplification
  • FIG. 6 is a plot of a pulse after applying a low pass filter
  • FIG. 7 is a plot of a reconstructed A-Scan
  • FIG. 8 is a circuit diagram of an impulse generator according an embodiment
  • FIG. 9 is a circuit diagram of a sampling mixer according to an embodiment
  • FIG. 10A is a perspective view of an embodiment with the top section of the housing removed exposing the printed circuit board;
  • FIG. 10B is a perspective view of an embodiment with the housing assembled;
  • FIG. 11 is a drawing of representative A-Scan data showing reflections from a plurality of target at different ranges;
  • FIG. 12 is a plot of representative B-Scan data.
  • microcontrollers and their relatively low-performance onboard analog-to-digital converter (ADC) units are capable of functioning in an impulse radar provided that an architecture according to the present invention is used.
  • embodiments accomplish this by limiting the CPU to performing operations that are timing-flexible.
  • the CPU may be used to setup the impulse radar, but thereafter the CPU does not intervene in the function of the ADC or in writing quantized data to memory.
  • embodiments apply a timing technology of the invention to spread sampling out over wide time intervals. In this arrangement, low- performance ADCs, like those onboard typical microcontrollers, are sufficient.
  • the CPU instructs a clock circuit to output two periodic square waves having slightly different pulse repetition frequencies.
  • the square waves are each separately communicated to one of two impulse generators having identical architectures, converting the square waves to broadband pulses having a sufficiently narrow pulse width to provide acceptable range resolution as determined by a given application.
  • both the higher and lower frequency pulses may have identical frequency content.
  • the lower frequency pulse is capable of serving as a local oscillator for triggering a sampling mixer to sample RF returns originating from the higher frequency pulse, and that it is also capable of phase shifting relative to the higher frequency pulse as described herein. Provided these conditions are met.
  • the output of the second impulse generator is substantially the same as the output of the first impulse generator.
  • the frequency content of both pulses is between 500 MHz and 5 GHz.
  • the content starts to taper off at 3 GHz and is very low as frequency increases to 5 GHz.
  • the most important frequencies are approximately 2 GHz.
  • Suitable frequencies are determined in part by their ability to transmit through a medium. For instance. 2 GHz has a useful penetration of damp soil due its extinction coefficient at that frequency. Therefore, this frequency can be used to detect targets such as buried utilities.
  • Suitable frequencies are also determined in part by signal strength at the given frequency. A frequency with good penetration of a given medium may be unusable if its power is too low. Conversely, even a relatively high power signal may be unusable if the medium’s extinction coefficient at that frequency is too high.
  • Embodiments include but are not limited to semiconductor-based impulse generators, such as those using step recovery diodes (SRDs), to generate narrow pulse width impulses from the high and low frequency square waves of the clock generator circuit.
  • SRDs step recovery diodes
  • Suitable pulse widths provide an acceptable range resolution according to Equation 3, where AR is range resolution, c is the speed of light, and r is pulse width. Accordingly, a pulse width of 500 picoseconds provides a range resolution of 7.5 cm.
  • c x T (3 x 10 8 m/s)(5 x 10" lo s)
  • FIG. 8 is a diagram of an impulse generator circuit 800 according to one embodiment.
  • a square wave clock signal 802 is shown as input on the left, and a corresponding pair of output impulses 804 is shown above-right.
  • the impulses 804 are relatively narrow in pulse width compared to the clock signal 802.
  • Step recovery diodes (SRD) 806A and 806B are used to narrow' the pulse w idth of the output.
  • SRDs are MACOM’s MMDB30 and MMDB45 step recovery diodes. According to the manufacturer’s specifications, the MMDB30 SRD has a lifetime betw een 1-4 ns and a transition time between 30-38 ps.
  • the MMDB45 SRD has a lifetime between 3-8 ns and a transition time between 45-58 ps.
  • the output pulse shape of the SRDs can be modified using an appropriate network to adjust the SRD’s input signal. More specifically, where the SRD is the MMDB45, making the resistors (808 and 810) 50 Ohms provides an output pulse width of about 500 ps. The person having ordinary skill will recognize that increasing the resistance of 808 and 810 would increase the pulse width. For example, changing the resistors 808 and 810 from 50 Ohms to 400 Ohms would roughly double the output pulse width. Accordingly, embodiments are not limited to 500 ps pulse widths. Rather, the skilled artisan, having the benefit of this description, will understand how to adjust the circuit to vary pulse width to suit range resolution requirements of a given application.
  • diodes 806A and 806B are also configured to modify the frequency content of the inputted clock signal 802. More specifically, the diodes function as a frequency doubler and a harmonic generator. Therefore, the clock circuit’s fundamental frequency can be converted to a broadband signal having much richer frequency content, which can be used to improve range resolution.
  • the impulse generator of FIG. 8 includes a schottky diode 810 that functions to rapidly switch the circuit between charging and discharging states.
  • the pulse generated from the higher frequency signal is amplified and communicated to an antenna where it is transmitted at a predetermined pulse repetition frequency (PRF).
  • PRF pulse repetition frequency
  • the individual transmitted pulses are known as radar pulses. These pulses are transmitted in frames of a predetermined number equally spaced in time. The transmitted pulses are reflected from a target, and a portion of the transmitted energy is returned to the antenna.
  • the slight PRF difference between the higher and lower frequency pulses causes the waveforms to phase shift at a constant rate so that each successive coinciding period of high and low frequency pulses shift in constant fractions of a period according to Eq. 4 where At is the time step in picoseconds,/? is the higher frequency, and fi is the lower frequency.
  • This phase shift corresponds to an effective sampling frequency, where the lower frequency pulse triggers a sampling mixer to sample the higher frequency radar return pulse.
  • the frequency difference is within a range limited by Nyquist’s Theorem (Eq. 5), which states that in order to sample a signal without loss of information, the sampling frequency fi must be at least twice the highest frequency fmax contained in the signal.
  • the highest frequency content f max of a radar pulse is 5GHz then, according to Nyquist, at the minimum sampling frequency f s of 10 GHz a measurement is made every 100 picoseconds. In other words, the difference between the higher and lower frequencies must be small enough to trigger sampling in 100 picosecond steps from period to period. At 3 GHz. the minimum is 6 GHz or 167 picosecond steps, and at 2 GHz the minimum is at least 4 GHz or 250 picosecond steps. Nyquist limits the minimum sampling rate, but the maximum sampling rate is only limited by the frequency resolution of the clock generator circuit. In other words, the minimum frequency difference between two clock signals that a clock circuit is capable of generating defines the upper limit of effective sampling rate. While the exact frequency difference is not critical, what is important is that the difference is set so that the resultant phase shift is small enough to sufficiently sample the returning radar pulses according to Nyquist’s Theorem.
  • Embodiments are capable of arbitrarily high effective sampling rates with low-end ADCs capable of no more than 1 or 2 megasamples per second (MSPS) because the individual pulses in a frame can be spaced arbitrarily far apart e.g., one or two microseconds apart. Accordingly, any ADC can be used in an embodiment provided that the ADC’s sampling rate is compatible with the application. Since lower ADC sampling rates correspond to slower A-Scan and B-Scan production, a slower ADC will extend data acquisition time. Data acquisition times on the order of seconds can be acceptable in applications such as a ground penetrating radar where the target is stationary, but could be unacceptable in applications where the target is in motion at high rate speed e g., a missile traveling at Mach 5. In such cases, a faster ADC would be warranted; however, a sampling technology according to embodiments of the invention may still be used.
  • MSPS megasamples per second
  • the lower frequency pulse is a local oscillator (LO) used to trigger a sampling mixer to sample the returning higher frequency radar pulse after it is reflected from a target. Therefore, the constant phase shift enables embodiments to sample the reflected wave across the waveform at enough points to satisfy Nyquist’s Theorem, regardless of when the pulse arrives, so long as it arrives during a time window defined by the PRF. The samples are then used to reconstruct a single waveform, representative of the individual RF return pulses.
  • LO local oscillator
  • FIG. 9 is an illustration of a sampling mixer 900 according to an embodiment of the invention.
  • the RF return signal is carried by line 904, while the positive and negative components of the local oscillator signal are carried on lines 902A and 902B.
  • the LO signal arrives at the schottky diodes 906 and 908 it biases the diodes for a period approximating the pulse width of the LO pulse.
  • the pulse width may be about 500 ps.
  • a reflected RF pulse may coincide in time. In such instances, the energy added by the RF pulse causes the schottky diodes 906 and 908 to switch from the off state to the on state.
  • the resulting signal is sent to an analog to digital converter 910 where it is quantized.
  • the analog to digital converter (ADC) 910 can be any relatively low performance ADC such as that which is found on a conventional low- cost microcontroller. Such ADCs typically operate around 1 to 2 MSPS. While slow among ADCs in general, this is still too fast for the CPU of such a low-cost microcontroller to handle writing the data to an onboard buffer memory. Therefore, embodiments use DMA circuit 920 to write to a buffer memory 930. From there the data can be transferred by conventional bus architectures such as USB 940 to a personal computer 950 or similar general purpose computing device e.g.. a tablet computer. The PC can then conduct asynchronous image reconstruction processes.
  • USB 940 such as USB 940
  • a personal computer 950 or similar general purpose computing device e.g.. a tablet computer.
  • the PC can then conduct asynchronous image reconstruction processes.
  • the ADC’s output can be used to construct A-Scans as shown in FIG. 11 which may be assembled into B-Scans as shown in FIG. 12.
  • the number of pulses transmited in a frame corresponds to the range R of the radar. In general, more pulses in a frame corresponds to greater range. The reason is that the higher and lower frequency pulses phase shift at a constant rate from period to period.
  • FIG. 4B illustrates the progressive phase shift between the 782 kHz and 782.01 kHz impulses, as produced by the impulse generator, and before amplification or transmission of the RF pulse.
  • Plot (a) of FIG. 4B shows a 16.4xl0 12 second phase shift (At) in the time domain between the second period impulses.
  • each successive period multiplies the phase shift At by an integer corresponding to the period.
  • the total phase shift is M*At, where M is a pulse in the middle of a frame.
  • the total phase shift Attotai is N*At, where N is the last pulse in the frame, as shown in Eq. 6.
  • Total phase shift Attotai corresponds to a time delay between generating the lower frequency pulse and generating the higher frequency pulse, thereby providing time for the RF pulse to travel down-range, reflect from a target, and reach the sampling mixer at about the same time as the lower frequency trigger pulse. If N is 1000, then the time delay Attotai given by Eq. 6 is 1.64xl0' 9 s, corresponding to a maximum distance to the target i.e., range R, of 2.5 meters according to Eq. 7.
  • the lower frequency pulse is generated 1.64xl0' 9 seconds after the higher frequency pulse, which provides sufficient time for the RF pulse to travel to a target 2.5 meters away and return to the antenna during the 500 ps window where the lower frequency pulse is forward biasing the schotky diode.
  • the higher frequency is 782.01 kHz, so the period between higher frequency pulses is 1.28xl0' 6 s.
  • the RF signal can travel to a target and back in less than 1.28x1 O' 6 seconds, no aliasing will result.
  • This provides a theoretical maximum range Rmax and corresponding maximum number of pulses Nmax and a maximum total phase shift Attotai. Given that 1.28x1 O' 6 seconds corresponds to 384 meters at the speed of light, a target up 192 meters away from the antenna can be detected without aliasing, assuming signal strength is sufficient to return a measurable RF pulse.
  • a range of 192 meters corresponds to about 780,000 pulses per frame where the pulses shift in increments of 1.64xl0 12 seconds.
  • the person having ordinary’ skill in the art will understand that such a large number of pulses per frame will greatly increase power consumption by nearly three orders of magnitude compared to a frame of 1000 pulses. Therefore, it is generally desirable to use as few pulses per frame as necessary’ to meet range requirements.
  • the person having ordinary skill will understand how to adjust At, signal strength, pulse width, frequency content, and N to produce a desired maximum operable range Rmax and range resolution AR.
  • PRF pulse repetition frequency
  • the faster the rate of transmission the shorter the maximum range Rmax, but the faster the rate of reconstructing A-Scans and the faster B-Scans can be generated.
  • competing factors place limits on the rate of transmission. For example, as discussed above, faster transmission rates correspond to smaller maximum effective ranges of a radar because a pulse must travel to a target, be reflected, and travel back to the receiver before the next pulse is transmitted. If the rate of transmission is too fast, aliasing will result, producing false signals.
  • Functional transmission rates within the scope of the invention leave sufficient time for a radar pulse’s maximum expected time-of-flight.
  • the maximum effective range Rmax is about 192 m. Provided the target is never more than 192 m from the embodiment, a transmission rate of 1.28 MHz is sufficient. Larger maximum ranges would require slower PRFs.
  • penetration is expected to be relatively’ low, on the order of a few meters at low power levels and broadband frequency content peaking at about 2 GHz.
  • Rmax sufficiently few pulses per frame, and permits an arbitrarily large PRF. Therefore, the PRF can be set to coincide with the sampling capability of the onboard ADCs of a typical microcontroller e.g., 1 to 2 MSPS.
  • the present invention enables the use of much slow er low-cost ADCs that are generally limited to sampling at rates of a couple MHz e.g., 2 MSPS.
  • the rate of transmission also corresponds to the number of frames that can be transmitted per second.
  • the number of pulses in a frame as the number increases, the length of time and pow er required to generate a B-Scan may increase to an unacceptable degree.
  • this w ould impact an embodiment's capacity to include A-Scan averaging to improve signal-to-noise due to the time required to generate additional pulses. Accordingly, embodiments must balance a number of factors to achieve desired performance results. The process of balancing these factors is within the ordinary skill in the art as a matter of design choice.
  • One non-limiting example comprises a Silicon Labs Mighty 7 Gecko (EFR32) microcontroller that sets up a Texas Instrument clock generator (CDCM6208).
  • the clock generator creates two frequencies very close to each other, but different enough that the rising edges strobe past each other.
  • the higher frequency pulse is 782.01 kHz and the lower frequency pulse is 782 kHz, a difference 10 Hz.
  • Both pulses are sent to identical impulse generators that convert the clock generator's step function to a 500 picosecond pulse width broadband pulse, approximating a delta function and providing a 7.5 cm range resolution, A/?.
  • the frequency content of both pulses is between 500 MHz and 5 GHz.
  • the content peaks around 2 GHz starts to taper off at 3 GHz and is very low as frequency increases to 5 GHz.
  • the higher frequency 782.01 kHz pulse is amplified to about 35 dBm and sent out through a transmit antenna.
  • this level of amplification provides sufficient power to compensate for losses in the return signal. It will be understood by the skilled artisan that some portion of the energy transmitted will interact w ith a target, if present, and be reflected back to the antenna and measured. The person having ordinary skill will understand how to adjust amplification to provide sufficient power for making measurements over given ranges in given media.
  • the lower frequency 782 kHz pulse is amplified to about 15 dBm.
  • This pulse triggers a sampling mixer.
  • the degree of amplification is selected to sufficiently forward bias the schottky diode of the sampling mixer so that a returning RF pulse will be sampled if it arrives during the roughly 500 ps that the trigger pulse maintains the bias.
  • the sampling mixer captures the value of the higher frequency 782.01 kHz pulse and uses the resulting signal to charge a capacitor.
  • the capacitor’s charge fluctuates until the lower frequency pulse stops forw ard biasing, at which point the value of the capacitor is fixed and can be measured.
  • FIG. 1 is a plot 100 showing the higher frequency 104 and lower frequency 102 waveforms.
  • the vertical line 106 show s the point 108 w here the peak of the lower frequency pulse coincides in time with the higher frequency pulse 104. This point is where the sampling mixer is triggered, and therefore where the higher frequency pulse 104 is sampled.
  • Pulses may be organized into frames. For instance, in the present example a 782.01 kHz radar pulse may be transmitted in a frame of 512 pulses at a rate of 1 MHz. That is, the identical radar pulse is transmitted 512 times, comprising a frame of 512 pulses, and they are transmitted at a rate of 1 MHz.
  • FIG. 2 is a plot 200 show ing a train of higher 104 and low er 102 frequency pulses spaced apart at approximately Ips intervals, corresponding to the transmission rate of 1 MHz.
  • FIG. 2 shows an entire frame 206 of 1024 pulses spaced at approximately I s intervals.
  • the person having ordinary skill in the art will understand that the number of pulses per frame can vary', as w ell as the rate at w hich they are transmitted.
  • FIG. 4A is a plot 400 showing the phase shift At of three successive idealized periods of high 104 and low 102 frequency pulses, with the earliest period being at the bottom, and the latest period at the top. Each successive pulse shifts by a constant amount At.
  • FIG. 5 is a plot 500 of a pulse train before 510 and after 520 amplification.
  • the lower plot 510 shows the output of the sampling mixer prior to amplification, while the upper plot shows the pulse train post amplification 520.
  • High frequency noise 522 is still visible in the amplified signal 520, resulting from charge leakage from the capacitors of the sampling mixer.
  • FIG. 6 is a plot 600 of the pulse 610 after applying a low-pass filter.
  • FIG. 7 is a plot 700 of the reconstructed A- Scan pulse 710.
  • Equation 9 Data obtained from the individual pulses of a frame is used to reconstruct a single A-Scan waveform, one frame corresponding to one A-Scan.
  • the number of samples across the waveform is given by Equation 9, where PW is pulse width.
  • PW pulse width.
  • the number of samples at unique phase angles (represented in terms of time as At) is a minimum of 30; how ever, the fractional component of N prevents/; pulses over 30 from triggering the sampling mixer at the same phase angles as the preceding 30.
  • One thousand successive samples will trigger the sampling mixer at one thousand unique phase angles.
  • a DMA circuit handles writing the ADC’s output to a buffer memory 7 . From there the data can be transmitted over known bus architectures such as USB 2.0 or USB-C to an off-board computer that assembles the A-Scans (FIG. 11) into B-Scans (FIG. 12) and conducts image analysis.
  • the vertical axis corresponds to sample number
  • the horizontal axis corresponds to A-scan number
  • a B-Scan is a plot of vertically 7 arranged A- Scans with the first A-Scan appearing on the left and the last A-Scan appearing on the right. If the antenna moves horizontally at a known constant rate, the horizontal axis can be converted to the distance traveled by the antenna. Generally, traveling down the vertical axis from top to bottom corresponds to increasing distance from the antenna. In some embodiments discussed herein, samples are taken every 7 16.4 picoseconds. Since the radar pulse travels at the speed of light each 16.4 ps sample corresponds 2.5 mm distance from the antenna. Therefore, sample number 2000 corresponds to 5 meters from the antenna.
  • the zero point is the first moment that the LO pulse triggers the sampling mixer.
  • no radar can be detected until the first pulse is reflected from a target. So, samples 0 through roughly 550 can be neglected.
  • Embodiments may make a vertical axis correction by subtracting out the samples prior to the first RF pulse being transmitted from the antenna, making zero correspond to the position of the transmitter.
  • the gray scale plot indicates signal strength.
  • White and black represent the strongest signals, but have opposite signs.
  • the gray shade midway between black and white is a zero reading.
  • the antenna is stationary, resulting in horizontal bands.
  • the hyperbolic waves 1202 correspond to a single target.
  • known mathematical methods may be applied to the B-Scan data to reveal the shape of the target; however, in other embodiments a machine learning data model may be applied to recognize targets in the B- Scan data.
  • Onboard ADCs may have an effective number of bits (ENOB) as low as 8 but typically range from 8 to 14.
  • ENOB effective number of bits
  • a 12 ENOB ADC provides a SNR, according to Eq. 10, of 74 dB, where N is the effective number of bits.
  • a 74 dB SNR leaves ample dynamic range.
  • the position of a target is determined based on the time-of-flight of a transmitted radar pulse (f ), which is determined by the difference in phase angle between the higher and lower frequency waveforms after accounting to phase shift effects due to their slight frequency difference. For example if, as in the present example, the difference between the lower frequency fi pulse and the higher frequency f2 pulse is 10 Hz, then according to Eq. 4 the At from one period to the next is 16.4 ps. If the 2 pulse is detected at 516.4 ps then the extra 500.0 ps is due to time-of-flight. A 500 ps time of flight corresponds to a travel distance dtravei of 1.5 m (Eq. 11) and a target distance dt arg et of 0.75 m (Eq. 12).
  • f transmitted radar pulse
  • FIGS. 10A and 10B show an embodiment 1000 comprising a small printed circuit board (PCB) 1004 in a two-part anodized aluminum heatsink case 1006 A, 1006B.
  • the circuitry on this board 1004 is described elsewhere herein.
  • board 1004 includes connectors 1002A, 1002B to an external power supply, and a USB port 1008.
  • the PCB 1004 is a fully self-contained miniaturized impulse radar containing all necessary components except a power source and data analysis capabilities, both of which are off-board.
  • the ruggedized case 1006A, 1006B and small size permits the embodiment 1000 to be used to provide radar capabilities to a wide range of devices including, without limitation, hand-held ground penetrating radar units, warehouse robots, vehicle collision avoidance, and machine vision.
  • the antenna of such a device may be directed in a user-selected direction. For instance, a user may expect a target to potentially be underground and may point the antenna dow nw ard where the target is suspected of being located. Such a direction is a user-selected direction according to the invention.
  • Embodiments may also include onboard geolocating components, such as global positioning system (GPS) components, for locating the radar. Accordingly, such embodiments are configured to map the radar data to GPS data so that targets can be associated with a specific geolocation. This may be especially beneficial in ground penetrating radars for locating buried utilities. Once found and recorded, the location of the utility can be stored and recalled as needed.
  • GPS global positioning system

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  • Radar Systems Or Details Thereof (AREA)

Abstract

L'invention concerne un radar à impulsions miniaturisé. Le radar utilise une légère différence de fréquence entre un oscillateur local et un signal RF pour générer des balayages "A-scan" de haute qualité à l'aide d'un convertisseur analogique-numérique d'un microcontrôleur peu coûteux. Un circuit DMA est utilisé pour écrire des sorties de convertisseur analogique-numérique dans une mémoire tampon. Le radar a diverses applications, par exemple, comme radar à pénétration de sol permettant de détecter les services publics enfouis, pour l'évitement de collision automobile, la navigation de robots autonomes et la vision artificielle de manière générale.
PCT/US2024/022583 2023-11-13 2024-04-02 Dispositif radar à impulsions et procédé associé Pending WO2025106102A1 (fr)

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US20090092284A1 (en) * 1995-06-07 2009-04-09 Automotive Technologies International, Inc. Light Modulation Techniques for Imaging Objects in or around a Vehicle
US20090102703A1 (en) * 2007-10-18 2009-04-23 Farrokh Mohamadi Scanning ultra wideband impulse radar
US20170363711A1 (en) * 2016-06-16 2017-12-21 Texas Instruments Incorporated Radar hardware accelerator

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20090092284A1 (en) * 1995-06-07 2009-04-09 Automotive Technologies International, Inc. Light Modulation Techniques for Imaging Objects in or around a Vehicle
US20040249257A1 (en) * 2003-06-04 2004-12-09 Tupin Joe Paul Article of manufacture for extracting physiological data using ultra-wideband radar and improved signal processing techniques
US20090102703A1 (en) * 2007-10-18 2009-04-23 Farrokh Mohamadi Scanning ultra wideband impulse radar
US20170363711A1 (en) * 2016-06-16 2017-12-21 Texas Instruments Incorporated Radar hardware accelerator

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
MALAJNER MARKO, ŠIPOŠ DANIJEL, GLEICH DUŠAN: "Design of a Low-Cost Ultra-Wide-Band Radar Platform", SENSORS, MDPI, CH, vol. 20, no. 10, CH , pages 2867, XP093317983, ISSN: 1424-8220, DOI: 10.3390/s20102867 *

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