WO2025171451A1

WO2025171451A1 - An improved metal detector

Info

Publication number: WO2025171451A1
Application number: PCT/AU2025/050129
Authority: WO
Inventors: Bruce Halcro Candy
Original assignee: Minelab Electronics Pty Ltd
Current assignee: Minelab Electronics Pty Ltd
Priority date: 2024-02-16
Filing date: 2025-02-17
Publication date: 2025-08-21
Anticipated expiration: 2026-08-16

Abstract

A method of reducing a transmit back-EMF decay period of a pulse induction metal detector, the method comprising: connecting a diode to a first inductive winding within a coil housing, the inductive winding at least for transmitting a magnetic field; wherein the pulse induction metal detector is arranged and configured such that the diode is forward biased during some of the transmit back-EMF period, and the diode is reverse biased during non-transmit periods of the inductive winding, the non-transmit periods comprise a period to process a receive signal of the pulse induction metal detector.

Description

AN IMPROVED METAL DETECTOR

TECHNICAL FIELD

[0001] The present disclosure relates to a metal detector.

BACKGROUND

[0002] “Pulse Induction” (PI) metal detectors transmit a waveform commencing with a low voltage period when a low voltage (e.g. -10 V, -5 V, 4 V etc.) is applied across a “transmit inductive winding” (or windings), that results in an increasing current flowing in the inductive winding during the low voltage period that in turn generates a correspondingly increasing transmitted magnetic field, and, thereafter the applied low voltage to the inductive winding during the low voltage period is switched off, and the energy stored in the transmitted magnetic field induces a “back-EMF” in the inductive winding, which is sometimes clamped to a high voltage (e.g. +180 V, +120 V, +100 V etc.). Once this transmitted magnetic energy is nearly exhausted, the back-EMF “collapses” or “decays” down to zero during a “back-EMF decay period”.

[0003] The transmit inductive winding is housed within a coil housing, and this transmit inductive winding is connected to an electronics housing by a cable, also known as a “coil cable”, that usually contains at least a coaxial cable, through which signals are sent.

[0004] When the so called “coil” (the inductive windings plus coil housing) is a “monoloop”, it usually contains just a single inductive winding which acts to both transmit the transmitted magnetic field and also to receive a receive magnetic field.

[0005] This receive magnetic field from an environment into which the transmitted magnetic field has been applied, induces a receive EMF in the said monoloop inductive winding. This is amplified and processed by the metal detector electronics housed within the electronics housing, wherein the processing includes producing a processed signal that is applied to an output indicator, that is usually in the form of modulated audio. Further, the processing may include processes known to those skilled in the art such as “ground balancing”.

[0006] Metal detectors rely on the detection of changing magnetic fields generated by eddy currents flowing in metal targets that result from EMFs induced in these metal targets via application of a changing transmitted magnetic field.

[0007] These metal targets may be modelled as a distributed resistance-inductive network, and for reasons of simple intuitive understanding to assist those skilled in the art, often as just a single first-order network of a single inductance in parallel with a single resistor, giving a time-constant (TC) of L/R where R is the resistor value and L the inductance value. Very small nuggets (e.g. 40 milligrams) are an example of a short (or a “fast”) TC metal target, whereas say a 10 kg copper ingot is an example of a target with a very considerably longer TC (albeit the model of such a target is overtly a high-order distributed inductive-resistive target).

[0008] One aim of some fit-for-purpose metal detectors designed for gold nugget prospecting is to endeavour to detect the smallest nuggets whilst this signal is not contaminated by a spurious signal from the back-EMF decay. This disclosure provides an alternative method to allow a very rapid period of transmit inductive back-EMF decay using relatively few components.

SUMMARY

[0009] According to a first aspect of the present disclosure, there is provided a method of reducing a pulse induction metal detector transmit back-EMF decay period by including a diode within a coil housing that is connected to a first inductive winding that acts at least to transmit a magnetic field, wherein the diode is forward biased during some of the back-EMF period, and reverse biased during nontransmit periods that include receive synchronous demodulation signal processing.

[0010] According to a second aspect of the present disclosure, there is provided a method of reducing a transmit back-EMF decay period of a pulse induction metal detector, the method comprising: connecting a diode to a first inductive winding within a coil housing, the inductive winding at least for transmitting a magnetic field; wherein the pulse induction metal detector is arranged and configured such that the diode is forward biased during some of the transmit back-EMF period, and the diode is reverse biased during non-transmit periods of the inductive winding, the non-transmit periods comprise a period to process a receive signal of the pulse induction metal detector.

[0011] According to a third aspect of the present disclosure, there is provided a pulse induction metal detector, comprising: a first inductive winding within a coil housing, the inductive winding for at least transmitting a magnetic field; a diode connected to a first inductive winding; wherein the diode is arranged and configmed such that the diode is forward biased during some of the transmit back-EMF period, and the diode is reverse biased during non-transmit periods of the inductive winding to reduce a transmit back-EMF decay period of the pulse induction metal detector, wherein the non-transmit periods comprise a period to process a receive signal of the pulse induction metal detector.

[0012] In one form, a first low capacitance coaxial cable within a coil cable is connected between the first inductive winding and receive electronics within an electronics housing of the pulse induction metal detector. In one form, the first low capacitance coaxial cable is connected to taps of the first inductive winding. In one form, the first receive inductive winding within the coil housing is connected to a receive preamplifier housed within the coil housing and an output of the preamplifier is fed to further electronics within the electronics housing via the coil cable.

BRIEF DESCRIPTION OF DRAWINGS

[0013] Embodiments of the present disclosure will be discussed with reference to the accompanying drawings wherein:

[0014] Figure 1 depicts an exemplary block diagram of a PI metal detector;

[0015] Figure 2 shows exemplary PI waveforms, and (a) when associated with example of Figure 1 and (b) when associated with embodiment of Figure 4;

[0016] Figure 3 depicts an impulse response for a first-order “idealized target” exponential decay of a very short time constant target;

[0017] Figure 4 depicts one embodiment of the present disclosure; and

[0018] Figure 5 depicts another embodiment of the present disclosure.

DESCRIPTION OF EMBODIMENTS

[0019] Figure 1 shows an exemplary block diagram of a PI metal detector. The electronics within the electronics housing 1 is connected to the inductive winding 6 within the coil housing 2 via a coil cable 3. In this example, the coil is a monoloop, wherein it consists of just one inductive winding 6 that acts to transmit a transmitted magnetic field and to receive a receive magnetic field. Other forms of coil are possible. The inductive winding 6 is shielded electrostatically by a relatively poorly conducting electrostatic shield 9 surrounding most of the inductive winding 6. This is connected to the electronics system “ground” 10 via an outer shield 5 of a coaxial cable within the coil cable 3. An inner core 4 of the coaxial cable is shielded by the outer shield 5. The inner core 4 of the coaxial cable is connected to one end 7 of the inductive winding 6, and outer shield 5 of the coaxial cable is connected to the other end 8 of the inductive winding 6. The metal detector electronics includes a clock 11, that generates a transmitter control signal at the gate 12 of a field effect transistor 13 (FET), whose source 14 is connected to a low voltage source 15, and this low voltage source 15 is connected to the electronics system ground 10. The drain 16 of FET 13 is connected to two diodes, diode 17 and diode 19. The cathode of diode 19 is connected to a voltage source 21 that clamp (limit) the back-EMF to approximately the EMF of voltage source 21. Figure 1 shows the EMF of voltage source 21 as being, for example, 180 V and that of voltage source 15 as being, for example, 10 V, but these may be different values. Diode 17 is connected to the inner core 4 of the coaxial cable in the coil cable. Within the electronics housing, the inner core 4 of the coaxial cable is also connected to a resistor 30 and a transmit/receive switch 22 (T/R switch) through connection 18. The other end of resistor 30 is connected to the electronics system ground 10.

[0020] The parallel network of inductance of inductive winding 6, and the capacitance of the coaxial cable, plus that of the T/R switch, plus that of the effective self-capacitance of inductive winding 6, plus that of diode 17, plus the effective capacitance between inductive winding 6 and the electrostatic shield 9, and, the resistance of resistor 30, is usually designed to form a critically damped parallel “LCR” network. This occurs when the following mathematically equal conditions are satisfied: where the undamped resonant frequency in radians per second is:

[0021] An output 24 of the T/R switch 22 is connected to an input of a receive preamplifier 27, whose output 28 feeds inputs of synchronous demodulators 30 with receive processing electronics 25. Outputs 35 of the synchronous demodulators 30 feed further signal processing electronics 36, that includes signal processing such as low and high-pass filtering, ground balancing, indicator output modulation and other processors known to people skilled in the art. The indicator output 37 is usually modulated audio and usually supplied to headphones or loudspeakers. The control 26 of the synchronous demodulators 30 is generated by the clock 11, and clock 11 may also control 38 signal processing within the further signal processing electronics 36, and control 23 the T/R switch 22. The reason for the diode 17 is that its capacitance is typically much lower than that of the drain (relative to ground 10) when the drain voltage is approximately the same as the voltage of voltage source 15 (e.g. 10 V) relative to the source 14 of the FET 13. This is especially true when a high valued reverse high voltage is held across diode 17. For example, an ES2D diode at about 180 V has a capacitance of about 8 pF, whereas, for example, an Infineon FET IPx320N20N3 (drain) has a considerably higher capacitance of about 1.3 nF at 10 V.

[0022] Figure 2 shows exemplary PI waveforms associated with Figure 1.

[0023] With reference to Figure 2, the gate 12 control of FET 13 is shown as graph 200, 202A, 202B, 204. During a receive period, that is a non-transmit period, the gate voltage 200 is held at a similar voltage to the source voltage of FET 13. This ensures that FET 13 is “turned off’. During a transmit low voltage period between times 201 and 203 , the gate of FET 13 is held at a voltage 202 A and 202B substantially higher than the gate threshold, in this example, 10 V relative to the source voltage, to ensure that the FET is “turned on”. This means that the drain 16 is approximated “short-circuited” to the source (at -10 V) during this transmit low voltage period. The current flowing between the ground 10, through the inner core 4 and outer shield 5 of the coaxial cable, the inductive winding 6, the power supply 15, the FET 13 and diode 17 is known to those skilled in the art and approximately follows

. (y - V_diode l - e- ^tL^r) (V - V_diode)

¹ = - r ¹ - R - - (0

[0024] At the termination of this period, where V is the voltage of the voltage source 15, V_diode is the forward voltage drop across diode 17 (varies between about 0 V and 1 V and is a function if current i), “r” is the equivalent series resistance of all the elements in the above said transmit circuitry, and t=0 at the commencement (time 201) of the transmit low voltage period between times 201 and 203, and P is the period of the low voltage period. Note that the forward voltage drop across diode 17 is non-linear. Thus, during this transmit low voltage period, the current flowing through the inductive winding 6 increases as per equation (i) and hence this generates a transmitted magnetic field into the searched environment of approximately the same form as equation (i). This is not exactly the same mainly because the (complex) magnetic permeability of the environment alters the magnetic field, due to elements such as soil viscous remnant magnetism (VRM), soil (eddy current) conductivity, conductivity of metals (generating eddy currents) in sought metal targets, but also due to eddy currents flowing in the coil cable connector metals, electronics PCB and so on. With reference to Figure 2, the EMF 210 across the inductive winding 6 is at approximately 0 V (relative to the electronics ground 10), save induced EMF generated by received rate of change of environmental magnetic fields. At the commencement of the transmit low voltage period at a time after time 201, an EMF 211 A close to the EMF of voltage source 15 appears across the inductive winding 6, whereas at the end of the low voltage transmit period just before time 203, the EMF 21 IB across the transmit inductive winding is that of the voltage source 15 less that of the product of sum of the resistances of conductors, FET 13, the ESR of voltage source 15, multiplied by the current flowing through the inductive winding 6, and also less that of the forward voltage across diode 17 at time 203.

[0025] After the termination of the transmit low voltage period at time 203, FET 13 is switched off again when its gate voltage 204 returns to approximately its source voltage, and a back-EMF high voltage period commences. This is limited (or “clamped”) to approximately the voltage of voltage source 21 (e.g. 180 V) via the forward action of diode 19. The energy supplying this back-EMF period is due to the transmitted magnetic field generated during the transmit low voltage period. It is known to those skilled in the art that during this clamped back-EMF period the current flowing through inductive winding 6 decreases approximately linearly as

, _c[_ampt _z...,

1 = Io - — (ill) where V_ciamp is the said clamped voltage (e.g. 180 V), and I_o the inductive winding 6 current at time 203, and t=0 is defined here at time 203. The clamped back-EMF period terminates at time 205 when the current flowing through inductive winding 6 equals V_ciamp/R (where R is the value of resistor 30 as stated above). At time 205 the back-EMF decay period commences and is of principal significance to this disclosure. It is known to those skilled in the art that this decays as where t=0 at the termination of the clamped back-EMF (V_ciamp), at time 205. Note that the late-time decay period is dominated by the term.

[0026] With reference to Figure 2, only the “top” of clamped back-EMF 212 is shown above the “break” in the waveform trace so as to show clearly features of the lower voltages 214 of the decaying back EMF, and this is yet further exaggerated in vertical scale for the purposes of explaining the detection issues to be overcome when endeavouring to detect very fast time constant targets. The graphs of Figure 2 assume no detected metal targets, nor any signals from soils. As can be seen, the back-EMF becomes near zero at about time 216. Only after this can synchronous demodulation commence without too much contamination of the back-EMF. The back-EMF shape of (iv) is exaggerated as 214. After time 216 the back EMF decay 215 continues but is effectively zero. Note the initial “rounded” commencement of the back-EMF decay with an initial slope of zero because = 0 when t=0.

[0027] The EMF at the drain 16 of FET 13, or more particularly, at the cathode of diode 17 is approximately that of the voltage source 15 during the transmit low voltage period 221 A and 22 IB when the FET 13 is “switched on”. The length of the low voltage period is also indicated via the waveform “breaks” to be longer than shown in order that details of the back-EMF period may be more clearly seen. During the clamped back-EMF, the EMF at this node between FET 13 drain 16, the cathode of diode 17 and anode of diode 19 equals approximately the EMF 212 of voltage supply 21, e.g. 180 V as shown, + the forward voltage drop across diode 19. Thereafter, the relative high capacitance of the drain 16 of FET 13 then maintains this EMF 235 and EMF 230 at the cathode of diode 17. Hence during the non-transmit periods, diode 17 is reverse biased at about 180 V, yielding its very low capacitive contribution to the transmit inductive associated circuitry. [0028] Eddy currents are induced in a metal target by the transmitted magnetic field. For short TC targets, most of the eddy currents are generated due to the back-EMF, because by the time receive synchronous demodulation commences, most of the eddy currents induced during the transmit low voltage period have decayed too considerably to be of any significance, and, the induced EMF in the target is considerably larger during the back-EMF period than during the low voltage transmit period. Thus, the eddy currents induced in a (very) short time constant target during the clamped back-EMF period, may be approximated as where T = , the late-time TC of the metal target with I being its effective eddy current loop inductance and p its resistance, and V_hv the effective induced EMF within the said metal target, and t=0 when the clamped back-EMF periods begin (at time 203 in Figure 2). At the end of the clamped back=EMF period (time 205 in Figure 1), the eddy current i_t is approximately where A is the width of the clamped back-EMF period (time 205 - time 203).

[0029] After the clamped back-EMF period (after time 205), the back-EMF decay period 213 and 214 continues contributing energy to the target eddy currents, and, its rate of change of transmitted magnetic field is proportional to equation (iv) above. After this back-EMF decay period effectively ceases (at time 216), the target eddy currents decay without any transmitted magnetic field energy left to affect them.

[0030] The rate of change of magnetic fields generated by the target eddy currents induces an EMF (typically very small for very small metal targets: microvolts or millivolts) in the inductive winding 6. Once the T/R switch 27 switches from “transmit” to “receive”, inductive winding 6 changes roles to act as a receive inductive winding. This switch-over occurs close to time 216, usually before. The induced EMF from the target eddy currents is convolved with the associated LCR network transfer function of the inductive winding 6.

[0031] In the mathematics below the induced EMF is arbitrarily normalized to commence at a value of one for a given idealized first-order time constant target when a> is idealized to zero. [0032] The back-EMF decay period begins when t=0, and the receive induced EMF from the idealized target rate of change of eddy currents only (excluding contributions the decaying back-EMF of inductor 6) is 2(fi - c ) 6(H - m) where fl = - .

[0033] The integrated induced receive voltage for (vii) between time and t₂ is proportional to 6co(fl — co)

[0034] Note: it is assumed that the energy dissipated in the of the metal target does not “load” the decaying transmitted magnetic field during the back-EMF decay period significantly (via mutual coupling)

[0035] A typical fit-for-purpose high-end gold nugget detector may have transmit inductive winding inductance of 300 pH, and the total parallel effective capacitance may be about 300 pF, making co approximately 3.33 megaradians per second.

[0036] With reference to Figure 3, this shows the impulse response for the said first-order “idealized target” exponential decay 300 of a very short time constant target (relative to PI, or Pl-like metal detector detection capability). Its TC shown is (arbitrarily) 1.43 microseconds with fl being 700 kiloradians per second. The time is defined as zero (t=0) at the boundary between the clamped back-EMF and back-EMF decay period, namely time 205 of Figure 2. The scales are relatively arbitrary; the signal levels entirely depend on where the target is relative to inductive winding 6, and the shape and size of the metal target. Figure 3 therefore implies that the idealized target’s induced impulse signal, is arbitrarily about 25 microvolts when the Figure 3’s graph X-axis arbitrarily begins at t=1.5 microseconds (not 1 volt at t=0 as above to define equations (vii) and (viii)). Shown are responses of three different values of a> One col is 3.333 megaradians per second as in the above “typical” LCR example, another u>3 is three times higher at 10 megaradians per second, and a third u>2 at the geometric mean between them, namely 5.77 megaradians per second. With no targets being detected and ignoring any transients due to the T/R switch being switched from “transmit” to “receive”, the back-EMF decays (starting at 180 V at t=0) are graphs 302, 304 and 306 for u>3, u>2, and <ul respectively. These follow equation (iv), and the received EMF induced in inductive winding 6 from the “idealized” target (200) plus the decaying back-EMFs are graphs 301, 303, and 305 respectively. If demodulation commences once the back-EMF has decayed down to a very low level as indicated at 310, 313 and 315 respectively for u>3, u>2, and <ul, then as can be indicated in Figure 3, the receive signal magnitudes commence demodulation as indicated at 301, 303 and 305 respectively. Note the very substantial advantage in signal level of the higher values of compared to lower for the detection of this short TC target. Thus increasing m, by decreasing the effectively parallel capacitance (C) whilst maintaining the same inductance (L), or decreasing the effectively parallel inductance (L) whilst maintaining the same capacitance (C), or decreasing both, substantially improves the detection capability of fast decaying eddy current targets. Note than no electronics noise is shown, which is about several microvolts RMS for the noise and preamplifier bandwidths of PI metal detectors, nor is the transfer function of the preamplifier included. Note that the gains of the preamplifiers in PI metal detectors are of the order of about 100, and the transmit waveform fundamental repeating periods usually less than 1 millisecond (e.g. may be 250 microseconds), and the transmit waveform may include multi-periods. Depending on the demodulation function, typically the processed signal from the idealised target (300) would be below the noise floor for the nil and u>2 cases, but not for u>3 for a typical well designed high-end metal detector.

[0037] This disclosure provides a method to improve the detection of fast time constant targets.

[0038] Methods for shortening this back-EMF decay period are given in two patents, US9348053 and US9829598, that usefully include T/R switches. This present disclosure may be implemented together with those methods, but may also be implemented independently from those methods.

[0039] Figure 4 shows an exemplary first embodiment of the present disclosure. It should be noted that the structure of Figure 4 is like the structure of Figure 1. The most significant changes for this embodiment are that diode 17 is now housed within the coil housing 402 as diode 417, rather than residing in the electronics housing 401.

[0040] A second coaxial cable 431, 432, a “receive coaxial cable”, has a capacitance designed to be of lower capacitance than that of the coaxial cable 4, 5 in Figure 1 for a similar diameter. This is possible because the coaxial cable in Figure 1 must be of total low resistance (ideally « 1 Ohm) in order to maintain a reasonable “Q” of the transmit series resistance circuitry and to avoid excessive power loss, whereas the receive coaxial cable 431, 432 in Figure 4 has no such constraint, and therefore can allow for a series resistance of at least an order of magnitude higher, thus providing much lower capacitance for a given diameter. Further, the capacitance of the “transmit coaxial cable” 404, 405, need not be of low capacitance, and indeed need not actually be a coaxial cable, but merely separate cables.

[0041] “Receive coaxial cable” 431, 432 may be connected to taps of the monoloop inductive winding 446, 447, 450, 451, or these inductive windings may be separated but well inductively coupled. For example, the central core 432 of the “receive coaxial cable” may be connected 455 to the inductive windings 447 and 450 node at 460, or, connected 457 to the inductive windings 447 and 446 node 448. Assuming that these taps are by quarters of the net inductive winding (! , !/, %), then the capacitive loading by the receive coaxial cable 431, 432 as seen at the “live” end 407 of the whole inductive winding is

^107 ~ ® ^coax where a is the tapped ratio of the whole inductive winding 446, 447, 450, 451. Thus, for example a selected tap, to connect the coaxial cable to, may be 457 (at ³A), and hence the coaxial cable 431, 432 capacitive loading seen at 407 is 9/16 = 56% and the attenuation by definition approximately 3/4. These tap ratios may be other than ! , !/₂, ³/4.

[0042] By placing diode 17 of Figure 1 within the coil housing, shown as diode 417 in Figure 4, means that there is no “transmit coaxial cable” relatively high capacitive loading in parallel with the transmit windings 446 and 447, only that of diode 417 which, as said, is very low, plus the self-capacitance of the windings 446, 447, 450, 451, plus the capacitance relative to the electrostatic shield 409, plus the transformed capacitance of the lower capacitance “receive coaxial cable” 431, 432. This will allow for a lower effective parallel capacitance of several times lower than that of Figure 1, all else being equal, bar signal attenuation due to the tap ratio. Another reason why the monoloop winding is split is given in US 11474274. Thus resistors 449, 429, 452, 454 respectively are in parallel with (transmit/receive) inductive windings 446, 447, 450 and 451 respectively act both to damp the fundamental frequency (see first equation above) and also higher-order internal resonances. Not all of the damping need be achieved by resistors 454, 452, 429 and 449, but more may be included between the T/R switch 427 and preamplifier 430 input 424. As said, the T/R switch 422 may include methods described in US9348053 and US9829598 to yet further speed up the back-EMF decay periods whilst have the advantage of a higher frequency resonance (co) receive LCR input.

[0043] Diode 417 (plus any associated PCB) is surrounded by a ferrite magnetic shield 445 to avoid induced eddy currents from the transmitted magnetic field being detected by the “receive inductive windings”. The resistors 449, 429 may also be magnetically shielded thus. The coil cable 403 houses conductor 433 that is connected to the electrostatic shield 409. This shield 409 also may be connected to the receive coaxial cable shield. [0044] Note that merely moving the “capacitance isolating diode” (417) into the coil housing, and connecting a “low capacitance receive coaxial cable” across all of the inductive windings (446, 447, 450, 451), which thus requires at least cable 404, 405, achieves little for the same coil cable (403) diameter as the prior art in Figure 1: This is because, cable 404, 405 must be included with in the said diameter, which must then force a smaller “receive coaxial cable” thereby increasing its capacitance closer to the capacitance of the coaxial cable of Figure 1, and thus gaining little for a given coil cable diameter. Hence, in essence, it is the “receive” tapping of the transmit inductive windings that achieves the major advantage, for a given coil cable diameter compared with the prior art.

[0045] Further, note that the tapping of the inductive winding for the receive signal attenuates the receive signal compared to the prior art. However, in practice, the soil magnetic and conductivity “ground noise” is usually significantly greater than the receive electronics noise for the state-of-the-art high-end “gold detectors”, and hence at least for searching for gold nuggets, this loss in gain is typically of little consequence in practice. However, even if this were not so, higher values of co afford the advantages of detecting faster TC targets without the attendant increase on “X” spurious signal contamination (that would be the case without increasing co).

[0046] With reference to Figure 2, the EMFs 220, 221A, 221B, 222, 223, 224, 225 for Figure 4 play the same role as that of EMFs 210, 211A, 21 IB, 212, 213, 214, 215 for Figure 1, but the back-EMF decay 223, 224 is shown as more rapid for the Figure 4 waveforms than the back-EMF decay 213, 214 for Figure 1. Note the difference in time duration 217, between times 205 and 216, and time duration 227 between times 205 and 226.

[0047] A second exemplary embodiment is given in Figure 5. This configuration is for a coil housing a separate nulled transmitter winding and receiver winding inductors, and the preamplifier 427 of Figure 4 is now housed within the coil housing 552 as preamplifier 577. It needs to be supplied with power from the control box 551. This is achieved by voltage source 593 supplying one voltage, e.g. -5 V via a conductor 595 within the electronics housing 551, through conductor 542 within the coil cable 553, and via conductor 591 within the coil housing 552, and, voltage source 592 supplying another voltage, e.g. +5 V via a conductor 596 within the electronics housing 551, through conductor 544 within the coil cable 553, and via conductor 590 within the coil housing 552. However, the negative supply voltage may be obtained by the simple use of a small signal diode rectifying the negative low voltage period plus a negative preamplifier supply storage capacitor within the coil housing. The input of the preamplifier 577 is connected to one end of receive inductive winding 585, and the electronics ground 560 is connected to an end of receive inductive winding 586 via conductor 541 within the coil cable 553. Receive inductive winding 586 is connected in series with receive inductive winding 585. Resistors 588 and 587 are selected to affect effective damping of the receive inductive winding wherein its effective parallel capacitance is due to the self-capacitance of each receive inductive winding, and the capacitance between them, plus the capacitances relative to the electrostatic shield 559 and to transmit inductive windings 580 and 581. The reason why the inductive receive winding is split, or, centre-tapped, and ditto for the transmit inductive winding, again is given in US11474274. As suggested in US11474274, these windings may have further taps such as quarter taps to reduce higher order internal resonances. (Note the term “critical damping” only applies to second order system, not to coupled coils such as 586 and 585, or, 580 and 581, which obey different equations by definition.) This arrangement of the preamplifier within the coil housing then removes the loading capacitance of the coil cable’s receive coaxial cable (such as 431, 432 in Figure 4) from the receive inductive circuit, thus providing for a rapid receiver impulse response damping decay and enabling the detection of shorter time constant targets than the capability of the system of Figure 4. Both diode 546 and preamplifier 577 are magnetically shielded by ferrites 545 and 573 respectively, which may be the same ferrite, to avoid the transmitted magnetic field inducing eddy currents in the associated metal such as PCB tracks and solder joints being detected by the receive inductive windings. The electronics ground 558 within the coil housing connects the electrostatic shield 559 and is connected to the electronics ground 560 via the conductor 541 within the coil cable 553. An output 578 of preamplifier 577 is fed to inputs 572 of the synchronous demodulators 597 via conductors 543 within the coil cable 553.

[0048] However, it is pointless designing such a high valued a> receive LCR network if the transmit effective parallel LCR network is as “slow” as the prior art (Figure 1). Again, this limiting short-coming may be overcome by housing diode 546 within the coil housing 552, that isolates the transmit LCR network from both the high capacitance of the cables (554, 555) feeding the transmit inductive windings (580 and 581 plus their respective attendant damping resistors 582 and 583), plus isolation from the high drain capacitance of FET 563.

[0049] Table 1 shows ratios of synchronous demodulation outputs for integrations between the times shown relative to the integration for a> = 3.333 megaradians per second, for fl = 700 kiloradians per second.

TABLE 1

[0050] The initial start time of the demodulation periods, 7.2 microseconds for =3.333 megaradians per second, 4.1 microseconds for a> =5.771 megaradians per second, and 2.3 microseconds for a> =10 megaradians per second correspond to times 315, 313 and 310 respectively in Figure 3. The graph labels 305, 303 and 301 are placed at approximately these same respective times to emphasize differences in magnitude advantages for higher values of u>. Note that the higher group delays do aid the lower values of for their given demodulation time periods, for example as shown by the values 0.52 and 0.65 in Table

1, but the absolute value improvements are greater for the higher values of m, especially say 16 times improvement for the 3 times higher value of a> =10 megaradians per second compared to 3.333 megaradians per second.

[0051] Table 2 provides ratios between the integrated residual back-EMF decay divided by the target integrated signal for the levels assumed above. This indicates that for example the percentage of residual back-EMF decay for a> = 3.333 megaradians per second is not insignificant (6%) for the example target signal.

TABLE 2

[0052] Table 3 indicates the percentage change of the ratios in Table 2 for a 1% change in L. L changes when the soil magnetic permeability modulates the inductance of the coil inductive windings as the coil is swept across magnetic soils. This modulation changes the values of a> (by about 0.5% for a 1% change in L), but Table 3 headings do not show these altered values for the sake of continuity. This data indicates that changes in these ratios are substantially sensitive to changes as the coil is swept across magnetic soils. Further, equations ii and iii show that the clamped EMF period duration is modulated by the transmit inductive winding modulation L as well. This may be compensated for, but either way, these outcomes stress the need to avoid commencement of synchronous demodulation before the back-EMF period has effectively died down to significantly low levels.

TABLE 3 [0053] With reference to Figures 5 and 4, the system clocks 561 and 411 control the gate 562 of FET 563, and gate 412 of FET 413 whose sources 564 and 414 are connected to power supplies 565 and 415 respectively, and both power supplies 565 and 415 are connected to the electronics grounds 560 and 410 respectively, as are the clocks 561 and 411 too. In Figure 5, drain 566 of FET 563 is connected to diode 569 that is connected to power source 571, that is also connected to the electronics ground 560, and in Figure 4, drain 416 of FET 413 is connected to diode 419 that is connected to power source 421, that is also connected to the electronics ground 410. These power sources are shown to be 180 V, for example, and act to limit or clamp the back -EMF to 180 V. In Figure 5, drain 566 of FET 563 is also connected to the central conductor 554 of a coaxial cable within the coil cable 553 , and in Figure 4 drain 416 of FET 413 is also connected to the central conductor 404 of a coaxial cable within the coil cable 403. In Figure 5, shield 555 of its coaxial cable is connected to the electronics ground 560 at one end, and the series pair of transmit inductive windings 580 plus 581 at the other end within the coil housing 552, while in Figure 4 the shield 405 of its coaxial cable is connected to the electronics ground 410 at one end, and the series pair of transmit inductive windings 447 plus 446 at the other end within the coil housing 402. In Figure 5, central core 554 of the “transmit coaxial cable” within the coil cable 553 terminates within the coil housing 552 as a connection to a diode 546 housed with the coil housing 552. However, the transmit cables 554 and 555 may be in the form of independent cables and not in the form of a coaxial cable. In Figure 4, central core 404 of the “transmit coaxial cable” within the coil cable 403 terminates within the coil housing 402 as a connection to a diode 417 housed with the coil housing 402. An output 424 of the T/R switch 422 is connected to an input of a receive preamplifier 427, whose output 428 feeds inputs of synchronous demodulators 430 within the receive processing electronics 425. Clock 411 output 423 controls the T/R switch 422. Referring to both Figures 5 and 4, the forward gains of synchronous demodulators 597, 430 are controlled by clocks 561 and 411 respectively, via busses 576 and 426 respectively. In Figure 4, outputs 435 of the synchronous demodulators 430 feed further signal processing electronics 436, that includes signal processing such as low and high-pass filtering, ground balancing, indicator output modulation and other processors known to people skilled in the art. The indicator output 437 is usually modulated audio and supplied usually to headphones or loudspeakers. Similarly, in Figure 5, the output 578, 572 of preamplifier 577 feeds inputs 572 of synchronous demodulators 597 within the receive processing electronics 575 via conductor 543 within the coil cable 553. Outputs 598 of the synchronous demodulators 597 feed further signal processing electronics 547, that includes signal processing such as low and high-pass filtering, ground balancing, indicator output modulation and other processors known to people skilled in the art. The indicator output 549 is usually modulated audio and supplied usually to headphones or loudspeakers. Referring to both Figures 5 and 4, clocks 561 and 411 may control 548, 438 signal processing within the further signal processing electronics 547 and 436 respectively.

[0054] In this specification, the terms “ground” and “soil” are used interchangeably. As understood by a person skilled in the art, the terms “ground” and “soil” mean surfaces of earth where targets may be contained within. The surfaces are often solid, may be homogenous or may be a combination of various soil types, and may contain moisture or water. In a circuit, however, “ground” refers to electronics ground.

[0055] Those of skill in the art would understand that information and signals may be represented using any of a variety of technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips referenced throughout the above description may be represented by voltages, currents, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination thereof.

[0056] Those of skill in the art would further appreciate that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the embodiments disclosed herein may be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative components, blocks, modules, circuits, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the present disclosure.

[0057] The steps of a method or algorithm described in connection with the embodiments disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. For a hardware implementation, processing may be implemented within one or more application specific integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs), field programmable gate arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, other electronic units designed to perform the functions described herein, or a combination thereof. Software modules, also known as computer programs, computer codes, or instructions, may contain a number of source code or object code segments or instructions, and may reside in any computer readable medium such as a RAM memory, flash memory, ROM memory, EPROM memory, registers, hard disk, a removable disk, a CD-ROM, a DVD-ROM or any other form of computer readable medium. In the alternative, the computer readable medium may be integral to the processor. The processor and the computer readable medium may reside in an ASIC or related device. The software codes may be stored in a memory unit and executed by a processor. The memory unit may be implemented within the processor or external to the processor, in which case it can be communicatively coupled to the processor via various means as is known in the art.

[0058] Throughout the specification and the claims that follow, unless the context requires otherwise, the words “comprise” and “include” and variations such as “comprising” and “including” will be understood to imply the inclusion of a stated integer or group of integers, but not the exclusion of any other integer or group of integers.

[0059] The reference to any prior art in this specification is not, and should not be taken as, an acknowledgement of any form of suggestion that such prior art forms part of the common general knowledge.

[0060] It will be appreciated by those skilled in the art that the disclosure is not restricted in its use to the particular application described. Neither is the present disclosure restricted in its preferred embodiment with regard to the particular elements and/or features described or depicted herein. It will be appreciated that the disclosure is not limited to the embodiment or embodiments disclosed, but is capable of numerous rearrangements, modifications and substitutions without departing from the scope of the disclosure as set forth and defined by the following claims.

Claims

1. A method of reducing a pulse induction metal detector transmit back-EMF decay period by including a diode within a coil housing that is connected to a first inductive winding that acts at least to transmit a magnetic field, wherein the diode is forward biased during some of the back-EMF period, and reverse biased during non-transmit periods that include receive synchronous demodulation signal processing.

2. The method of claim 1, wherein a first low capacitance coaxial cable within a coil cable is connected between the first inductive winding and receive electronics within an electronics housing of the pulse induction metal detector.

3. The method of claim 2, wherein the first low capacitance coaxial cable is connected to taps of the first inductive winding.

4. The method of claim 1, wherein a first receive inductive winding within the coil housing is connected to a receive preamplifier housed within the coil housing and an output of the preamplifier is fed to further electronics within the electronics housing via the coil cable.

5. A method of reducing a transmit back-EMF decay period of a pulse induction metal detector, the method comprising: connecting a diode to a first inductive winding within a coil housing, the inductive winding at least for transmitting a magnetic field; wherein the pulse induction metal detector is arranged and configured such that the diode is forward biased during some of the transmit back-EMF period, and the diode is reverse biased during non-transmit periods of the inductive winding, the non-transmit periods comprise a period to process a receive signal of the pulse induction metal detector.

6. The method of claim 5, wherein a first low capacitance coaxial cable within a coil cable is connected between the first inductive winding and receive electronics within an electronics housing of the pulse induction metal detector.

7. The method of claim 6, wherein the first low capacitance coaxial cable is connected to taps of the first inductive winding.

8. The method of claim 5, wherein a first receive inductive winding within the coil housing is connected to a receive preamplifier housed within the coil housing and an output of the preamplifier is fed to further electronics within the electronics housing via the coil cable.

9. A pulse induction metal detector, comprising: a first inductive winding within a coil housing, the inductive winding for at least transmitting a magnetic field; a diode connected to a first inductive winding; wherein the diode is arranged and configured such that the diode is forward biased during some of the transmit back-EMF period, and the diode is reverse biased during non-transmit periods of the inductive winding to reduce a transmit back-EMF decay period of the pulse induction metal detector, wherein the non-transmit periods comprise a period to process a receive signal of the pulse induction metal detector.

10. The pulse induction metal detector of claim 9, wherein a first low capacitance coaxial cable within a coil cable is connected between the first inductive winding and receive electronics within an electronics housing of the pulse induction metal detector.

11. The method of claim 10, wherein the first low capacitance coaxial cable is connected to taps of the first inductive winding.

12. The pulse induction metal detector of claim 9, wherein a first receive inductive winding within the coil housing is connected to a receive preamplifier housed within the coil housing and an output of the preamplifier is fed to further electronics within the electronics housing via the coil cable.