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WO2025068755A1 - Circuit amplificateur - Google Patents

Circuit amplificateur Download PDF

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Publication number
WO2025068755A1
WO2025068755A1 PCT/IB2023/059808 IB2023059808W WO2025068755A1 WO 2025068755 A1 WO2025068755 A1 WO 2025068755A1 IB 2023059808 W IB2023059808 W IB 2023059808W WO 2025068755 A1 WO2025068755 A1 WO 2025068755A1
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WO
WIPO (PCT)
Prior art keywords
amplifier circuit
transmission lines
phase
resistors
amplifier
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Pending
Application number
PCT/IB2023/059808
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English (en)
Inventor
Richard Hellberg
Tony FONDÉN
Rui Hou
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Telefonaktiebolaget LM Ericsson AB
Original Assignee
Telefonaktiebolaget LM Ericsson AB
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Priority to PCT/IB2023/059808 priority Critical patent/WO2025068755A1/fr
Publication of WO2025068755A1 publication Critical patent/WO2025068755A1/fr
Pending legal-status Critical Current
Anticipated expiration legal-status Critical

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/60—Amplifiers in which coupling networks have distributed constants, e.g. with waveguide resonators
    • H03F3/602—Combinations of several amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/56—Modifications of input or output impedances, not otherwise provided for
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/451—Indexing scheme relating to amplifiers the amplifier being a radio frequency amplifier
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
    • H03H7/48—Networks for connecting several sources or loads, working on the same frequency or frequency band, to a common load or source

Definitions

  • Embodiments of the present disclosure relate to amplifier circuits, and particularly to amplifier circuits for reducing the effects of reverse waves in the amplifier circuit interfering with signals output from the amplifiers.
  • Power amplifiers (PA) in radio systems can be negatively affected by reverse waves (also referred to herein as interfering waves) which occur in a radio system when signal energy travels backwards in an amplifier (e.g., into an antenna branch and towards the amplifier).
  • Reverse waves can be caused by reflections from impedance discontinuities in transmission lines (e.g., from contacts, filters, and antennas) and/or they can enter the circuit as external interference from other (uncoordinated) radios.
  • the negative effects of reverse waves reaching amplifiers include: fluctuating power capability; fluctuating gain; and increased distortion due to reverse intermodulation (IM).
  • intermodulation can occur between the desired amplifier output and the reverse/interfering wave.
  • isolators and/or circulators are commonly used for improving backwards termination of reverse waves (including antenna reflections).
  • these comprise: identical amplifiers connected to two ports of a quadrature hybrid coupler, that also has an antenna port and a resistively terminated port.
  • the antenna port is where the combined power of the amplifiers is output.
  • the terminated port is where reverse waves that are linearly and equally reflected by the two amplifiers end up, instead of being reflected and going back to the antenna port.
  • Balanced amplifiers can have the side effect of steering many reverse intermodulation products into the same termination resistor. This occurs because many low-order reverse intermodulation products at individual amplifier output nodes have the same phase as the reflected reverse wave. Therefore, they also end up in the terminated port. This side effect can be used to improve the linearity of high-efficiency amplifiers including Doherty amplifiers (see, e.g., Pashaeifar, M. et al., (2021), “A Millimeter- Wave Mutual-Coupling-Resilient Double-Quadrature Transmitter for 5G Applications”, IEEE Journal of Solid-State Circuits, 56(12), 3784- 3798).
  • Doherty amplifiers see, e.g., Pashaeifar, M. et al., (2021), “A Millimeter- Wave Mutual-Coupling-Resilient Double-Quadrature Transmitter for 5G Applications”, IEEE Journal of Solid-State Circuits, 56(12), 3784- 3798).
  • one challenge is that efficient amplifiers like Doherty amplifiers are usually more nonlinear than non-efficient amplifiers. Therefore, replacing efficient amplifiers with more linear types of amplifiers (e.g., simple class AB amplifiers) can lead to lower efficiency.
  • a first aspect of the present disclosure provides an amplifier circuit comprising: a first input configured to receive a first signal; N first transmission lines, wherein JV > 3; JV second transmission lines; a signal combining node; N amplifiers; and a network of resistors comprising N third inputs.
  • Each amplifier comprises: a second input configured to receive the first signal with a respective first phase shift; and an output connected to a first end of a respective first transmission line.
  • a second end of the respective first transmission line is connected to a first end of a respective second transmission line.
  • the respective first transmission line is configured to introduce a respective second phase shift in a signal transmitted by the respective first transmission line.
  • the respective second transmission line is configured to introduce a third phase shift of substantially 90 degrees in a signal transmitted by the second transmission line.
  • Second ends of the second transmission lines are connected to the signal combining node.
  • the first transmission lines and the second transmission lines are configured such that signals output from each amplifier arrive at the combining node substantially in phase.
  • the network of resistors is a star or delta network of resistors.
  • Each of the third inputs are connected to the second end of a respective one of the first transmission lines.
  • a second aspect of the invention provides a communication device comprising an amplifier circuit according to the first aspect of the invention.
  • the amplifier circuit according to embodiments of the disclosure may have one or more of the following technical advantages. Reverse intermodulation distortion levels in signals output from the amplifier circuit may be reduced, which in turn enables the use of more efficient amplifiers and less over-dimensioned amplifiers in the amplifier circuit.
  • circulators/isolators can be omitted from the amplifier circuit, resulting in a cheaper and more efficient amplifier circuit which is capable of operating at higher frequencies.
  • amplifier circuits according to embodiments of the disclosure do not require hybrid couplers, and the numbers of amplifiers in the circuit do not need to be powers of two or have factors of two in them.
  • the amplifier circuits according to embodiments of the disclosure are more flexible than known solutions since any number of amplifiers (3 or more) can be used (e.g., the amplifier circuits may comprise 3, 5, 7, 9, etc., amplifiers). Numbers of amplifiers can be achieved which are not possible with known solutions. The reverse intermodulation characteristics of a specific type of amplifier in a specific setting can thus often be handled more efficiently.
  • the potential omittance of balanced amplifiers or hybrids from amplifier circuits according to embodiments of the disclosure can make hierarchical combiner arrangements unnecessary. This enables lower loss implementations of the amplifier circuits, as using a single layer combiner results in signals having a shorter path from the amplifiers to the output.
  • figure 1 is a circuit diagram of an amplifier circuit according to some embodiments
  • figure 2 is a collection of graphs illustrating the properties of waves in three amplifier systems
  • figure 3 is a circuit diagram of an amplifier circuit according to some embodiments
  • figure 4 is a circuit diagram of an amplifier circuit according to some embodiments
  • figure 5 is a graph illustrating the power of reverse waves relative to the power of carrier waves in a system with respect to wave frequency
  • figure 6 is a circuit diagram of an amplifier circuit according to some embodiments
  • figure 7 is a circuit diagram of an amplifier circuit according to some embodiments
  • figure 8 is a communication device according to embodiments of the disclosure.
  • Figure 1 is a circuit diagram of an amplifier circuit 100 according to some embodiments.
  • the amplifier circuit 100 of figure 1 can be used in some examples to amplify a first signal whilst reducing the effects caused by reverse waves existing in the circuit.
  • reverse waves in the circuit e.g., caused by reflections from impedance discontinuities in transmission lines and/or entering the circuit as external interference from other radio sources
  • This can generate reverse intermodulation distortion products (also referred to herein as simply “intermodulation products”) at the output of the amplifiers that may be of the following types:
  • the amplifier circuit of figure 1 may, in some examples, address issues caused by the first and third types of intermodulation products. That is, the amplifier circuit 100 causes reverse intermodulation distortion products to be steered into resistive terminations (a network of resistors) while combining all wanted output power at one output (a signal combining node).
  • the amplifier circuit 100 comprises a first input 102, N amplifiers 104 (where N is an integer and JV > 3), N first transmission lines 106 (which may also be referred to herein as “delay” or “phasing” lines), N second transmission lines 108 (which may also be referred to herein as 90 degree transmission lines), a signal combining node 110, and a network of resistors 1 12. At least one of the amplifiers 102 may be a PA.
  • the first transmission lines 106 and the second transmission lines 108 are configured to transmit signals output from the amplifiers 102 to the signal combining node 1 10.
  • the first input 102 is configured to receive a first signal that is to be amplified by the amplifiers 104. In some embodiments, the first signal may be received from a signal source that is not illustrated in figure 1 . In some embodiments, the phase of the first signal at the first input 102 is ⁇ p 0 .
  • the first input 102 is connected to each one of the amplifiers 104, and each amplifier comprises a respective second input 114.
  • the second inputs 114 are configured to each receive signals from the first input 102.
  • each amplifier receives, from the first input 102, the first signal with (i.e., modified to include) a respective first phase shift, 0 n (where 1 ⁇ n ⁇ N, and n is an amplifier index).
  • the respective first phase shift may include: any phase change caused by the first signal travelling from the first input 102 to the second input 114 of the respective amplifier; and/or any additional phase change to be introduced into the first signal.
  • the relative phase differences between first signals arriving at a pair of amplifiers 104 may:
  • the respective first phase shift may be any one or more of the following: different for each amplifier;
  • 0 t may be 0 degrees
  • 0 2 may be 60 degrees
  • 0 3 may be 120 degrees.
  • there may be:
  • phase shifts 0 n may be arbitrary, but nevertheless result in signals from the amplifiers 104 arriving at the signal combining node 110 substantially in phase (e.g., the relative phase shifts between each of the first signals are configured to achieve this purpose).
  • the first phase shift may be introduced to the first signals by third transmission lines (not illustrated) which connect the first input 102 to each of the second inputs 144.
  • the third phase shift may be introduced by configuring the (electrical) lengths of the third transmission lines.
  • the length of each third transmission line may be l 0 + l 0n , where l 0 is a constant value for each third transmission line and l Sn is a length introducing a phase shift in the first signal of 0 n (where 1 ⁇ n ⁇ W).
  • the third transmission lines may have electrical length that differ by 180/tV degrees between a first and second third transmission line, and between a second and third transmission line, and so on.
  • At least some of the third transmission lines may comprise respective phase shifters that introduce the respective first phase shift.
  • at least one of the respective phase shifters may comprise a modified Schiffman phase shifter, as discussed below in reference to figure 4.
  • each amplifier 104 further comprises an output 116 which is connected to a first end of a respective first transmission line 106, such that signals output from a respective amplifier 104 are input into the first end of a respective first transmission line 106. Signals that are input into a first end of a respective first transmission line 106 are output from a second end of the respective first transmission line 106. The second end of the respective first transmission line 106 is connected to a first end of a respective second transmission line 108, such that signals output from the second end of the respective first transmission line 106 are input into a first end of the respective second transmission line 108. Signals that are input into the first end of the respective second transmission line 108 are output from a second end of the respective second transmission line 108.
  • the second ends of respective second transmission lines 108 are connected to the signal combining node 110, and signals that are output from second ends of second transmission lines 108 are input into the signal combining node 110.
  • the first transmission lines 106 and the second transmission lines 108 are therefore configured such that signals output from the amplifiers 104 (at substantially the same time) arrive at the combining node 1 10 substantially in phase.
  • the first transmission lines 106 are configured such that signals input into the first ends of the first transmission lines 106 from the amplifiers 104 are output from the second ends of the first transmission lines 106 substantially in phase. This is achieved by configuring each first transmission line 106 such that it introduces a respective second phase shift, a n , (where 1 ⁇ n ⁇ N) in a signal transmitted by the respective first transmission line 106.
  • the introduction of the respective second phase shift in the signal transmitted by the respective first transmission line 106 may, for example, compensate for the effect of the respective first phase shifts applied to the first signal.
  • the respective second phase shift may include: any phase change caused by the first signal travelling from the output 1 16 of an amplifier 104 to the second end of the connected first transmission line 106; and/or any additional phase change to be introduced into the first signal.
  • the respective second phase shift a n applied to the first signal may be X - 0 n , where X is a constant (e.g. 180 degrees).
  • X is a constant (e.g. 180 degrees).
  • the introduction of the respective first phase shift results in the first signal being delayed by 0 n degrees
  • the introduction of the second phase shift a n may result in the first signal being delayed by a further 180 - 0 n degrees (or an equivalent phase shift, such as 540 - 0 n ).
  • the phase shifts a N may be arbitrary, but nevertheless result in signals from the amplifiers 104 arriving at the signal combining node 110 substantially in phase (e.g., the relative phase shifts between each of the first signals are configured to achieve this purpose).
  • the first signals output from the second ends of the first transmission lines 106 may have relative phase differences substantially equal to 0 degrees (or equivalent, such as integer multiples of 360 degrees).
  • the second phase shift is introduced by the first transmission lines 106 by configuring their (electrical) lengths.
  • the length of each first transmission line 106 may be l 0 + l an , where l 0 is a constant value for each first transmission line 106 and l an is a length introducing a phase shift in the first signal of a n (where 1 ⁇ n ⁇ JV).
  • the first transmission lines 106 may have electrical length that differ by 180/tV degrees between a first and second first transmission line 106, and between a second and third first transmission line 106, and so on.
  • the first transmission lines 106 may comprise respective phase shifters that introduce the respective second phase shift.
  • the respective phase shifters may comprise a modified Schiffman phase shifter, as discussed below in reference to figure 4.
  • each second transmission line 108 is configured to introduce a third phase shift, ⁇ >, of substantially 90 degrees in a signal transmitted by respective second transmission lines 108 (i.e., a substantially 90 degree phase shift is applied to each of the signals transmitted by the second transmission lines 108).
  • the third phase shift may be introduced by the second transmission lines 108 by configuring their (electrical) lengths.
  • the length of each second transmission line 108 may be Z ⁇ , where 1 is a length configured to introduce a phase shift in the first signal of 90 degrees.
  • At least some of the second transmission lines 108 may comprise respective phase shifters that introduce the respective third phase shift.
  • at least one of the respective phase shifters may comprise a modified Schiffman phase shifter, as discussed below in reference to figure 4.
  • the first signals output from the amplifiers 104 arrive at the signal combining node 1 10 substantially in phase.
  • the signal combining node 110 is configured to combine signals that are input substantially simultaneously into the signal combining node 1 10.
  • signals output from each amplifier 104 may arrive at the signal combining node 110 at substantially the same time and may be combined into a combined signal.
  • the first signals are substantially in phase when they arrive at the signal combining node 110, they may interfere constructively, providing a combined signal with a larger amplitude than each individual first signal arriving at the signal combining node 110.
  • the signal combining node 1 10 is configured to output the combined signal to one or more antennas and/or an antenna array included in the circuit of figure 100 (not illustrated).
  • the one or more antennas and/or the antenna array may be configured to transmit the combined signal to a wireless receiving device (e.g., a user equipment (UE), a radio access network (RAN) node, etc.).
  • a wireless receiving device e.g., a user equipment (UE), a radio access network (RAN) node, etc.
  • the network of resistors 112 comprises a plurality of resistors (e.g., a star or delta network of resistors).
  • the network of resistors 112 also comprises N third inputs 118. Each respective one of the third inputs 118 is connected to the second end of a respective one of the first transmission lines 106.
  • the network of resistors 112 may be configured such that at least one resistor is located between each pairing of second ends of the first transmission lines 106. For example, for each pairing, each end of the at least one resistor may be connected to one of the second ends of the pairing.
  • the network of resistors 112 is configured to dissipate signals arriving at the second ends of the first transmission lines 106 out of phase (in which case a potential difference exists across at some of the third inputs 118). The more out of phase the signals, the more the signal energy can be dissipated.
  • signals arriving at the second ends of the first transmission lines 106 in phase are not dissipated by the network of resistors 112 (as no potential difference exists across the third inputs 118).
  • the signals arriving in phase also arrive at the second ends of the first transmission lines 106 with substantially the same amplitude.
  • the amplifier circuit 100 deals with intermodulation products of the first type by being configured such that signals input into the amplifiers 104 (derived from a single source, e.g., first signals) arrive at the third inputs 118 with a relative phase difference of substantially 0 degrees (or equivalent). Since all amplified forward waves are in phase and of equal amplitude at the third inputs 118, they produce no voltage differentials over the network of resistors 1 12, resulting in no ohmic loss for the forward amplified first signals. Instead, the output power from the amplifiers 104 usefully combines in phase after the second transmission lines 108 at the signal combining node 110 (as the second transmission lines 108 do not change the relative phase differences between each of the first signals).
  • phase of any intermodulation products in the reverse wave reflected from the amplifier 104 are shifted by twice the respective second phase shift before the reflected reverse wave arrives at a third input 118 of the network of resistors 112.
  • each of the respective second phase shifts introduced by the first transmission lines 106 have magnitudes that are spread evenly over 180 degrees (or are equivalent thereto)
  • the phases of the intermodulation products in the reflected reverse wave at the third inputs 1 18 of the network of resistors 1 12 will be spread over a 360 degree range.
  • a large voltage differential may exist over the network of resistors 1 12, causing a large percentage of the reverse wave energy to be input into the network of resistors 112, causing it to be resistively terminated.
  • the intermodulation products can be greatly reduced in amplitude and/or cancelled out completely. This may also apply to phases that are not evenly spread over a 360 degree range (even if less of the reverse wave energy may be dissipated over the network of resistors 112 in some examples).
  • the network of resistors 112 can similarly deal with intermodulation products of the third typel if their phases vary randomly. That is, as the value of N increases, it becomes more likely that intermodulation products of the third type in the reflected reverse waves arrive at the second ends of the first transmission lines 106 out of phase, thus causing a similar power differential to that discussed above across the network of resistors 112.
  • the value of N may be selected such that the average phase difference between intermodulation products of the third type arriving at the third inputs 118 is non-zero.
  • the amplifier circuit 100 can reduce distortion and variations in power output from amplifiers 104 by selectively terminating reverse intermodulation products of the first and third type.
  • reverse intermodulation products that are dealt with by known balanced amplifiers are also reduced by the amplifier circuit 100.
  • the amplifier circuit 100 reduces intermodulation products more effectively. This is illustrated by the graphs in figure 2.
  • the three columns in figure 2 each correspond to a respective amplifier system.
  • the left-hand side column is for a system comprising a Doherty amplifier
  • the middle column is for a system comprising two Doherty amplifiers in a balanced configuration
  • the right-hand side column is for a three amplifier system according to embodiments of this disclosure (in this example an amplifier system comprising three Doherty amplifiers).
  • Each column contains, from top to bottom, an efficiency vs. output amplitude plot, an output phase vs. input amplitude plot, and an output amplitude vs. input amplitude plot.
  • First traces (black lines) 202 are for forward waves in the amplifier systems
  • second traces (light grey lines) 204 are for standing waves in the system.
  • the single Doherty amplifier suffers large variations in the output amplitude, phase and efficiency.
  • the balanced configuration corresponding to the middle column provides a reduction of these variations.
  • the amplifier circuit 100 (even when comprising just three amplifiers) improves upon these known solutions. This is especially noticeable when comparing the output phase vs. input amplitude plots of each column.
  • the first transmission lines 106 may have characteristic impedances of Z m (the impedance after output matching).
  • the network beyond the signal combining node 110 may have some other impedance if these lines are also used for impedance transformation.
  • the parallel combination of the second transmission lines 108 results in a load impedance of Z m .
  • the impedance at the network beyond the signal combining node 110 is the same as the impedances of the first transmission lines 106.
  • Wilkinson power combiner/divider The combination of a star or delta network of resistors 112 and impedancetransforming 90-degree transmission lines to a common point (i.e., a signal combining node 110) may in some examples be referred to as a Wilkinson power combiner/divider. Whilst embodiments of the present disclosure may use Wilkinson power combiner/dividers as examples, the skilled person would appreciate that embodiments of the disclosure are also suitable for use with any other in-phase power combiner which resistively terminates differential signals arriving at its inputs.
  • An input signal is provided via each third transmission line 302 to respective amplifiers 104.
  • the three third transmission lines 302 introduce the respective first phase shift to the inputs 114 of each of the amplifiers 104 such that the otherwise identical input signals arrive at the amplifiers 104 with a relative phase difference between them.
  • the three amplifiers 104 are connected at their output side, via three respective first transmission lines 106, to a Wilkinson combiner 304 (or an equivalent power combiner/divider).
  • a Wilkinson combiner 304 or an equivalent power combiner/divider.
  • the path of each signal travelling along a respective third transmission line 302 and respective first transmission line 106 have a substantially equal total delay or phase shift.
  • signals output from the amplifiers 104 should effectively be modified by the opposite delay or phase shift to the ones applied to the input signals before being amplified by the amplifiers 104.
  • the third transmission lines 302 may introduce delays on signals they carry of a common delay, pbase, minus the delays of the corresponding third transmission line 302 (e.g., pbase- pdell , pbase-pdel2, and pbase-pdel3, respectively). This is illustrated in figure 3.
  • the reverse wave parts may interact with the linear and nonlinear elements output from the amplifiers 104 and produce a number of different intermodulation products in addition to a linearly reflected signal.
  • the reflected signals and the intermodulation products travel back to the Wilkinson combiner 304 through the same first transmission lines 106, essentially doubling the phase shift applied to the reverse wave parts.
  • any phase introduced to signals due to a common line length of the first transmission lines 106 is inconsequential.
  • the output matching networks of the amplifiers 3104 can thus have any length/number of stages, if they are equal.
  • the third and first transmission lines 302, 106 are preferably arranged such that the double phase shifts applied to the reverse wave parts (e.g. 2*pdel1 , etc) are spread evenly over 360 degrees (or are equivalent thereto). As such, linearly reflected signals and some intermodulation products arrive at the Wilkinson combiner 304 with large (and potentially maximal) voltage differences over the network of resistors 112 forming part of the Wilkinson combiner 304.
  • the resistors of the network of resistors 112 substantially terminate the reverse wave parts and intermodulation products (i.e., resistively terminate).
  • the 90-degree transmission lines 108 also forming the Wilkinson combiner 304 transform the differential short circuit at the output side (i.e., at the combination point 110) to have very high impedance, meaning that the differential signals all go into the resistors.
  • the network of resistors 112 is a star network with a floating node.
  • a delta network of resistors comprising resistors 310 between each pair of nodes could instead be used.
  • the resistors 310 may have three times the resistance of the resistors in the star network, as suggested above for example.
  • An advantage of using a delta network is that it does not have a floating node.
  • the degree of cancellation of the reverse wave parts by the Wilkinson combiner 304 of figure 3 depends on the amplitude and phase balance between the paths from the amplifier outputs to the Wilkinson combiner 304 (e.g., how well matched/similar the phases and amplitudes of the first signals reaching the third inputs 118 are). Both the amplitude balance and phase balance depend on production and signal variations, but the phase has extra complications.
  • the lengths of the third and first transmission lines 302, 106 may be configured to introduce phase shifts to the signals they carry.
  • first and third transmission lines 302, 106 impart phase shifts on the signals they carry which vary linearly with the frequency of the signals, meaning the phase balance between the signals is sensitive to frequency variations in the signals carried by the third and first transmission lines 302, 106.
  • the effects of these phase variations may be reduced by using transmission lines 302, 106 with a smaller variation in phase differences over the whole frequency band of interest. Only the phase differences between signals carried by the first transmission lines matter 106 with regards to the Wilkinson combiner 304, so any common phase component can, in principle, have arbitrary frequency dependence.
  • a modified Schiffman phase shifter may be formed in at least one of the first or third transmission lines 302, 106. This may provide an arrangement that gives an approximation to constant phase differences between the signals carried by the third and/or first transmission lines 302, 106, regardless of any frequency variations they may experience.
  • a Schiffman phase shifter comprises two branches.
  • a first branch comprises a pair of 90-degree coupled transmission lines bridged at their far ends.
  • a second branch comprises a 270-degree transmission line (i.e., a transmission line which introduces a phase delay/shift of 270 degrees in a signal it carries) which is uncoupled from the pair of 90-degree transmission lines.
  • the Schiffman phase shifter has close to 90 degrees of phase difference between signals transmitted by each branch over a relatively wide band.
  • Figure 4 illustrates the amplifier circuit 100 of figure 14, but comprising modified Schiffman phase shifters 402.
  • modified Schiffman phase shifters 402 have been adapted so that they may be used: in the amplifier circuit 100 with comprises three branches; and to result in phase differences other than 90 degrees (e.g., 60 degrees or 120 degrees).
  • Schiffman phase shifters may also be adapted for use in amplifier circuits with more than three branches.
  • An advantage of the amplifier circuit 100 of figure 4 is that its bandwidth is increased. That is, as the effects of phase variations on the signals carried by the third and first transmission lines 302, 4106 is reduced, they have more constant phase differences over a wider frequency band.
  • figure 5 illustrates the performance of transmission lines for a relative bandwidth of 10%.
  • the upper trace 502 illustrates transmission lines which introduce phase shifts on signals they carry by configuring their electric lengths.
  • the lower trace 504 illustrates transmission lines which introduce phase shifts on signals they carry by using modified Schiffman phase shifters.
  • the total reverse intermodulation distortion power relative to the carrier power for the lower trace 504 is lower (by several decibels) for wider carrier/reverse wave frequency separations than the upper trace 502. Therefore, amplifier circuits according to embodiments of the disclosure can be further improved for use with wider range frequency separations/to have larger bandwidth by stacking several coupled line sections with different impedances and coupling factors.
  • the Wilkinson combiner 304 has the advantage that it can easily be adapted for use with more amplifier branches (i.e., when N > 3). This can result in incrementally lower levels of reverse intermodulation distortion.
  • a multi-input Wilkinson combiner or equivalent may be used with a set of first transmission lines 106 (e.g., delay lines or transmission lines comprising phase shifters) that introduce respective second phase shifts into the signals they carry, wherein the respective second phase shifts average out the phase of the reflected waves and intermodulation products over 360 degrees.
  • first transmission lines 106 e.g., delay lines or transmission lines comprising phase shifters
  • the higher order arrangements have more freedom than the lower ones in that they can be grouped into individually balanced subgroups.
  • first transmission lines 106 which introduce a 90 degree phase difference shift on the signals they carry, and a triplet of first transmission lines 106 which introduce a 60 degrees phase difference on the signals they carry, and a freely chosen difference relative phase difference between signals transmitted by these groups. This degree of freedom could be useful in some applications.
  • a star resistor arrangement may have the same number of resistors as the number of amplifier branches, whilst a delta resistor arrangement may have increasingly more (since it has resistors between all pairs of nodes). Therefore, star resistor arrangements may be favorable for higher-order implementations of the amplifier circuits according to embodiments of the disclosure, that is, with an increasing number of amplifiers in the amplifier circuit.
  • Wilkinson combiners with more than two inputs can be problematic in that it is sometimes challenging to get all input nodes of the Wilkinson combiner into proximity with the resistors they are connected to.
  • the length of transmission line used for the connection can change the electrical behavior of the Wilkinson combiner, input nodes that are not in proximity with the resistors they are connected to can cause the Wilkinson combiner to have a reduced terminating effect.
  • this challenge can be overcome by having series capacitors in these lines, so that they become effectively zero-length (or equivalently zero-phase) over a certain bandwidth.
  • a Gysel power divider/combiner is used in place of a Wilkinson combiner.
  • Gysel power divider/combiners use an additional transmission line network to enable all resistors to have one end coupled to a ground instead of a port node. This can reduce the effect of parasitic capacitance from the termination resistors on the network.
  • the combiner employed may be made from a number of Wilkinson combiners arranged hierarchically, wherein the number of third inputs for each combiner is less than the number of inputs of a single Wilkinson combiner, if the single Wilkinson combiner were to be used in the amplifier circuit.
  • Figure 7 illustrates a composite amplifier circuit 700.
  • the asymmetrical 2-input Wilkinson combiner 708 is similar to the previously discussed Wilkinson combiners (e.g., they comprise second transmission lines 108 and a network of resistors 112) but is different in that resistors of their network of resistors 112 may differ in resistance and the second transmission lines 108 have different characteristic impedances.
  • one resistor of the asymmetrical 2-input Wilkinson combiner 708 is Z m /3 ohms, whilst the other is m l2 ohms.
  • the characteristic impedances for the second transmission lines 108 connecting the asymmetrical 2-input Wilkinson combiner 708 to the first amplifier circuit 702 is m l
  • the characteristic impedances for the second transmission lines 108 connecting the asymmetrical 2-input Wilkinson combiner 708 to the second amplifier circuit 702 is m/2.
  • the asymmetrical 2-input Wilkinson combiner 708 can in some examples include a resistance between each branch of the asymmetrical 2-input Wilkinson combiner 708. This is shown in figure 7 as two series resistors 112, with resistances Z m /3 and Z m /2 respectively. In other examples, however, a single resistor or any suitable combination of resistors may be used, serial combinations of resistors that provide the required resistance can be used in each branch.
  • fourth transmission lines 706 may connect the outputs of the first and second amplifier circuits 702, 704 (e.g., the signal combining nodes 110) to inputs of the asymmetrical 2-input Wilkinson combiner 708.
  • the fourth transmission lines 706 may be configured to introduce respective fourth phase shifts on signals carried by the fourth transmission lines 706, such that signals output from the first and second amplifier circuits 702, 704 arrive at a signal combining node 710 substantially in phase (wherein the signal combining node 710 of figure 7 performs the same function as the signal combining node of figure 1 ).
  • the respective fourth phase shifts introduced to signals carried by the fourth transmission lines 706 may be configured such that they account for the respective second phase shifts introduced to signals carried by the first transmission lines 106 of each of the first and second amplifier circuits 702, 704.
  • a composite circuit according to embodiments of the disclosure may comprise any number of amplifier circuits, wherein at least one or more of the amplifier circuits has a respective number N of amplifiers, where JV > 3.
  • pairs of amplifier circuits may be combined with any asymmetrical 2- input Wilkinson combiner 708 (or any equivalent asymmetrical combiner).
  • an equivalent combiner may comprise a network of resistors comprising fourth inputs, wherein the fourth inputs are respectively connected to the signal combining nodes of the first amplifier circuit and the second amplifier circuit via the fourth transmission lines 706, and may also comprise resistors with different resistances.
  • each “amplifier” in an amplifier circuit according to this disclosure could be an amplifier circuit according to this disclosure, e.g. an amplifier circuit where JV > 3 could itself feed a Wilkinson combiner of two or more inputs.
  • the Wilkinson combiner may be symmetrical instead of asymmetrical.
  • the amplifiers in an amplifier circuit are of substantially the same size or output substantially the same power.
  • Amplifier circuits according to embodiments of the disclosure are more flexible than known solutions as there is no requirement for balanced amplifiers or hybrids.
  • This enables systems with odd numbers of amplifiers and makes hierarchical combiner arrangements with several layers unnecessary.
  • the former is beneficial for avoiding unnecessarily large systems (e.g., a three-amplifier system can be used rather than a four-amplifier system, or a five-amplifier system can be used rather than an eightamplifier system).
  • the latter allows for lower loss in the circuit, since fewer layers means that the signals have a shorter path from the amplifiers to the output.
  • FIG 8 is a schematic diagram of a communication device 802.
  • the communication device 802 may be a wireless communication device, such as a UE or a Radio Access Network (RAN) node.
  • the communication device 802 comprises an amplifier circuit according to the embodiments discussed in relation to figures 1 -7.
  • the communication device 802 may be configured to communicate with another wireless receiving device.
  • RAN Radio Access Network

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Amplifiers (AREA)

Abstract

La présente invention concerne un circuit amplificateur qui comprend : une première entrée conçue pour recevoir un premier signal ; N premières lignes de transmission, avec N > 3 ; N deuxièmes lignes de transmission ; un noeud de combinaison de signaux ; N amplificateurs ; et un réseau de résistances comprenant N troisièmes entrées. Chaque amplificateur comprend : une deuxième entrée conçue pour recevoir le premier signal avec un premier décalage de phase respectif ; et une sortie connectée à une première extrémité d'une première ligne de transmission respective. Une deuxième extrémité de la première ligne de transmission respective est connectée à une première extrémité d'une deuxième ligne de transmission respective. La première ligne de transmission respective est conçue pour introduire un deuxième décalage de phase respectif dans un signal transmis par la première ligne de transmission respective et la deuxième ligne de transmission respective étant conçue pour introduire un troisième décalage de phase de sensiblement 90 degrés dans un signal transmis par la deuxième ligne de transmission. Des deuxièmes extrémités des deuxièmes lignes de transmission sont connectées au noeud de combinaison de signaux. Les premières lignes de transmission et les deuxièmes lignes de transmission sont conçues de telle sorte que des signaux émis par chaque amplificateur arrivent au niveau du noeud de combinaison sensiblement en phase. Le réseau est un réseau en étoile ou en triangle de résistances. Chacune des troisièmes entrées est connectée à la deuxième extrémité d'une ligne respective parmi les premières lignes de transmission.
PCT/IB2023/059808 2023-09-29 2023-09-29 Circuit amplificateur Pending WO2025068755A1 (fr)

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Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4547745A (en) * 1983-02-28 1985-10-15 Westinghouse Electric Corp. Composite amplifier with divider/combiner
US20120258677A1 (en) * 2008-05-28 2012-10-11 Hollinworth Fund, L.L.C. Power amplifier architectures
US8952752B1 (en) * 2012-12-12 2015-02-10 Nuvotronics, Llc Smart power combiner
KR102546533B1 (ko) * 2021-05-31 2023-06-23 한밭대학교 산학협력단 쉬프만 위상 천이기를 이용한 다중 대역 도허티 증폭기

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4547745A (en) * 1983-02-28 1985-10-15 Westinghouse Electric Corp. Composite amplifier with divider/combiner
US20120258677A1 (en) * 2008-05-28 2012-10-11 Hollinworth Fund, L.L.C. Power amplifier architectures
US8952752B1 (en) * 2012-12-12 2015-02-10 Nuvotronics, Llc Smart power combiner
KR102546533B1 (ko) * 2021-05-31 2023-06-23 한밭대학교 산학협력단 쉬프만 위상 천이기를 이용한 다중 대역 도허티 증폭기

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
PASHAEIFAR, M. ET AL.: "A Millimeter-Wave Mutual-Coupling-Resilient Double-Quadrature Transmitter for 5G Applications", IEEE JOURNAL OF SOLID-STATE CIRCUITS, vol. 56, no. 12, 2021, pages 3784 - 3798, XP011888963, DOI: 10.1109/JSSC.2021.3111126

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