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WO2023247947A1 - Dispositif de commande amélioré - Google Patents

Dispositif de commande amélioré Download PDF

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Publication number
WO2023247947A1
WO2023247947A1 PCT/GB2023/051616 GB2023051616W WO2023247947A1 WO 2023247947 A1 WO2023247947 A1 WO 2023247947A1 GB 2023051616 W GB2023051616 W GB 2023051616W WO 2023247947 A1 WO2023247947 A1 WO 2023247947A1
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WO
WIPO (PCT)
Prior art keywords
phase
current
rotating machine
electric rotating
commutation
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
PCT/GB2023/051616
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English (en)
Inventor
Martin Palmer
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Aeristech Control Technologies Ltd
Original Assignee
Aeristech Control Technologies Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Aeristech Control Technologies Ltd filed Critical Aeristech Control Technologies Ltd
Publication of WO2023247947A1 publication Critical patent/WO2023247947A1/fr
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
    • H02P6/182—Circuit arrangements for detecting position without separate position detecting elements using back-emf in windings
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/15—Controlling commutation time
    • H02P6/157—Controlling commutation time wherein the commutation is function of electro-magnetic force [EMF]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
    • H02P2207/055—Surface mounted magnet motors

Definitions

  • This invention relates to improved controllers for electric machines, in particular to improved controllers for controlling a motor.
  • a permanent magnet motor comprises a permanent magnet rotor and a stator with windings.
  • the permanent magnet rotor includes a number of magnetic pole pairs. Electrical energy is converted into mechanical energy due to magnetic forces between the permanent magnets and the magnetic field induced due to the current in the stator windings.
  • Each commutation event corresponds to the period for which current flows to the same windings. For example, for a 4-pole 3-phase motor, a single commutation event lasts through 30 degrees of shaft rotation. At the end of the commutation event, the permanent magnets are adjacent to different windings and switching of the current is required.
  • controllers regulate the current in the motor windings. Pulse width modulation is a well-known method for regulation of current in motor windings.
  • the present invention provides improvements to the robustness of sensorless commutation of BLDC motor controllers.
  • FIG. 1a is a waveform diagram of a conventional motor
  • FIG. 1 b represents the structure of the conventional motor of FIG. 1a;
  • FIG. 1c is a waveform diagram for an alternative conventional motor
  • FIG. 2a is a waveform diagram of a motor that can be utilised in embodiments of the present invention.
  • FIG. 2b represents the structure of the motor of FIG. 2a
  • FIG. 3 is a functional block circuit diagram of a control circuit embodying the present invention.
  • FIG. 4 is a block diagram showing a detail of the circuit of FIG. 3;
  • FIG. 5a is a waveform diagram showing an ideal back EMF in a three phased motor (named a, b, and c) utilized in an embodiment of the present invention
  • FIG. 5b is a phase to phase back EMF derived from measuring total back EMF across two phases (a and b, b and c, a and c);
  • FIG. 5c is a filtered waveform diagram of the phase to phase waveform of FIG. 5b;
  • FIG. 6 is a circuit diagram of a low-pass filter that may be used in the control circuit embodying the present invention;
  • FIG. 7b is a wave form diagram of the commutation of the current to the individual phases that may be derived by the control circuit embodying the present invention.
  • FIG. 8a is a wave form diagram of phase currents generated when the motor of FIG. 2b is utilised as a generator;
  • FIG. 8b is a waveform diagram of the rectified phase currents shown in FIG. 8a;
  • FIG. 8c is a waveform diagram of the rectified phase current from a conventional generator
  • FIG 9 is a signal path illustration of the phase voltages and comparator signals
  • FIG 10 is a prior art method of measuring phase voltage differences
  • FIG 11a and 11 b are graphs of the integrated signals of Figure 9 (FIG 11a) and the prior art method (FIG 10) respectively;
  • FIG 12 is an output of measured phase-phase voltage showing the simulated output when the motor is being driven vs the output while the motor is not being driven;
  • FIG 13 shows a calibration curve for selecting a desired delay time for a delay timer
  • FIG 14a and 14b show implementation of the delay from the delay timer on the comparator signals in relation to the current supplied to each inverter gate signal, with multiple phases/timers shown in Figure 14b;
  • FIGS 15a and 15b show circuit used in the present invention.
  • FIG. 1a shows the ideal current 10 that must be supplied to each phase winding in a prior art synchronous AC motor, as shown in FIG. 1 b, (or conversely the current generated by a prior art synchronous AC generator).
  • a sinusoidal current 10 (with commutation frequency 16) is optimal for this type of prior art motor and care and effort is taken to provide as close a representation 12 of a sinusoidal wave pattern as possible when driving such a prior art motor.
  • PWM Pulse Width Modulation
  • the average pulse width and the timing of the pulse By varying the average pulse width and the timing of the pulse (switching frequency 14) an overall current approximating a sinusoidal wave can be generated. Amplitude is varied by controlling the average pulse width and commutation is controlled by changing the timing of the pulses. Generally, the current pulses 12 are applied with multiple phases, most preferentially three different phases differing by 120 degrees.
  • FIG. 1 b shows a prior art brushless AC motor 20 with a four pole permanent magnet rotor 22 which is mounted on a shaft 24.
  • the motor has four magnetic poles spaced around its circumference.
  • the magnetic poles are provided by four permanent magnets 26, 28, 30, 32 spread around 360 degrees; however each magnet spans only 60 degrees, separated from its neighbours by 30 degrees dead space.
  • the magnets 26, 28, 30, 32 naturally generate a ‘blocky’ north- south-north-south magnetic field around the motor.
  • FIG. 1b An example of how the windings 34 of the 3-phase voltages are distributed around the magnets as shown in FIG. 1b. Only one loop of the winding 34 has been shown for clarity. It may be seen that the winding 34 emerges from the slot 36 adjacent to magnet 28, before passing through the slot 36 adjacent to the edge of magnet 30. This winding pattern creates differing magnetic fields within the windings dependent upon the relative position and direction of the winding relative to the magnets of the motor. It may, of course be appreciated that by varying the winding pattern the properties of the motor may be tailored.
  • this machine is a generator, the potential created in each coil (all of which are connected in series for one phase) changes more-or-less independently from one another as the rotor moves, and the distribution of the coils is chosen such that the total potential rises and falls in a nearly-sinusoidal pattern
  • IGBTs In conventional brushless DC control, six IGBTs (A+, A-, B+, B-, C+, C-) such as illustrated in FIG. 4 are utilized to control both the commutation (timing) and voltage regulation (quantity). Voltage regulation is implemented by Pulse Width Modulation (PWM) as shown in FIG. 1c. Pulses 12a of constant voltage amplitude are supplied to the motor and form a square-wave voltage 10a. In this case the amplitude of the voltage 10a is determined by the number of pulses and their duration or width (the duty cycle).
  • PWM Pulse Width Modulation
  • Pulses 12a of constant voltage amplitude are supplied to the motor and form a square-wave voltage 10a. In this case the amplitude of the voltage 10a is determined by the number of pulses and their duration or width (the duty cycle).
  • the inductance and resistance of the motor provides an inherent regulation of current. However, the inductance and resistance of the motor reduce efficiency and make the motor unsuitable for
  • the IGBT switching frequency 14 would need to be at least 100* higher than the commutation frequency 16. With the high operating speeds desired of embodiments of the present invention, this control approach becomes impractical.
  • the motor 40 employed by the current invention uses a 12 slot design.
  • a representation of this motor is shown in FIG. 2b.
  • the four magnets 41-44 span the full 360 degrees of the rotor 46 without any dead space, creating a continuous permanent magnet shell, so the motor is (in general) 50% more powerful for a given size compared to the 15-slot motor.
  • 12 slots 48 and three phases 50 allow 4 coils or slots per phase, which corresponds to the 4 magnetic poles on the rotor. Therefore each coil 50 can always be fully excited by the magnets 41-44.
  • the coils 50 in any one phase are wound clockwise-anticlockwise-clockwise- anticlockwise, so the north-south-north-south magnetic field reinforces and drives the maximum current through the stator (in the case of a generator) or creates maximum torque from a given current (in the case of a motor).
  • this machine would supply a square-wave output that is difficult to deal with.
  • smooth rotation of the shaft 47 requires a square-wave current input 60 which is difficult to supply.
  • a motor with the features of this 12- slot machine described here is not a popular choice in most prior-art applications. If this motor were chosen for its compactness and efficiency and then driven using a prior-art PWM controller, the result would be unsmooth (varying with time) motor output and additional electrical losses, negating some of motor's inherent benefits.
  • FIG. 2a shows the current 60 that must ideally be imparted upon each phase winding 50 for a motor 40 (FIG. 2b) designed for square-wave input.
  • gaps 66 are necessary as a function of stator geometry and to prevent non-ideal excitation of the rotor when the magnetic poles of the rotor are not aligned with the permanent magnet energised by the coil. During these gaps 66, current is imparted by a different phase.
  • the relevant switching points 68 between the application or removal of current is the commutation timing and ideally occurs when the pole of the rotor passes into or out of the influence of the magnet energised by the windings.
  • the resistance and inductance of the windings 50 are much smaller than in a typical DC brushless motor, such as the prior art motor of FIG. 1 b.
  • one phase 50 is connected to the positive (current travelling in), one phase is connected to the negative (current travelling out), and one phase is floating (no current).
  • the current should be injected to each phase 50 when that phase exhibits the maximum back EMF compared to the other phases and should return from each phase when that phase exhibits the minimum back EMF.
  • the commutation timing 68 must be accurately controlled. If the back EMF is ideal, the commutation timing 68 can be obtained by comparing the three phase voltage (e.g., when a phase exhibits the greatest back EMF, then that phase current is switched ‘on’).
  • FIG. 3 shows a main embodiment of the present invention and details the controller 80 used.
  • a principle feature of this controller 80 is that it addresses power separately from commutation. This control approach is achieved by a logical separation between the control of aggregate current i182 flowing to the motor 84 and the commutation of that current iu, iv, iw 86a-c on the phase connectors of the motor 84.
  • the aggregate current 82 has two proportional-integral (PI) feedback control loops 88, 90 that regulate aggregate current 82.
  • the inner loop 88 controls the current amplitude directly and the outer loop 90 adjusts the current in response to the torque requirement (speed/target speed mis-match) of the motor 84.
  • the inner loop 88 comprises a duty cycle 92 that provides the amplitude of the aggregate current 82 and a (amplitude) regulator 94 that compares the present aggregate current 82 to the current requested by the outer loop 90. If the aggregate current 82 requested by the outer loop 90 is greater than the currently supplied aggregate current then the current is adjusted to match the desired current by the duty cycle 92. It can be appreciated that the inner loop 88 can be considered to be a regulating feedback loop for regulating the current amplitude.
  • the outer loop 90 also comprises a (speed) regulator 94 that compares a speed target 96 with the current speed of the motor 98 and determines the aggregate current 82 required to accelerate to the speed target 96.
  • a saturation check 100 is provided to ensure that the current requirements are within the capability of the controller 80 and the motor 84.
  • the speed of the motor is provided by a FA/ converter 102 that analyses back EMF signals Vw, Vv, Vu 104 obtained from the motor and converts them to determine the motor speed 98 and the angular position of the motor (and the magnets).
  • the components used to regulate the aggregate current 82 (the inner and outer feedback control loops 88, 90) may be considered as a current supply feedback loop for providing a current amplitude to the motor 84 windings.
  • the use of the back-EMF signal generated by the strong permanent magnets moving past the windings in the motor is advantageous because the back-EMF manifests itself as an oscillating variation in the apparent electrical resistance across each phase connection of the motor's stator winding. This gives an indication of the instantaneous position of the rotor relative to the stator and thus the appropriate timing for the electrical excitation of the stator.
  • the motor's phase connections carry the output of the motor controller (oscillating current to excite the motor's stator windings) as well as one of its inputs (back-EMF to determine the commutation pattern).
  • the present invention utilises back-EMF signals to determine motor speed and position
  • alternative ways of monitoring the motor and producing reference signals may be utilised.
  • alternatives include: the use of an external rotor position sensor, most likely an optical type or electromagnetic interference (Hall effect) sensor type responding to markings or shapes (e.g., compressor blades) on the motor shaft; the use of a timekeeping device internal to the controller, which is regularly calibrated or reset (e.g., once per motor shaft rotation) by a coarse sensor; a measurement of the commutation current, or a signal indicative thereof, relating to the current induced in the motor windings (not the total current going to the motor); and the use of purely internal logic and timekeeping which makes assumptions about the position of the rotor and the required commutation without expecting or without caring that this may fall out of synchronicity with the true, optimal commutation timing (e.g., the rotor may ‘slip’ relative to the electrical excitation).
  • an external rotor position sensor most likely an
  • This two-tier approach is implemented in order to prevent an over-current condition, because the motor 84 is optimally designed for very low internal inductance and is therefore highly sensitive to damage unless current 82 is tightly controlled on a short timescale.
  • the control system 80 measures the frequency of the motor back EMF 104 to get the motor speed 98.
  • the control system can control the torque. If the motor 86 needs to accelerate, the controller 90 will increase the current command to increase the torque.
  • the commutation of the aggregate current 82 is implemented separately and is shown to the right of the motor 86.
  • the commutation pattern 110 responds passively to the motor position as measured by tracking the back-EMF 104 displayed on the phase connectors.
  • the preferred embodiment uses the phase-to-phase voltage to measure back-EMF. This would normally lead in phase by 90 degrees relative to the optimal current commutation timing, based on the typical properties of motors (see below).
  • the preferred embodiment therefore implements a low-pass filter 112 which produces a 90 degree phase shift in the measured phase-to-phase voltages. This low-pass filter 112 additionally removes errors from the back- EMFsignal 104 and simultaneously adjusts the phase angle so that the timing is appropriate for use as a current commutation control signal.
  • the commutation pattern 110 is determined, it is provided to the IGBT module 114.
  • the aggregate current 84 can then be regulated by the IGBT module 114 in the required commutation pattern 110 to deliver the required current iu, iv, iw 86a-c to the motor 84.
  • This combination of components 110, 112 and 114 act as a commutation feedback loop for controlling the timing and duration of excitation current supplied to the motor windings.
  • FIG. 4 highlights the duty cycle 92 and the IGBT module 114.
  • the duty cycle 92 acts as a “DC/DC current source” part and creates a nearly continuous current of controlled aggregate amperage 82.
  • the duty cycle has two IGBTs 120, 122 and by switching on and off the IGBTs, the aggregate current 82 can be regulated.
  • the duty cycle 92 is connected to the IGBT module 114, which acts for a three phase signal as a six-leg inverter. Because of the high fundamental frequency of the motor, this IGBT module 114 only controls the commutation, and need never interrupt the aggregate flow of current to control power (as it would have to do in a more conventional control layout).
  • the “inverter” part takes as input a commutation signal from a digital controller (not shown) and the aggregate current 82 produced by the duty cycle 92.
  • the IGBT module 114 produces square wave current signals to drive the PM motor.
  • the function of the IGBT module 114 is to deliver whatever aggregate current 82 is available from the duty cycle 92 directly to the motor 84 using the simple switching pattern shown in FIG. 2a.
  • IGBT's 116a, 116b that switch on and off the aggregate current 82 supply to the required commutation pattern 110.
  • Similar IGBT's 118a, 118b, 120a, 120b perform the same function for each additional phase of current iv 86b, iw 86c. Therefore the current supplied by each phase can be either positive, negative or zero.
  • the back-EMF signals 104 generated by the motor 86 are shown if FIG. 5a.
  • the three back-EMF signals 104a, 104b and 104c correspond to the three phases of the input currents 86a, 86b, 86c.
  • the back-EMF shown in FIG. 5a is idealised. In reality the back-EMF signal 104 is often choppy and distorted, making determining the angular position of the motor and therefore determination of the commutation timing difficult. Furthermore, in practical motor control, the commutation itself disturbs the back-EMF 104 because of the rapid phase current changes. This disturbance can deform the shapes of the back-EMF wave forms, which makes the comparison between them no longer reliable. Additionally, due to the practicalities of wiring the controller, measuring a single back-EMF 104a is difficult.
  • the reliability of the back-EMF signal is further improved by measuring the phase-to-phase voltages 130 of the back EMF 104 (allowing the controller to monitor the same wires that are used to impart current to the motor) as shown in FIG. 5b.
  • the phase-to-phase voltages 130 (back EMF's) are not aligned with the phase voltages (back EMF's) 104.
  • the Phase A and Phase B crossing point (marked as Point 1 ) in FIG. 5a would be the Phase A-B zero crossing point (marked as Point 2) in FIG. 5b.
  • the phase to phase crossing point determines the optimal position for switching the current supplied to the motor to the next corresponding phase, determination of this position is critical to ensure efficient use of the motor.
  • the phase-to-phase voltage is the difference between two phase voltages, and so the difference in phase between these two signals can be calculated as follows:
  • Phase B voltage sin(x-pi/3)(120 degree phase offset in 3-phase motor)
  • phase to phase crossing point (point 1 in FIG. 5a) is no longer within an easily determined position (point 2 in FIG. 5b).
  • the three phase back EMF is therefore filtered before the comparisons are carried out.
  • a low-pass filter may be used.
  • the ideal switching timing is obtained by considering the crossing points between phase voltage signals.
  • the controller uses filtered phase-to-phase voltage signals, which are in total 60 degrees behind the phase voltages (30 degrees-90 degrees). Because a commutation event occurs every 60 degrees (see FIG. 5c), these filtered phase-to-phase voltages can be used, although the mapping of which crossover points are associated with which phase current signals is different to the mapping that would apply if the phase voltages were used.
  • FIG. 5c shows the filtered phase to phase signals with 90 degree phase shift considered. Aligned with Point 1 and 2 in FIGS. 5a and 5b, the corresponding point in FIG. 5c is marked as Point 3 which is the crossing point between Phase B-C and Phase C-A.
  • FIG. 5c shows that, despite the constant phase shift, the commutation timing 110 for current switching can still be determined by comparing the magnitudes of the filtered, phase-to-phase voltages.
  • the three voltage signals produced by the phase-to-phase voltage filter can then be compared using well-known electronic components. The results of the comparisons can be decoded to generate the commutation output, as can be seen in FIG. 7a.
  • C1152 is the comparison results between the filtered Va-b and Vb-c.
  • C2154 is for Vb-c and Vc-a.
  • C3156 is for Vc-a and Va-b.
  • the six IGBTs 116a, 116b, 118a, 118b, 120a, 120b which control the commutation pattern of current to the motor can be controlled completely and optimally by the signals C1 , C2, C3, as shown in the bottom graph of FIG. 7b.
  • the controller 80 imparts upon the motor's phase connections 86a-c oscillating current signals whose waveforms are shaped and spread (phase offset) in such a way that the sum of their absolute values is always equal to the constant (aggregate current 82) signal from which they were constituted.
  • This multiphase commutation pattern 110 is the controller's output and is sent to the phase connection points available in the motor's stator winding.
  • This controller 80, including the commutation part, is electronic, rather than mechanical, which improves efficiency and reliability compared to rubbing or sliding mechanical switches embedded in the motor and potentially moving at high speed.
  • the controller 80 may be used to run the motor 84 as a generator.
  • movement of the rotor of the generator relative to the stator causes a current to flow within windings of the stator.
  • the commutation circuit in such embodiments pulls the current off the windings, creating (in the above example) a three phase power signal.
  • the controller 80 operates the motor 84 as a generator, the controller 80 continues to operate in the same manner as described above, independently of the current source.
  • the motor 84 is run as a generator, the current source is essentially reversed and so the direction of the current flow is also reversed causing current to flow out of the commutation circuit and motor 84. Due to the arrangement of the motor or generator, such output is a DC signal or current. By varying this DC current between positive and negative setpoints, the controller allows rapid and seamless transition from motor to generator to motor.
  • the commutation pattern produced by the generator embodiment of the present invention is a series of square waveforms as shown in FIG. 8a and similar to those shown in FIG. 7a.
  • the generator (of the form of the motor 40) produces three phase current signals 180 (comparison between A and B of FIG. 7b), 182 (comparison between B and C of FIG. 7b) and 184 (comparison between C and A of FIG. 7b).
  • Each of the three phase signals generated 180, 182, 184 produce a signal that varies from a positive phase current 180a to a negative phase current 180e via zero net phase current plateaus 108c, 180g and stepped functions 180b, 180d, 180f, 180h between the positive, negative and net zero plateaus. It can be appreciated that the exact form of the waveforms is likely to vary from this idealised normal representation.
  • the type of generator described here produces square wave output 180, 182, 184 from the individual generator phases, which, when rectified to DC, is smooth 190 (except for harmonics 192).
  • a representation of the rectified DC current is shown in FIG. 8b.
  • Harmonics 192 may occur during the transition from one square-wave phase output 180 to the next 182 (at each step function interval).
  • the overall signal 190 is smoother than a rectified 3- phase sinusoidal signal 194 as shown in FIG. 8c.
  • the signal 190 varies by approximately 0%, rather than the 10% typical of rectified 3-phase sinusoidal signals 194.
  • phase to phase terminal voltages use a differential amplifier with attenuation as a first stage in the circuit.
  • the signal path uses phase voltages II, V and W. These are combined as shown in Figure 9. They are combined via differential amplifiers and integrators to give II’, V’ and W’ signals. These are then combined using comparators to provide comparator signals Cap1 , Cap2 and Cap3.
  • the crossing points of the integrated signals II’, V’, W’ can be provided by feeding each pair (U’,V’; V’,W’; W.IJ’) into a comparator as in the schematic on Figure 9.
  • the present invention does not neglect the diode conduction (“freewheeling”).
  • the artefacts it introduces to the phase-phase voltages have an advancing effect on the commutation timing.
  • the graph shown in Figure 12 shows measured phasephase voltages just before and just after stopping driving the motor.
  • the time-bases have been shifted by a multiple of the electrical period so that the curves can be overlaid.
  • the measured phase-phase voltage is the pure back-EMF.
  • the blue curve is the simulated output of the integrator circuit with the motor being driven.
  • the grey curve is the output while the motor is not being driven. It can be seen that the integrator largely filters out the spikes due to freewheeling I diode currents, but it results in a slight shift to the left. The commutation timing is therefore advanced.
  • a small advance in timing is beneficial since it can remove any phase lag in current waveform compared to the back-EMF.
  • This phase lag is a natural consequence of the inductance of the motor which causes the current to rise and fall more slowly after voltage is applied or removed.
  • the length of the freewheeling periods depends on the relative magnitudes of motor current, motor inductance and back-EMF voltage. In some early designs, where the freewheeling periods were long, the phase advance was too much, and the timing needed to be corrected.
  • the decoding of zero crossing signals into inverter gate signals by a small extra microcontroller may be implemented.
  • Integrated timers capable of being triggered directly by the change of state of an input GPIO pin allows the timing to be delayed.
  • the delay duration is calculated from a lookup table using the measured speed as a parameter.
  • the software is calibrated by seeing how the motor current changes as the delay was varied and selecting the value where the current was a minimum or at least was at the midpoint of the “bathtub” curve: shown in Figure 13.
  • FIG. 14a shows how the CAP signal received from the zero crossing signal is delayed by a timer.
  • a change of state event in any of the signals causes a timer to begin a count. Once a programmed count value is reached, the timer triggers an interrupt which changes the inverter gate signals.
  • Figure 14b shows three timers are used - one for each CAP signal. This may reduce the complexity of the microcontroller used such that separate timers are used, rather than triggering a single timer from three different signals. Additionally, multiple timers may be run simultaneously.
  • a change of state of in a CAP signal causes the inverter signal to trigger a predetermined length of time thereafter.
  • the integrator circuit as shown in Figure 6 has a resistor R16 to slowly discharge the capacitor C2. This prevents any small DC offset in the input from charging C2, causing the op-amp output to saturate close to +15V or -15V.
  • the combination of R16 and R23 still gives a gain of 50 to any DC offsets.
  • the DC offsets can be caused by nonidealities of the op-amps used, tolerances of components or asymmetry of motor phase-to-phase voltages. Any small offsets would be amplified 50 times. This can cause asymmetry of the output signals (Cap1 , Cap2, Cap3) and so incorrect timing of commutation switching.
  • a standard positive feedback was added to the comparator to give some hysteresis (R604 in the schematic of Figure 15b).
  • C602 a capacitor was added (C602) which causes a larger hysteresis, decaying over a time of around 50 ps. This helps to prevent any electrical noise due to the inverter switching (and elsewhere) from causing false transitions of the commutation circuit outputs while two of the signals II’, V’, W are necessarily close in voltage since they are in the process of crossing each other. It also helps to limit the maximum switching frequency of the inverter in the case of oscillation and prevent damage to components. It will be related to the natural frequency of the RC network created by R605, R606 and C602 ( ⁇ 20 kHz in this case).

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

L'invention concerne une machine tournante électrique à aimant permanent et un système de commande. Une pluralité d'enroulements de phase peuvent être excités par un courant d'excitation unique provenant d'une alimentation électrique. Un circuit de commutation, fonctionnellement indépendant de l'alimentation électrique, est opérationnel pour commander une temporisation et une durée d'alimentation du courant d'excitation transmis à ou tiré individuellement des enroulements de phase. Le circuit de commutation comprend en outre un retardateur, ledit retardateur tenant compte du retard de phase dans l'impulsion de courant par rapport à la force contre-électromotrice provenant de la machine tournante électrique à aimant permanent.
PCT/GB2023/051616 2022-06-20 2023-06-20 Dispositif de commande amélioré Ceased WO2023247947A1 (fr)

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GBGB2209066.6A GB202209066D0 (en) 2022-06-20 2022-06-20 Improved controller
GB2209066.6 2022-06-20

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Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2011161408A2 (fr) 2010-06-22 2011-12-29 Aeristech Control Technologies Limited Dispositif de commande
WO2017029502A1 (fr) 2015-08-17 2017-02-23 Aeristech Control Technologies Limited Commande de moteurs et générateurs à aimant permanent

Patent Citations (4)

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