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WO2020092849A1 - Système et procédé de télémétrie ultra-haute résolution à l'aide d'une identification par radiofréquence - Google Patents

Système et procédé de télémétrie ultra-haute résolution à l'aide d'une identification par radiofréquence Download PDF

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Publication number
WO2020092849A1
WO2020092849A1 PCT/US2019/059274 US2019059274W WO2020092849A1 WO 2020092849 A1 WO2020092849 A1 WO 2020092849A1 US 2019059274 W US2019059274 W US 2019059274W WO 2020092849 A1 WO2020092849 A1 WO 2020092849A1
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WIPO (PCT)
Prior art keywords
signal
tag
frequency
transceiver
additional
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
PCT/US2019/059274
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English (en)
Inventor
Xiaonan HUI
Edwin C. Kan
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Cornell University
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Cornell University
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Publication date
Application filed by Cornell University filed Critical Cornell University
Priority to US17/289,702 priority Critical patent/US11519996B2/en
Priority to CA3117204A priority patent/CA3117204A1/fr
Priority to EP19878965.3A priority patent/EP3874298A4/fr
Publication of WO2020092849A1 publication Critical patent/WO2020092849A1/fr
Anticipated expiration legal-status Critical
Priority to US17/900,480 priority patent/US11914018B2/en
Ceased legal-status Critical Current

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Classifications

    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S1/00—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith
    • G01S1/02—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith using radio waves
    • G01S1/08—Systems for determining direction or position line
    • G01S1/20—Systems for determining direction or position line using a comparison of transit time of synchronised signals transmitted from non-directional antennas or antenna systems spaced apart, i.e. path-difference systems
    • G01S1/30—Systems for determining direction or position line using a comparison of transit time of synchronised signals transmitted from non-directional antennas or antenna systems spaced apart, i.e. path-difference systems the synchronised signals being continuous waves or intermittent trains of continuous waves, the intermittency not being for the purpose of determining direction or position line and the transit times being compared by measuring the phase difference
    • G01S1/306—Analogous systems in which frequency-related signals (harmonics) are compared in phase, e.g. DECCA systems
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S1/00—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith
    • G01S1/02—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith using radio waves
    • G01S1/022—Means for monitoring or calibrating
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S1/00—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith
    • G01S1/02—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith using radio waves
    • G01S1/04—Details
    • G01S1/042—Transmitters
    • G01S1/0428—Signal details
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G01S13/32—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated
    • G01S13/36—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated with phase comparison between the received signal and the contemporaneously transmitted signal
    • G01S13/38—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated with phase comparison between the received signal and the contemporaneously transmitted signal wherein more than one modulation frequency is used
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/74—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems
    • G01S13/82—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems wherein continuous-type signals are transmitted
    • G01S13/84—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems wherein continuous-type signals are transmitted for distance determination by phase measurement
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/87—Combinations of radar systems, e.g. primary radar and secondary radar
    • G01S13/878—Combination of several spaced transmitters or receivers of known location for determining the position of a transponder or a reflector
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/74—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems
    • G01S13/75—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems using transponders powered from received waves, e.g. using passive transponders, or using passive reflectors
    • G01S13/751—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems using transponders powered from received waves, e.g. using passive transponders, or using passive reflectors wherein the responder or reflector radiates a coded signal

Definitions

  • the present disclosure relates to range finding, and more particularly, to radio frequency-based range finding.
  • CMOS complementary metal-oxide-semiconductor
  • ToF time-of-flight
  • LIDAR structured- light range scanning and light detection and ranging
  • ETltrasound ranging and imaging is an alternative approach, but has issues in terms of impedance matching when going through different layers of materials, especially for air gaps.
  • Another approach is to use location sensors based on microelectromechanical systems (MEMS), such as the servo motor and encoder.
  • MEMS microelectromechanical systems
  • MEMS microelectromechanical systems
  • the size and mechanical structure of such systems cannot effectively fit onto the human body, delicate robotic structures, or soft materials.
  • MEMS accelerometers and gyroscopes also suffer from slow drift and their power requirement limits deployment options.
  • RFID RF identification
  • RSSI received signal strength indication
  • SAR synthetic aperture radar
  • iSAR inverse SAR
  • the landmark-tag method uses known coordinates and RSSI of the reference tags to retrieve the position of the unknown target tag, but suffers from insufficient range sensitivity and ambiguity from multipath interference.
  • the SAR and iSAR methods require relative motion between the tags and reader, which is not feasible for many scenarios.
  • the phased-array radar forms narrow beams to isolate the coverage areas, but the scanning time and spatial resolution of the beam will constrain the system localization capability.
  • FMCW frequency modulation continuous wave
  • FFT fast Fourier transform
  • phase information of the RF signal is more sensitive to the distance between the transmitter (Tx) and receiver (Rx) than RSSI, and can give higher precision if the wavelength ambiguity can be resolved.
  • Rx can use the demodulated phase of the received signal to retrieve the Tx-to-
  • the present disclosure provides devices and methods able to achieve a ranging resolution smaller than, for example, 50 micrometers using a harmonic ultrahigh frequency (UHF; for example, 300 MHz-3 GHz) RF transponder system with a sampling rate of 1 kHz or higher.
  • UHF harmonic ultrahigh frequency
  • high spatial resolution can be achieved using extremely high frequency, such as in the collision avoidance radar system of 79 GHz with 4 GHz bandwidth.
  • lack of dielectric penetration is a severe limitation for extremely high frequency ranging in many applications.
  • the presently-disclosed UHF system can potentially achieve the maximum distance of conventional RFID systems, for example, around 15 meters in free space with a Tx power which may be below 30 dBm, and can see through dielectrics such as water and common building materials.
  • a radio-frequency method for range finding includes: modulating a reference signal having an intermediate frequency, f IF , to a downlink signal having a carrier frequency, f c , using a clock signal; transmitting the downlink signal to a tag using a transmitter, the tag being located at a distance from the transceiver; receiving an uplink signal backscattered from the tag, the uplink signal having a frequency that is a harmonic of the carrier frequency; demodulating the uplink signal using the clock signal; and calculating a distance between the tag and the transceiver based on a phase of the demodulated uplink signal.
  • the distance is calculated by comparing the coherent reference signal to the demodulated uplink signal.
  • the uplink signal may be at a second harmonic of the carrier frequency, for example, to lower phase noise interference from leakage of the downlink signal.
  • the step of calculating the distance may be repeated at a sampling rate to update the distance.
  • the intermediate frequency, f IF may be greater than a frequency where Flicker noise power density is equal to the thermal noise density.
  • the method may include calculating a moving average comprising a predetermined number of most recent calculated distances.
  • the uplink signal may have a unique digital identification code to provide isolation from ambient noise.
  • the unique digital identification code may be encoded using a code-division multiple access (CDMA) protocol to provide isolation from other tags.
  • CDMA code-division multiple access
  • the method may include modulating the reference signal to one or more additional downlink signals each having an additional carrier frequency and each of the additional carrier frequencies generated using a corresponding clock signal, and wherein each of the additional carrier frequencies is not equal to / c ; transmitting, using a corresponding one or more additional transceivers, the one or more additional downlink signals to the tag; receiving one or more additional uplink signals backscattered from the tag, each of the one or more additional uplink signals being at a second harmonic of a corresponding one of the one or more additional carrier frequencies; demodulating each of the one or more additional uplink signals using the clock signal of the corresponding one or more carrier frequencies; calculating a distance between the tag and each additional transceiver based on a difference between a phase of the reference signal and a phase of a corresponding one of the one or more demodulated uplink signals.
  • Obtaining the distance may further include dividing a result by the square root of a relative permittivity of a medium between the transceiver and the tag, wherein the relative permittivity of the medium is known and relative to a vacuum.
  • the method may include determining a relative permittivity of a medium in which the tag disposed, using the obtained distance and a known range of the tag, wherein the relative permittivity of the medium is relative to a vacuum.
  • a radio-frequency system for range finding includes a transceiver and a processor.
  • the transceiver is configured to modulate a reference signal having an intermediate frequency, f IF , to a downlink signal having a carrier frequency, f c , using a clock signal; transmit the downlink signal; receive a backscattered uplink signal from a tag, wherein the uplink signal is at a harmonic frequency of the carrier frequency; and demodulate the uplink signal using the clock signal such that transceiver is a coherent transceiver.
  • the processor is configured to receive the demodulated uplink signal and calculate a distance between the tag and the transceiver using a phase of the demodulated uplink signal.
  • the processor may be further configured to repeatedly calculate the distance at a sampling rate.
  • the processor may be further configured to calculate a moving average comprising a predetermined number of most recent calculated distances.
  • the system may include a tag configured to receive the downlink signal at f c and to backscatter the uplink signal at the harmonic frequency of the carrier frequency.
  • the harmonic frequency of the carrier frequency may be the second harmonic.
  • the tag may be configured to encode a digital identification code onto the uplink signal.
  • the tag may be configured to encode the uplink signal using a code-division multiple access (CDMA) protocol.
  • CDMA code-division multiple access
  • the system may include one or more additional tags, each configured with a unique digital identification or CDMA code.
  • the tag may be configured to transform the downlink signal to the uplink signal without offsetting a phase of the downlink signal.
  • the tag may include a non-linear transmission line.
  • the system may include an analog-to-digital converter to convert the
  • the system may include a clock for generating the clock signal, wherein the clock is in communication with the transceiver.
  • the system may further one or more additional transceivers, each configured to modulate and transmit the reference signal at a corresponding one or more additional carrier frequencies, and to receive and demodulate corresponding uplink signals, each uplink signal at a harmonic of a corresponding one of the one or more additional carrier frequencies.
  • the processor may be further configured to determine a relative permittivity of a medium along a path between the tag and each transceiver of the one or more additional transceivers, using an obtained distance from the corresponding transceiver and a known range of the tag from the
  • Figure 1 is a diagram of a system for range finding an object according to an embodiment of the present disclosure.
  • Figure 2 is a chart of a method according to another embodiment of the present disclosure.
  • FIG 3 is a diagram of an experimental embodiment of a harmonic RFID ranging system according to the present disclosure.
  • the system comprises a harmonic reader and tag, and the system is shown with a micrometer platform.
  • the reader is connected to a dual -band antenna, which transmits an interrogating signal and then receives a backscattered signal from the harmonic tag.
  • a demodulated and digitalized signal is then sent to a processor to calculate ranging information.
  • FPGA field-programmable gate array
  • MCU micro control unit
  • LPF low-pass filter
  • HPF high-pass filter
  • FIG. 5 A photograph of a prototype passive harmonic tag.
  • ANT 1 Antenna 1
  • Ant 2 Antenna 2
  • NLTL Non-linear transmission line
  • Figure 7 Experimental results of quasi-static ranging (a) Time-domain measurements in air and water with steps of 50 pm. (b) Probability density curves when the tag was in air, sand, and water with steps of 50 pm. The solid and dashed lines correspond to data collection within 2 s and 0.1 s, respectively (c) Resolution analysis when the tag was in water.
  • Figure 8 Experimental results of tag movement and permittivity based on a two- tag structure (a) The long-travel round-trip ranging accuracy with the DFCW method in water.
  • the forward trip is in blue (line with square markers) and the backward trip in pink (line with circular markers).
  • the average errors for the forward and backward trips are shown as dashed lines
  • the curves Cl to C7 correspond to a tag speed of 65.60 mm s -1 for Cl and the successive half scaling for C2 to C7.
  • the bottom and top edges of each box are the 75 th and 25 th percentiles of the permittivity ratio, and the middle line inside each box indicates the median.
  • the bottom and top edges of the whiskers are the minimum and maximum values of the analyzed data.
  • Figure 9 A backscattering EIHF RFID system
  • (a) A block diagram of a conventional RFID system, where phase-based ranging would suffer from the high noise skirt of the Tx signal through direct leakage and antenna reflection to Rx.
  • (b) A block diagram of a harmonic system with high Tx and Rx isolation according to an embodiment of the present disclosure
  • (c) A diagram of an Rx signal chain of a harmonic receiver according to an embodiment of the present disclosure.
  • Figure 10 A photograph of an experimental embodiment of a harmonic RFID ranging system and test platform.
  • Figure 11 Simulation results of RMS ranging variations under different local oscillators (LOs).
  • LOs local oscillators
  • Figure 12 Simulation results of reducing the ranging variation by different moving average window sizes (a) Ranging variation with windows Wl to W4 at 1, 5, 20, and 100, respectively (b) The corresponding frequency response under each window size.
  • Figure 13 Simulation results of the ranging error caused by the frequency inaccuracy and IQ imbalance (a) Ranging error caused by the frequency inaccuracy where the error also accumulates with the distance (b) Ranging error caused by the IQ imbalance at different frequencies.
  • FIG. 14 Root-mean-square (RMS) ranging variation caused by analog-to- digital (ADC) RMS aperture jitter.
  • Signal -to-noise ratio, SNR ADC is set as 73 dB.
  • f c 1.5 GHz with the same f IF condition.
  • Figure 15 (a) A photograph of a tag antenna in water in an experimental embodiment (b) A photograph taken during the experiment using the embodiment of
  • Figure 15(a) wherein the main photograph shows the part of the antenna connector shown in the red boxed area of Figure 15(a).
  • Figure 16 Experimental ranging results of 50-pm stepping (a), (d), and (g) are transient signals when the tag was in the air, sand, and water, respectively (b), (e), and (h) are the probability densities of the ranging data in (a), (d), and (g), respectively.
  • the solid curves are the density collected in 2-s duration, while the dash curves are in 0.1 -s duration (c), (f), and (i) show the analyses of ranging resolutions in air, sand and water, respectively.
  • the present disclosure may be embodied as a radio-frequency method 100 for range finding— e.g., finding the distance to an object.
  • the method 100 includes modulating 106 a reference signal having an intermediate frequency, f IF (an“intermediate frequency signal”).
  • the reference signal may be generated 103 from a digital signal, for example, using an digital-to-analog converter.
  • the reference signal is modulated 106 to a downlink signal having a carrier frequency, f c , using a clock signal.
  • the downlink signal is transmitted 109 using a transceiver.
  • the downlink signal may be transmitted 109 to a tag located at a distance from the transceiver.
  • the downlink signal is not necessarily directed only to the tag, but may be, for example, wirelessly broadcast so as to be received at the tag.
  • the transceiver may be an RFID reader system or a part of such a system.
  • the method 100 includes receiving 112 an uplink signal that is backscattered from the tag, the uplink signal being at a harmonic of the carrier frequency.
  • the uplink signal may be at a second harmonic, 2 f c of the carrier frequency.
  • the harmonic RFID system makes use of harmonic backscattering to isolate the downlink (reader to tag) and uplink (tag to reader), which results in a much lower noise floor to achieve accurate ranging. Because of the backscattering scheme, the tag and reader carrier synchronization problem is also readily avoided. A detailed comparison of conventional and harmonic RFID systems is discussed below, which includes analyses of the operational range and link budget.
  • the uplink signal is demodulated 115 using the clock signal.
  • the modulating and demodulating steps use the same clock signal and are coherent.
  • the clock signal need not be at the carrier frequency (or a harmonic of the carrier frequency).
  • a mixer in the transmitter may use a local oscillator (at the carrier frequency) which is derived from the clock signal.
  • a mixer in the receiver may use a local oscillator (at a harmonic of the carrier frequency) which is derived from the clock signal.
  • a distance between the tag and the transceiver is calculated 118 based on a phase of the demodulated uplink signal.
  • the distance may be calculated 118 based on a difference between a phase of the reference signal and a phase of the demodulated uplink signal.
  • the step of calculating 118 the distance is repeated at a sampling rate.
  • the steps of transmitting the downlink signal, receiving the uplink signal, demodulating the uplink signal, and calculating the distance may each be repeated at a sampling rate.
  • the reference signal may be continuously modulated and transmitted (for some period of time), and the backscattered uplink signal may be continuously received and demodulated.
  • the demodulated uplink signal be used to repeatedly calculate the distance between the tag and the transceiver. In this way, a moving object can be tracked over its movement.
  • the sampling rate may be 10 Hz - 10 kHz, inclusive, including every integer Hz value therebetween ( e.g ., 20 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 1.5 kHz, 5 kHz, 8 kHz, 9 kHz).
  • the sampling rate may be higher or lower.
  • the demodulated uplink signal may be sampled 121 at a sampling rate so as to digitize the demodulated uplink signal.
  • the resulting samples may be averaged 124 over a moving average window.
  • the method 100 may further comprise modulating 150 the reference signal to one or more additional carrier frequencies, f c2 ... / cm.
  • Each of the one or more additional carrier frequencies may be generated using a corresponding clock signal.
  • the additional carrier frequencies may be coherent (using a same clock signal) or incoherent (using separate clock signals).
  • Each of the additional carrier frequencies is not equal to f c . In this way, multiple frequencies may be used to resolve wavelength ambiguity (providing range finding over a broader range).
  • One or more additional transceivers may be used to transmit 153 the one or more additional downlink signals to the tag, and a corresponding number of additional backscattered uplink signals are received 156 by the transceiver.
  • Each of the received 156 additional uplink signals is modulated at a harmonic of its corresponding one of the one or more additional carrier frequencies.
  • Each of the received 156 additional uplink signals is
  • a distance between the tag and each additional transceiver is calculated 162 based on a difference between a phase of the reference signal and a phase of a corresponding one of the one or more demodulated uplink signals.
  • the intermediate frequency, f IF , and/or the carrier frequency, f c may be selected to minimize Flicker noise and sampling jitter.
  • f IF may be chosen to be at a frequency greater than a frequency where the Flicker noise power density is equal to the thermal noise density.
  • sampling jitter f IF may be selected to be at a frequency below that at which performance is degraded by sampling jitter. Further examples are provided below.
  • the present disclosure may be embodied as a radio-frequency system 10 for range finding (see, e.g., Figure 1).
  • the system 10 includes a transceiver 20.
  • the transceiver 20 is configured to modulate a reference signal having an intermediate frequency f IF to a carrier frequency, f c .
  • the transceiver 20 may have a mixer to modulate the signal using a clock signal.
  • the system may include a clock to provide the clock signal.
  • the clock may be a part of the transceiver or separate from the transceiver.
  • the clock signal may be provided by a clock that is not a part of the system.
  • the transceiver 20 is configured to transmit the downlink signal.
  • the transceiver may have an antenna or be connected to an antenna.
  • the transceiver 20 is further configured to receive a backscattered uplink signal modulated at a harmonic of the carrier frequency.
  • the uplink signal may be backscattered by a tag, such as, for example, a tag 50 attached to the object 90 of interest (an object for ranging finding).
  • the uplink signal may be modulated at the second harmonic, 2 f c , of the carrier frequency.
  • the transceiver 20 is configured to demodulate the uplink signal using the clock signal.
  • the transceiver may use a frequency doubler to demodulate an uplink signal modulated at the second harmonic of the carrier frequency.
  • the transceiver may have a transmitter and a receiver, both operating (modulating and demodulating) using a same clock signal. In this way, the transmitter and the receiver are coherent.
  • the transmitter and receiver of the transceiver may be separate— e.g., without sharing a housing or circuitry. In other embodiments, the transmitter and receiver of the transceiver may in a common housing and/or share common circuitry.
  • the system 10 includes a signal processor 30.
  • the signal processor 30 is in communication with the transceiver 20.
  • the signal processor 30 is configured to receive the demodulated uplink signal from the transceiver 20, and to calculate a distance between the tag and the transceiver using a phase of the demodulated uplink signal.
  • the signal processor 30 may be a field-programmable gate array (FPGA).
  • the processor may include one or more modules and/or components.
  • the processor may include one or more hardware-based modules/components (e.g., an FPGA, a digital signal processor (DSP), an application specific integrated circuit, a general purpose processor, etc.), one or more software-based modules (e.g, a module of computer code stored in a memory and/or in a database), or a combination of hardware- and software-based modules.
  • hardware-based modules/components e.g., an FPGA, a digital signal processor (DSP), an application specific integrated circuit, a general purpose processor, etc.
  • software-based modules e.g, a module of computer code stored in a memory and/or in a database
  • a combination of hardware- and software-based modules e.g., a module of computer code stored in a memory and/or in a database
  • the tag 50 may form a component of the system 10.
  • a tag may be a separate component from the system in some embodiments.
  • the tag (harmonic tag) 50 is configured to receive the downlink signal from the transceiver 20 and to backscatter the uplink signal at a harmonic frequency of the carrier frequency.
  • the tag may be configured to backscatter an uplink signal at the second harmonic, 2 f c , of the carrier frequency, f c .
  • the tag may be a passive harmonic tag configured with a non-linear transmission line (NLTL) configured to backscatter the downlink signal at a harmonic frequency.
  • the tag may be configured to encode a digital identification onto the uplink signal.
  • the digital identification may be an identification code unique to the tag.
  • the digital identification may be encoded using a code- division multiple access protocol. In this way, the system may be able to distinguish between multiple tags which may be present.
  • the system may be configured for range finding of multiple objects using multiple tags.
  • the system 10 comprises one or more additional transceivers 25.
  • Each additional transceiver 25 is configured to modulate and transmit the reference signal at a corresponding one or more additional carrier frequencies, and to receive and demodulate corresponding uplink signals.
  • each uplink signal is at a harmonic of a corresponding one of the one or more additional carrier frequencies.
  • An exemplary system 60 (shown in Figure 3) was built to benchmark the performance of the presently-disclosed harmonic RFID ranging scheme.
  • the exemplary system included a harmonic reader 62 and tag 64.
  • a micrometer platform 66 was also provided to accurately move the tag.
  • a photograph of the experimental setup is shown in Figure 10.
  • a software defined radio (SDR, Ettus X310, UBX-160) was used as the harmonic reader, and the external clock 72 was derived from a rubidium frequency standard (FE-5650A) to provide a stable frequency reference.
  • a field-programmable gate array (FPGA) 74 was used to feed the digital signal to a digital-to-analog converter (DAC) to generate the intermediate-frequency (IF) signal, which was mixed with the downlink RF frequency (carrier frequency) of f c .
  • the Tx signal after a power amplifier and a low-pass filter (LPF) 78 was fed to a splitter 80, which was used as part of a broadband duplexer.
  • the downlink signal (the blue arrow) was received by a harmonic tag 64, which was mounted on a carriage block 92 to provide linear motion with micrometer-level accuracy and resolution through a worm shaft.
  • the passive harmonic tag received the downlink signal and harvested the energy to power up.
  • the tag modulated the backscattered signal with the code-division multiple access (CDMA) protocol together with the unique tag identification, so the system was able to distinguish each tag with simultaneous multiple access.
  • a nonlinear transmission line (NLTL) was designed on the tag to convert the backscattered signal to the second harmonic, which goes back to a reader antenna 75 and then a high-pass filter (HPF) 82 through the splitter 80.
  • the Rx signal was amplified by a low-noise amplifier (LNA) and down converted by a local oscillator (LO) at 2/ to the intermediate frequency, which was sampled by an analog-to-digital converter (ADC).
  • LNA low-noise amplifier
  • LO local oscillator
  • ADC analog-to-digital converter
  • the digitized intermediate frequency was processed by the FPGA and transmitted to a host computer. Because the Tx and Rx chains shared the same clock reference (indicated by the green arrows in
  • the harmonic reader was configured as a coherent transceiver.
  • FIG. 4 A schematic of the passive harmonic transponder is shown in Figure 4(a), and the transponder is pictured in Figure 5.
  • Antenna 1 (Ant. 1) received the downlink RF signal at f c .
  • the transponder was configured to harvest the RF signal using a charge pump to power up the tag receiver and the digital logic unit.
  • Antenna 2 (Ant. 2) also received the downlink signal and fed it to the non-linear transmission line (NLTL) to generate the 2 nd harmonic backscattering signal at 2 f c .
  • the other end of the NLTL was designed as an open circuit (OC), so the signal would be reflected and converted to the harmonic signal again, which increased the conversion efficiency.
  • the RF switch before the NLTL modulated the uplink signal by on-off keying (OOK), providing the uplink baseband information.
  • OOK on-off keying
  • FIG 10 is a photograph of the setup.
  • a field-programmable gate array (FPGA) feeds a digital signal to a digital-to-analog converter (DAC) to generate an intermediate frequency (IF) signal, which is mixed with the downlink frequency at f c .
  • the Tx signal after a power amplifier and LPF is fed to a splitter, which is utilized as part of the broadband duplexer.
  • the harmonic tag is mounted on the linear-module platform to provide motion with micrometer accuracy.
  • the backscattering signal (orange dashed arrow) from a transponder of the tag is received at the reader antenna, and goes through a two-way splitter and a high-pass filter (HPF) to an Rx chain of the SDR.
  • HPF high-pass filter
  • the Rx signal is amplified by a LNA and down-converted by the LO at 2 f c to f IF , which is then sampled by the ADC.
  • the digitized Rx signal (at f IF ) is processed by the FPGA and transmitted to the host computer.
  • the same digital clock signal (green dashed line) in Figure 4(b) is fed into two frequency synthesizers to make the SDR as the 2 nd harmonic coherent transceiver.
  • RF frontend methods can provide the harmonic coherent transceiver as well, such as using the nonlinearity of the power amplifier to obtain the LO signal at 2 f c and halving the Rx LO frequency by the divided-by-two module to serve as Tx LO.
  • the method of using the same frequency reference here presents reasonable performance and high reconfigurability.
  • phase noise can be a fundamental limit for the ranging system
  • a technique to achieve high resolution is to employ an adequate intermediate frequency f IF to avoid the low- frequency Flicker noise.
  • the ranging variation is related to the resolution, one of the most efficient ways to counter random noise is to apply the moving average.
  • the ranging variation with different window sizes of 1, 10, 100, lk, and lOk are shown in Figure 6(c), with the
  • the ADC sampling rate can be correspondingly tuned down to reduce power consumption, if such a feature is desirable for the reader transceiver.
  • a moving average is not an essential procedure in embodiments of the present scheme if, for example, the signal-to-noise ratio (SNR) of the backscattered LoS signal is sufficiently high under the low-noise system. Therefore, the trade-off between the frequency response caused by the window size and the ranging resolution is not limited by the same uncertainty principle as in the Fourier-based methods. A detailed discussion of the ranging variation related to the system configuration is provided below.
  • the initial tag position was calibrated as 0 to cancel the constant system phase offset, and then advance the carriage block to 50 pm and 100 pm.
  • the recorded time-domain signals at the three positions of 0, 50, and 100 pm are shown in Figure 7(a) in blue, red, and green, respectively.
  • the wavelengths at the uplink frequency of 2 GHz are 15 cm and 1.69 cm in air and water, respectively.
  • the downlink signal was set as 1 GHz.
  • the equivalent sampling rate was 1 kSps (kilo-samples per second), and a lk moving-average window was applied.
  • Figure 7(b) shows the ranging probability density when the tag was in air (low peaks), sand (middle peaks), and water (high peaks) at 0 (blue), 50 (red), and 100 pm (green).
  • the solid curves correspond to 2-s data collection and the dashed curves to O.l-s data collection, where hardly any difference can be observed.
  • the ranging probability density in water (blue solid curve) is examined in Figure 7(c) in more detail to calculate the resolution.
  • the full-width at half-maximum (FWHM) is at 5.9 pm. If the shape of the distribution is considered as the average of the rise and fall distances, the 10-90% probability then gives 4.7 pm for both rise and fall sides.
  • the FWHM resolutions in air and sand are 39.1 pm and 17.8 pm, respectively.
  • the carrier frequency can also affect the resolution as discussed above (regarding Figure 7(a)). Downlink signals of 0.5, 1, and 2 GHz were provided, and the tag was stepped by 50 pm in air in Figure 7(d). It can be seen that the higher frequency will make the ranging resolution slightly higher, but the compromise includes the shorter wavelength ambiguity and the larger attenuation at the same ranging distance.
  • the wavelength integer from cyclic ambiguity needs to be resolved to extend the maximum operation range for the phase-based methods.
  • the dual -frequency continuous- wave (DFCW) method was used to demonstrate implementation of the range extension.
  • Other techniques can be used.
  • sophisticated multi-frequency methods can provide more robust estimation with fewer constraints on the maximum range.
  • the sensing uplink signal of the experimental embodiment was around the 2 GHz band, in air the single-frequency method can cover a distance of about 15 cm, but only 1.69 cm in water.
  • the computer-control step motor drove the tag carriage forward for 5 cm in water, and backwards to the 0 point, for a travel of about three wavelengths.
  • the travel distance monitored by the motor rotation angle was chosen as the ground truth to benchmark the ranging accuracy of the experimental embodiment, as shown in Figure 8(a).
  • the square marks (forward) and circular marks (backward) denote the ranging error at every millimeter. From Figure 8(a), it can been see that DFCW is effective in providing accurate ranging with travel distances over several wavelengths in water. Besides, because two frequencies were applied, after resolving the wavelength integer, the final ranging result was obtained by averaging over measurements from the two frequencies, which further reduced the random noise. If multiple incoherent frequencies are used, more improvement can be expected.
  • Figure 8(b) shows the cumulative distribution function (CDF) of the ranging accuracy from the 5-cm-travel experiment.
  • the benchmark instances are extracted at every 1 mm of the ground-truth measurement.
  • the blue and pink curves are the CDF for forward and backward tag motion, respectively.
  • the yellow curve is the overall CDF.
  • the ground truth may be polluted by the mechanical accuracy and structural vibration during tag movement. It can be seen that the backward curve is worse than the forward one, which may be caused by the lost motion clearance of the linear module in the backward travel.
  • Figure 8(a) where the forward average is very close to 0 and the backward average has a positive bias, as shown in the dashed lines.
  • the tag antenna was about 0.4 m away from the reader Rx antenna. In view of the shorter wavelength and larger attenuation in water, it is estimated that the same SNR can equivalently operate in air at 3.5 m from the reader Rx antenna.
  • phase errors and uncertainties caused by multipath interference play an important part in ranging accuracy and resolution.
  • a worst-case multi-path signal at an orthogonal phase to the LoS path signal with 55 dB lower magnitude can already pose a phase error of 0.1°, which is at the phase noise tolerance limit.
  • the constant part of the phase offset can be reduced by the calibration step in a reasonably controlled indoor environment, which did not greatly contribute to the ranging errors in the experiments shown in this work.
  • the use of a high-directivity reader antenna also helped reduce the multi-path effect by providing low antenna gain for undesirable directions (though such an antenna is not required).
  • the ranging system may need to be adapted with broader bandwidth antennas with high directivity and/or a stable phase center, and/or more sophisticated algorithms, possibly with a compromised ranging accuracy and resolution.
  • some other hardware aspects may need to be considered as well in the system setup. For example, large signal-power dynamic range due to large coverage of the operation distance may need to be adaptively compensated by improved tag and reader designs to reduce the variations in harmonic conversion by the nonlinear element.
  • the harmonic reader was implemented using an SDR (Ettus X310 and UBX 160 MHz RF daughter boards).
  • the SDR was controlled by a computer with Lab VIEW and was connected to the computer via PCIe cable to provide a broad data bandwidth.
  • the sampling rates of the DAC and ADC are both configured at 66.7 MSps (mega-samples per second).
  • the LO of Rx is set as twice that of the LO of the Tx, so the SDR is configured as a coherent harmonic transceiver.
  • the external clock is provided by a rubidium frequency standard (FE-5650A, frequency stability: ⁇ 10-11, phase noise: -100 dBc at 10 Hz, -125 dBc at 100 Hz, and -145 dBc at 1 kHz), giving the 10 MHz sinusoidal wave reference.
  • FE-5650A frequency stability: ⁇ 10-11
  • phase noise -100 dBc at 10 Hz, -125 dBc at 100 Hz, and -145 dBc at 1 kHz
  • the square wave can provide a more stable clock reference, so the system performance can be further improved by using a better clock source.
  • the harmonic tag PCB (printed circuit board) prototype is based on the open-source WISP (wireless identification sensing platform).
  • NLTL is implanted on the tag to generate the second-harmonic signal, which is a ladder structure of inductors and varactors.
  • NLTL provides high harmonic conversion efficiency over broad bandwidth even when the received signal is weak.
  • Other frequency doublers can also be applied with passive or active tag designs. In the small-signal regime with the tag impinging power less than 0 dBm, the dependence of the backscatter phase shift on the power level is negligible.
  • the tag is mounted on a wooden slab, which is then connected to the carriage block of the linear module driven by the step motor.
  • the motor controller is connected to the computer through the real-time controller area network (CAN) bus, where the motor status can be recorded by the computer.
  • CAN controller area network
  • the bit rate on the CAN bus is set at 1 Mbps (megabits per second).
  • wavelength integer ambiguity and/or multipath effects can be further mitigated by techniques such as frequency diversity, channel coherence, and/or angle-of-arrival (AoA) variation.
  • the reader and tag antennas would be detuned with different gain and phase offset, where we used one calibration point (denoted as position 0) to cancel the initial phase offset.
  • position 0 the tag under test moved within the given media but the reader antenna remained stationary with respect to the other boundaries of the setup, this calibration was sufficient for all subsequent ranging measurements.
  • unknown inhomogeneity in the media, direct blockage of LoS, and reader location changes without new calibration will make our present system fail in terms of its performance in precision and accuracy, similar to other RF methods.
  • the phase information of the RF backscatter signal can offer accurate ranging of the transponder tag, which modulates its identification (ID) code on the RF signal to differentiate against other non-specific ambient reflection and inter-tag interference.
  • ID identification
  • the phase noise and transmitter/receiver (Tx/Rx) synchronization hence determine the ranging accuracy.
  • the conventional EPC Gen2 (electronic production code generation 2) RFID system however suffers high phase noises. As shown in Figure 9(a), the Tx spectrum is around f c .
  • the phase noise of the Tx signal is shown as the blue skirt around the injected local oscillator (LO) carrier.
  • the Tx signal leaks to Rx at the same f c .
  • the backscattered signal from the tag is modulated, so the spectrum is slightly offset from the carrier frequency, shown as the double sideband (DSB, orange arrows) on the Rx spectrum.
  • DSB double sideband
  • the reader needs to transmit relatively high power to wake up the passive tag, increasing the overall power level of the phase noise skirt.
  • the received backscattered signal will often suffer very high noise level from the low signal-to-interference ratio (SIR).
  • SIR signal-to-interference ratio
  • a harmonic backscattering system is employed, as shown in Figure 9(b).
  • the Tx signal spectrum is around f c , transmitted through the Tx antenna after the low-pass filter (LPF) to the harmonic transponder as the downlink.
  • the harmonic transponder receives and harvests the downlink signal at f c , which is also converted to the 2 nd harmonic signal at 2f c as the backscattering uplink to be received by the reader Rx antenna.
  • FIG. 9(c) shows the Rx signal chain of the harmonic receiver.
  • the signal goes through the high pass filter (HPF) and then amplified by the low noise amplifier (LNA).
  • the 2 f c LO is synthesized from the same clock source of the Tx LO and is hence coherent to Tx.
  • the down-converted quadrature signals are digitized by the analog-to-digital converter (ADC), and passed to the digital system for further processing.
  • ADC analog-to-digital converter
  • phase-based ranging which is affected by the LO performance in addition to external phase noise.
  • Figure 11(a) shows four phase-noise curves of different LOs.
  • LO 2 is the phase noise curve close to 1 GHz synthesized from the atomic clock reference applied in other experiments. Others are the phase noise curves with worse (LO 1) and better (LOs 3 and 4) performance for comparison.
  • the phase noise can also be expressed as the phase jitter, which is usually described as the zero-crossing jitter.
  • the root mean square (RMS) phase jitter can be evaluated by
  • the simulated ranging variations under four window sizes of 1, 5, 20, and 100 are 41.4, 14.8, 7.3, and 3.2 pm at the 75 th percentiles, and 82.9, 74.4, 40.7, and 17.7 pm at the maximum, respectively.
  • the moving average is effective to reduce the ranging variation caused by the white phase jitter.
  • the moving average may be equivalent to a low-pass finite impulse response (FIR) filter.
  • the penalty of applying a broader window width is the limiting ranging frequency response BW rr , which is equal to half of the ranging sampling rate.
  • Figure 12(b) shows how BW rr affects the frequency response of each window size.
  • the window size increases, the ranging variation is reduced but BW rr also decreases.
  • the ranging sampling rate can be easily maintained above several kHz as the ADC (analog-to-digital converter) is often in the MHz level. Moving average provides more efficient computation than the digital filter.
  • an n th -order digital FIR filter will need at least n multiplication-addition operations for each real-time cycle (e.g, sub milliseconds for kHz sampling), which can be a significant computation cost especially when the locating system employs the multi-static structure (multiple Tx/Rx pairs), large numbers of transponders, and multiple frequencies for wavelength ambiguity resolution. In moving average, it is not necessary to first calculate the sum within the window and then obtain the average. In practical, it is not necessary to first calculate the sum within the window and then obtain the average. In practical
  • the system buffers the initial m data in a FIFO (first in first out) queue, and calculates an initial sum value.
  • the system obtains a new ranging entry, adding it to the sum value, and stores it into the FIFO.
  • the output of the FIFO is the oldest ranging entry, which is subtracted from the sum value.
  • the moving average can be obtained by a simple division. After the first m cycles, the output rate becomes stable for every sampling cycle.
  • the total operations in one cycle are 2 floating-point additions and 1 multiplication, which can be easily operated in real time for the multichannel, multi -frequency and multi-transponder system.
  • the carrier frequency f c can be utilized to calculate the wave number or wavelength. Inaccurate f c in the hardware system will directly introduce a ranging offset, which also accumulates along the ranging distance. In the LO specifications, ppm (parts per million) or ppb (parts per billion) describes the frequency inaccuracy. The ppm deviation can be converted to the maximum frequency difference by Eq. (2): f c ppm
  • the receiver is designed with the quadrature structure, where the phase is calculated from I/Q (in phase/quadrature).
  • the received signal is split to two RF mixers for down conversion, and sampled by two ADC’s.
  • the imbalance of the I/Q signal is another frequency- related ranging error source.
  • Figure 13(b) shows the ranging errors caused by the I/Q imbalance with different downlink carrier frequencies. It can be seen that even though the imbalance is just about 1%, the ranging error of 600, 1000, 1400, and 1800 MHz can be as high as 115, 102, 85, and 64 pm, respectively. Hence, improved results can be obtained with the I/Q imbalance calibration, which can be done using the processor.
  • IF sampling is introduced, where the I/Q signals sampled by the ADC are not at the DC band.
  • the ADC clock aperture jitter should also be considered, which causes additional ranging variation.
  • the ADC aperture jitter is usually described in the time domain.
  • the digitized signal phase variation and the resulting ranging error depend on the input signal frequency.
  • the ranging variation dR caused by the ADC jitter and the SNR of the ADC, SNR ADC can be described as: where t jitter is the ADC clock jitter (in second), f IF is the IF frequency sampled by the ADC, and X h is the wavelength of the uplink carrier.
  • Figure 14 shows the RMS ranging variation caused by the ADC RMS aperture jitter with various f IF choices.
  • SNR ADC is constant at 73 dB.
  • fIF is fixed
  • the ranging variation increases with the ADC jitter.
  • the ADC jitter is constant
  • the ranging variation depends on f c and f IF.
  • the aperture jitter is around 200 femtoseconds (fs) to 1 picosecond (ps).
  • Figures 13(a) and 13(b) are the simulation results with the downlink frequency of 1 GHz and 1.5 GHz, respectively. Under the same SNR condition, higher f c provides better ranging variation due to shorter h.
  • FIG. 15 shows the tag antenna in water when the experiments were conducted.
  • Figure 15(a) is the photograph of the tag antenna. Because the 25-pm movement is too small comparing to the size of the antenna, a 60-mm (full-frame equivalent) macro lens was applied to magnify the image. After the digital zoom-in, the white-boxed area in Figure 15(a) is shown in Figure 15(b). The high-contrast vertical edge of the connector was used as the moving marker, and a 50-pm scale bar was added close to the vertical edge (the white box in Figure 15(b)) to clearly illustrate the moving steps. The round shape bubble on the connector in Figure 15(b) is an air bubble.
  • the high permittivity of the media can enhance the ranging resolution.
  • the measured ranging distance change is AD m , which is related to the phase change A ⁇ pD caused by tag movement distance AD and media permittivity e r , phase change Afh caused by hardware noise, and the wavelength in the media l e
  • Equation (4) clearly shows the linear relationship of the ranging distance and the phase when the wavelength ambiguity is not considered.
  • Figure 16 shows the experimental ranging results of 50-pm stepping in various medias of air, sand and water, where (a), (b), and (c) are in air; (d), (e), and (f) in sand; (g), (h), and (i) in water.
  • the transient data are shown in (a), (d) and (g), and the probability density curves of each ranging data are shown in (b), (e), and (h).
  • the Tx power from the tag is -15 dBm.
  • the received power at the reader Rx will be -52 dBm.
  • the base bandwidth is set at 100 kHz
  • the received power of the tag will be -12 dBm and the Rx SNR will drop to 42 dB, which will reduce the raw data accuracy but can still be mitigated by the moving average method due to the still high SNR.
  • the simplified link budget analysis above is based on ideal devices and setup without consideration of variations and strong interferences.
  • the received power at the tag and the SNR at the reader Rx can be further reduced in realistic scenarios.
  • the present estimate of SNR at 42 dB still has room to give, as many ranging receivers can be reasonably operated with SNR > 20 dB.
  • the present concepts include a radio-frequency method for range finding, the method comprising the acts of modulating an analog signal having an intermediate frequency, f IF , to a carrier frequency, f c , using a clock signal, transmitting the modulated signal to a tag using a transceiver, the tag being located at a distance from the transceiver, receiving an uplink signal backscattered from the tag, the uplink signal being modulated at a harmonic of the carrier frequency, demodulating the uplink signal using the clock signal and calculating a distance between the tag and the transceiver based on a phase of the demodulated uplink signal.
  • this method further includes modulating the analog signal to one or more additional carrier frequencies, f c2 ... / cm , each of the one or more additional carrier frequencies derived from a common accurate reference clock signal, and wherein each of the additional carrier frequencies is not equal to f c , transmitting, using a corresponding one or more additional transceivers, the one or more additional modulated signals to the tag, receiving one or more additional uplink signals backscattered from the tag, each of the one or more additional uplink signals being modulated at a second harmonic of a corresponding one of the one or more additional carrier frequencies, demodulating each of the one or more additional uplink signals using the clock signal of the corresponding one or more carrier frequencies, calculating a distance between the tag and each additional transceiver based on a difference between a phase of the analog signal and a phase of a corresponding one of the one or more demodulated uplink signals, and choosing optimal f IF and / c ’s to minimize the phase noise and uncertainties arising from the F
  • the present concepts includes a radio-frequency system for range finding, including a transceiver configured to modulate an analog signal having an intermediate frequency, f IF , to a carrier frequency, f c , using a clock signal, transmit the modulated signal, receive a backscattered uplink signal from a tag, wherein the uplink signal is transformed to modulated at a harmonic frequency of the carrier frequency and modulated with a digital code, and demodulate the uplink signal using a coherent reference clock signal, wherein the radio-frequency system further includes a processor configured to receive the demodulated uplink signal and calculate a distance between the tag and the transceiver using a phase of the demodulated uplink signal in which the coherent reference clock signal relates to the local oscillator of the receiver being derived from the same reference clock in the transmitter. In at least some aspects, the radio-frequency system processor is further configured to repeatedly calculate the distance between the tag and the transceiver at a sampling rate between 20 Hz - 10 kHz

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Abstract

Un procédé radiofréquence de télémétrie comprend la modulation d'un signal de référence ayant une fréquence intermédiaire en un signal de liaison descendante ayant une fréquence porteuse à l'aide d'un signal d'horloge. Le signal de liaison descendante est transmis à une étiquette à l'aide d'un émetteur-récepteur. Un signal de liaison montante rétrodiffusé à partir de l'étiquette est reçu et démodulé à l'aide du signal d'horloge. Le signal de liaison montante a une fréquence qui est une harmonique de la fréquence porteuse. Une distance entre l'étiquette et l'émetteur-récepteur est calculée sur la base d'une phase du signal de liaison montante démodulé. L'invention concerne un système de télémétrie comprenant un émetteur-récepteur et un processeur. L'émetteur-récepteur module un signal de référence en un signal de liaison descendante et transmet le signal de liaison descendante. L'émetteur-récepteur reçoit et démodule un signal de liaison montante. Le processeur est configuré pour recevoir le signal de liaison montante démodulé et calculer une distance entre l'étiquette et l'émetteur-récepteur à l'aide d'une phase du signal de liaison montante démodulé.
PCT/US2019/059274 2018-10-31 2019-10-31 Système et procédé de télémétrie ultra-haute résolution à l'aide d'une identification par radiofréquence Ceased WO2020092849A1 (fr)

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EP3874298A1 (fr) 2021-09-08
CA3117204A1 (fr) 2020-05-07
US20210373111A1 (en) 2021-12-02

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