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WO2019001179A1 - Alimentation électrique de commutateur à transfert indirect - Google Patents

Alimentation électrique de commutateur à transfert indirect Download PDF

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Publication number
WO2019001179A1
WO2019001179A1 PCT/CN2018/088025 CN2018088025W WO2019001179A1 WO 2019001179 A1 WO2019001179 A1 WO 2019001179A1 CN 2018088025 W CN2018088025 W CN 2018088025W WO 2019001179 A1 WO2019001179 A1 WO 2019001179A1
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Prior art keywords
primary winding
channel fet
capacitor
power supply
winding
Prior art date
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PCT/CN2018/088025
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English (en)
Chinese (zh)
Inventor
王保均
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Mornsun Guangzhou Science and Technology Ltd
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Mornsun Guangzhou Science and Technology Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from DC input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to the field of switching power supplies, and in particular to a flyback switching power supply.
  • the switching power supply is widely used.
  • the fly-back switching power supply has a beautiful advantage: the circuit
  • the topology is simple and the input voltage range is wide. Since the number of components is small, the reliability of the circuit is relatively high, so the application is wide.
  • many documents are also known as flyback switching power supplies, flyback power supplies, and flyback converters. In Japan and Taiwan, they are also called flyback converters, flyback switching power supplies, and flyback power supplies.
  • a common topology for an AC/DC converter is shown in Figure 1. The prototype is from the "Switching Power Supply Converter Topology and Design" by Dr. Zhang Xingzhu, ISBN 978-7-5083-9015-4.
  • the rectifier bridge 101, the filter circuit 200, and the basic flyback topology unit circuit 300 are also referred to as the main power stage.
  • the practical circuit is also provided with a varistor, an NTC thermistor, and an EMI (Electromagnetic Interference) in front of the rectifier bridge. ) Protect the circuit to ensure that the electromagnetic compatibility of the flyback power supply meets the requirements for use. Under normal circumstances, the flyback switching power supply requires that the leakage inductance between the primary and secondary windings is as small as possible, so that the conversion efficiency is high, and the withstand voltage of the primary side main power switch V is also reduced, for using the RCD network as a go Magnetic and absorbing flyback converters also reduce losses in the RCD network.
  • RCD absorption refers to the absorption circuit composed of resistors, capacitors and diodes.
  • the literature in China is the same as the international one.
  • the letter R is used to give the resistance number and represents the resistance.
  • the letter C is used to number the capacitor and represent the capacitor.
  • the letter D is used to give the diode. Numbered and represents the diode, the resistor and capacitor are connected in parallel, and then connected in series with the diode to form an RCD network.
  • the rectifier bridge 101 is generally composed of four diodes. When there is no rectifier bridge 101, 200, 300 can constitute a DC/DC switching power supply or converter. Because it is DC power supply, there is no power factor requirement, and the power can be more than 75W. . In fact, the use of flyback topology in low-voltage DC/DC switching power supplies is not the mainstream. This is because the input current of the flyback power supply is discontinuous and the ripple is large at low voltage, which requires higher requirements for the power supply equipment of the former stage.
  • Low-voltage DC/DC switching power supplies generally refer to input voltages below 48V. Some low-voltage DC/DC switching power supplies can operate up to 160V DC, such as railway power supplies.
  • the inductance of the primary winding is also low. It is often found that the calculated number of turns cannot be tiled from the left to the right of the slot of the full frame. Especially when the working voltage is high, the sandwich series winding method can be used. Under the voltage, the sandwich parallel winding method is forced. Since the two primary windings are not in the same layer, there is a leakage inductance between the two primary windings, and the leakage inductance will cause loss, thereby making the efficiency of the switching power supply variable.
  • the advantage is non-destructive demagnetization, the efficiency is higher, but the choice of the wire diameter of the third winding is also a problem: the selection is relatively thin, and the winding of the primary winding is more troublesome, easy to put The thin wire is broken; if the same wire diameter is selected as the primary winding, the cost is high.
  • the third winding demagnetization flyback converter is also referred to as a "three-winding absorption flyback converter".
  • the energy of the excitation may be directly returned from the DC power source U DC by D1, instead of appearing in the secondary winding N S , resulting in There is no current in the secondary side D2, so that the output voltage is low or no output; and the reflected voltage generated when D2 is turned on cannot be greater than the DC power supply U DC , and if the duty ratio cannot be greater than 0.5, the power density cannot be further improved.
  • the demagnetization circuit itself is a more classic topology, the duty cycle can be greater than 0.5, but the leakage inductance energy is not recycled.
  • the inventors have defined the topology used in the flyback switching power supply with the Chinese application numbers 201710142832.0 and 201710142797.2, respectively, and will include the forward topology using the inventive concept, and the basic topology excluding the demagnetization mode is defined.
  • the LCL converter it consists of two primary-side magnetizing inductances and a capacitor connected in series with them.
  • Such as LCL flyback converter also refers to LCL flyback switching power supply.
  • the present invention solves the above-mentioned shortcomings of the existing low-voltage LCL flyback switching power supply, and provides a flyback switching power supply, which requires loose leakage between the primary and secondary windings and achieves demagnetization.
  • the energy recovery of the circuit further realizes zero voltage switching of the main power switch tube, further reducing losses and improving conversion efficiency.
  • a flyback switching power supply including a transformer, a first N-channel FET, a first capacitor, a second capacitor, a first diode, a clamp network, and a transformer including a first primary winding, a second primary winding and a secondary winding, the clamping network comprises at least an anode and a cathode, and the secondary winding is connected to the first diode anode, the first diode cathode and the second capacitor One end is connected, and the output is positive, the second side of the secondary winding is connected with the other end of the second capacitor, and the output is negative; the positive end of the input DC power supply is simultaneously connected with the cathode of the same name of the first primary winding and the cathode of the clamp network, a primary winding different name end is connected to the drain of the first N-channel FET; the anode of the clamp network is connected to the second end of the second primary winding, and the source of the first N-channel FET
  • One end of the third capacitor is the cathode of the clamp network, the other end of the third capacitor is connected to the drain of the second N-channel FET, and the source of the second N-channel FET is the anode of the clamp network, a gate connection clamp control signal of the two N-channel FET;
  • one end of the third capacitor is the anode of the clamp network, the other end of the third capacitor is connected to the source of the second N-channel FET, and the drain of the second N-channel FET is the cathode of the clamp network, The gate of the two N-channel FET is connected to the clamp control signal.
  • the first N-channel FET can be replaced by a P-channel FET, and the body diode inside the P-channel FET is in the same polarity as the body diode inside the first N-channel FET.
  • the present invention also provides an equivalent solution of the foregoing solution 1.
  • the second embodiment of the present invention can also be implemented.
  • a flyback switching power supply includes a transformer, a first N-channel FET, a first capacitor and a second capacitor.
  • the transformer includes a first primary winding, a second primary winding, and a secondary winding
  • the clamping network includes at least an anode and a cathode
  • the secondary winding has a different name and a first diode
  • the anode of the tube is connected, the first diode cathode is connected to one end of the second capacitor, and the output is positive, the second end of the secondary winding is connected with the other end of the second capacitor, and the output is negative;
  • the positive end of the input DC power supply is simultaneously
  • the drain of the N-channel field effect transistor and the second primary winding are connected at different ends, and the source of the first N-channel field effect transistor is connected to the same name end of the first primary winding; the second primary winding has the same name end and the clamp network
  • the cathode of the first primary winding is connected to the anode of the clamp network, and the connection point is simultaneously connected to the negative terminal of the input DC power supply; the gate of the first N-channel F
  • One end of the third capacitor is the cathode of the clamp network, the other end of the third capacitor is connected to the drain of the second N-channel FET, and the source of the second N-channel FET is the anode of the clamp network, a gate connection clamp control signal of the two N-channel FET;
  • one end of the third capacitor is the anode of the clamp network, the other end of the third capacitor is connected to the source of the second N-channel FET, and the drain of the second N-channel FET is the cathode of the clamp network, The gate of the two N-channel FET is connected to the clamp control signal.
  • the first N-channel FET can be replaced by a P-channel FET, and the body diode inside the P-channel FET has the same polarity as the body diode inside the first N-channel FET.
  • the first primary winding and the second primary winding have the same wire diameter.
  • the physical path of the excitation current of the first primary winding and the second primary winding is reversed in the PCB layout.
  • the beneficial effects of the invention are: allowing the leakage inductance between the primary and secondary windings to be large, and the energy of the leakage inductance is recycled by the clamp network, and the duty ratio can be greater than that by those skilled in the art by selecting an appropriate control strategy. 0.5, the power density is high, and the conversion efficiency is not reduced, and the zero voltage switch of the switch tube can be realized, thereby further improving the conversion efficiency.
  • FIG. 1 is a schematic diagram of a conventional flyback switching power supply for alternating current to direct current
  • FIG. 2 is a schematic diagram of a disclosed technical solution of Chinese Application No. 201710142832.0;
  • Figure 3 is a schematic diagram of the disclosed technical solution of Chinese Application No. 201710142797.2;
  • 4-1 is a schematic diagram of a first embodiment of the present invention, and the clamp network adopts (1) mode;
  • 4-2 is a second schematic diagram of the first embodiment of the present invention, and the clamp network adopts (2) mode;
  • 4-3 is a schematic diagram showing the generation of two excitation currents 41, 42 when Q1 is saturated in the first embodiment
  • 4-4 is a schematic diagram showing the Q1 cutoff in the first embodiment, generating a freewheeling current 43 and a demagnetizing current 44;
  • 5-1 is a schematic diagram of a second embodiment of the present invention, and the clamp network adopts (1) mode;
  • FIG. 5-2 is a second schematic diagram of the second embodiment of the present invention, and the clamp network adopts the (2) mode.
  • FIG. 4-1 and 4-2 illustrate a schematic diagram of a flyback switching power supply according to a first embodiment of the present invention, including a transformer B, a first N-channel FET Q1, a first capacitor C1, and a second capacitor.
  • C2 a first diode D2, a clamp network 400, and a transformer B includes a first primary winding N P1 , a second primary winding N P2 and a secondary winding N S
  • the clamp network 400 includes at least an anode and a cathode, and a pair
  • the edge of the side winding N S is connected to the anode of the first diode D2, and the cathode of the first diode D2 is connected to one end of the second capacitor C2, and forms an output positive, which is the + end of Vout in the figure, and the secondary winding N S
  • the same name end is connected to the other end of the second capacitor C2, and forms an output negative, which is the end of Vout in the figure; the positive end of the input DC
  • One end of the third capacitor C3 is the cathode of the clamp network 400, the other end of the third capacitor C3 is connected to the drain d of the second N-channel FET Q2, and the source s of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal, as shown in FIG. 4-1;
  • One end of the third capacitor C3 is the anode of the clamp network 400, the other end of the third capacitor C3 is connected to the source s of the second N-channel FET Q2, and the drain d of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal, as shown in Figure 4-2.
  • the anode and cathode of the clamp network 400 and the body diode of the second N-channel FET Q2 therein are corresponding.
  • the anode of the body diode of Q2 is the anode of 400.
  • the cathode of the body diode of Q2 is the cathode of 400 after passing C3.
  • the cathode of the body diode of Q2 is the cathode of 400, and the anode of the body diode of Q2 is the anode of 400 after passing C3.
  • Heterogeneous end one end of the winding in the figure where there is no black mark
  • Driving control signal including various pulse waves such as PWM pulse width modulation signal and PFM pulse frequency modulation;
  • Clamp control signal includes various square waves such as PWM pulse width modulation signal and PFM pulse frequency modulation, but appears differently from the drive control signal;
  • Transformer B the first primary winding N P1 and the second primary winding N P2 are in the figure, the cores are connected by a broken line, indicating that they are wound around a transformer and share the same core, not a separate transformer, just for The graphics are clear and the connection relationship is simple, and the drawing method in the figure is used.
  • the source of the N-channel FET Q1 is connected to the same end of the second primary winding N P2 , and the connection point is simultaneously connected to the negative terminal of the input DC power supply U DC - that is, the FET
  • the source of Q1 is connected to the negative terminal of the input DC power supply U DC - which does not exist directly in practical applications. This is because in the field of switching power supply, the analysis of the working principle of the basic topology will omit unnecessary factors.
  • the source of the FET is connected to a current sense resistor or a current transformer to detect the average current or peak current to implement various control strategies.
  • the current sense resistor or current transformer is connected to the source.
  • the current transformer can appear anywhere in the excitation circuit, such as the drain of a FET, such as the same or different end of the first primary winding, and the current transformer has a conventional primary side. It is also a Hall sensor that is a "wire" and a magnetic core transformer whose secondary side is a multi-turn coil.
  • the charging current of the first primary winding N P1 is: flowing from the same name end to the different name end; the charging current of the second primary winding N P2 is: flowing from the different name end to the same name end; N P1 and N P2 are double lines and Winding, the two currents are equal in magnitude, and the generated magnetic flux is opposite, completely canceled. That is, at the time of power-on, the power supply U DC charges C1 through the two windings of the transformer B. These two windings cancel out due to the mutual inductance, and do not work.
  • C1 is equivalent to the DC internal resistance of N P1 and N P2 in parallel with the power supply U DC .
  • C1 still functions as power supply filtering and decoupling; as time passes, the terminal voltage of C1 is equal to the voltage of U DC , left positive And right negative.
  • Q1 When Q1 receives the control signal normally, taking one cycle as an example, when the gate of Q1 is high, Q1 is saturated and its internal resistance is equal to the on-state internal resistance R ds(ON) . For the convenience of analysis, this is the case. It is regarded as a straight-through, which is a wire. As shown in Figure 4-3, Q2 is in the off state and does not participate in the work. In the figure, 400 is drawn as an open state; at this time, two excitation currents are generated, 41 in Figure 4-3. And 42;
  • the excitation currents of 41 and 42 are in parallel. Since the inductances of N P1 and N P2 are the same, the excitation voltages are the same, and they are equal to U DC , 41 and 42 are completely equal.
  • the secondary winding N S is pressed.
  • the induced voltage is the same.
  • the induced voltage is: a positive voltage is induced at the same name, and a negative voltage is induced at the opposite end.
  • the magnitude is equal to U DC multiplied by the turns ratio n, that is, N S induces a positive and negative voltage.
  • N S induces a positive and negative voltage.
  • the currents of 41 and 42 increase linearly upward; the current direction flows from the same name end to the different name end in the inductance;
  • the gate of Q1 changes from high level to low level, Q1 also turns from saturation conduction to off. Since the current in the inductor cannot be abrupt, even though Q1 is off at this time, the currents of 41 and 42 still flow from the same name end. At the opposite end, since the current loop of the primary side has been cut, the energy in the core flows from the same name to the opposite end on the secondary side. Referring to Figure 4-4, the secondary winding N S appears to flow from the same end to the different end.
  • the initial magnitude of the current (the sum of 41 and 42 at the instant of Q1 turn-off) / ⁇ ratio n, which causes D2 to conduct a forward conduction and pass the forward-directed D2 to Capacitor C2 is charged and Vout establishes voltage or continuously outputs energy. This process is also the process of demagnetization.
  • the output of the flyback switching power supply is named after the primary winding is disconnected from the power supply.
  • the transformer B is not the function of the voltage conversion, but is the isolated version of the Buck-Boost converter. Therefore, transformer B is often referred to as a flyback transformer;
  • the circuit for demagnetizing the leakage inductance of the present invention is composed of a clamp network 400 composed of Q2 and C3 and a second primary winding N P2 , and the working principle is:
  • the first primary winding N P1 and the second primary winding N P2 are double-wired and wound, and the leakage inductance between the two windings is zero. At the instant of Q1 turn-off and after, the energy on the leakage inductance is not transmitted to the secondary
  • the electric energy of the leakage inductance in the second primary winding N P2 is in the same direction as the direction of the excitation, flowing from the same end to the opposite end, that is, in FIG. 4-4, flowing from bottom to top, opening the body of Q2 a diode, a current flowing from the source s of Q2 to the drain d, and this electrical energy is charged to C3 to form a leakage inductance demagnetization current as indicated by 44;
  • the leakage energy of the first primary winding N P1 is coupled to the second primary winding N P2 without leakage inductance, and is demagnetized by the body diode of Q2, and the leakage inductance demagnetization current shown by 44 is also formed. ;
  • the output voltage Vout is divided by the turns ratio n, which is the "reflected voltage" formed on the primary side when the secondary winding N S is turned on at D2. Since there is C3 blocking, the reflected voltage is greater than the value of the DC power source U DC , the circuit It can also work normally.
  • C2 is equivalent to a voltage source, which is "excited” to the secondary winding N S and the primary side forms a "reflected voltage".
  • the primary winding is equivalent to a voltage source having a voltage equal to the reflected voltage and
  • the leakage inductance is connected in series, and the current in D2 drops to zero, and D2 turns off, and the primary winding is restored to the series connection of the magnetizing inductance and the leakage inductance.
  • ZVS Zero voltage switch
  • the terminal voltage of C3 will be close to or equal to twice the U DC voltage at a certain time, and it is up and down, due to the terminal voltage of C1. It is always left positive and right negative, and equal to U DC . At this moment, the left terminal voltage of C1 is zero volt, that is, the terminal voltage of Q1 is also zero volt. If Q1 is saturated at this moment, then zero of Q1 is realized. The voltage is turned on, and this mode must be the current interrupt mode, and the time when Q1 is turned on is extremely easy to detect.
  • the wire diameters of the first primary winding and the second primary winding are the same, so that the winding is convenient, the wire diameters described herein are the same, and they are all of the same size Litz wire, the color can be Different, that is, multi-strand stranding, for the convenience of identification, the same specification wire including the Litz wire can have different colors. As the operating frequency increases, the high frequency current tends to flow on the surface of the enameled wire. In this case, the Litz wire can solve this problem.
  • the invention has many differences, mainly: allowing the leakage inductance between the primary and secondary windings to be large, and the leakage inductance energy is recycled by the clamp network, thus achieving high efficiency. . It also increases the current density of the primary winding, increases the power density of the converter, and is suitable for applications with lower operating voltages.
  • a flyback switching power supply includes a transformer B, a first N-channel FET Q1, a first capacitor C1, a second capacitor C2, a first diode D2, a clamp network 400, and a transformer B including a first primary winding N P1 , a second primary winding N P2 , and a secondary winding N S , a clamp network
  • the 400 includes at least an anode and a cathode, and the secondary winding N S is connected to the anode of the first diode D2, and the cathode of the first diode D2 is connected to one end of the second capacitor C2, and forms an output positive, which is Vout in the figure.
  • the secondary winding N S has the same name end connected to the other end of the second capacitor C2, and forms an output negative, which is the end of Vout in the figure; the positive terminal of the input DC power supply U DC + simultaneously with the N-channel FET Q1
  • the drain and the second primary winding N P2 are connected to each other.
  • the source of the N-channel FET Q1 is connected to the same end of the first primary winding N P1 ; the second primary winding N P2 is the same name and the clamp network 400 is connected to the cathode, the anode connected to a first primary winding N P1 dotted end of the clamp network 400, while the connection point of the DC input connector Source U DC negative terminal; a connection control signal gate N-channel MOSFET Q1; N Pl a first primary winding and a second primary winding P2 is N bifilar, further comprising a first capacitor C1, a first One end of the capacitor C1 is connected to the same end of the first primary winding N P1 , and the other end of the first capacitor C1 is connected to the same end of the second primary winding N P2 .
  • the clamping network 400 includes at least a third capacitor C3 and a second N-channel.
  • the MOSFET Q2, the third capacitor C3 and the second N-channel FET Q2 are connected in series, and the series connection is one of the
  • One end of the third capacitor C3 is the cathode of the clamp network 400, the other end of the third capacitor C3 is connected to the drain d of the second N-channel FET Q2, and the source s of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal, as shown in FIG. 5-1;
  • One end of the third capacitor C3 is the anode of the clamp network 400, the other end of the third capacitor C3 is connected to the source s of the second N-channel FET Q2, and the drain d of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal as shown in Figure 5-2.
  • the second embodiment is a modification of the first embodiment: on the basis of FIG. 4-1 of the first embodiment, the series devices of the two excitation circuits are interchanged, that is, the positions of N P1 and Q1 are interchanged. At the same time, the clamp network 400 and N P2 are interchanged, and C1 is still connected between the connection points of the two series devices, and the circuit of FIG. 5-1 of the second embodiment is obtained. Since the source voltage of Q1 is fluctuating, Therefore, this circuit is floating drive, but obtains the direct drive of the second N-channel FET Q2 for clamping. On the basis of Figure 5-1, C3 and Q2 in the clamp network 400 are interchanged. Position, you can get the circuit of Figure 5-2.
  • the terminal voltage of C1 is equal to the voltage of UDC, right and left negative;
  • the first way is: the positive end of the power supply UDC enters through the drain of Q1, the source of Q1 is out, and then enters the same name of the first primary winding NP1, and the different name of NP1 is output, and returns to the negative end of the power supply UDC;
  • the second way is: the right positive end of the capacitor C1 passes through the same name of the second primary winding NP2, the opposite end of NP2 is output, the drain of Q1 enters, the source of Q1 comes out, and returns to the left negative end of the capacitor C1;
  • the first and second excitation currents are in parallel relationship. Since the NP1 and NP2 have the same inductance and the same excitation voltage, they are equal to UDC, and the two paths are completely equal.
  • the secondary winding NS is the same as the ⁇ ratio. The induced voltage is generated, the positive voltage is induced by the same name, and the negative voltage is induced by the different name. The size is equal to UDC multiplied by the turns ratio n, that is, NS induces a positive and negative voltage. This voltage is connected in series with the terminal voltage of C2. At both ends of D2, D2 is reversed and not turned on. At this time, the secondary side is equivalent to no load, and there is no output;
  • the first and second excitation currents increase linearly upward; the current direction flows from the same name end to the different name end in the inductance;
  • the circuit for demagnetizing the leakage inductance is composed of the clamp network 400 and the second primary winding NP2, and the working principle is:
  • the energy on the leakage inductance is not transmitted to the secondary side, and the electrical energy of the leakage inductance in the second primary winding NP2 is in the same direction as the direction of the excitation, flowing from the same name to the opposite end.
  • the Q2 body diode in the clamp network 400 is turned on to charge C3, and this electric energy is absorbed by C3 to form a leakage inductance demagnetization current loop;
  • the leakage energy of the first primary winding NP1 is coupled to the second primary winding NP2 without leakage inductance, and is demagnetized by the clamp network 400 to form a leakage inductance demagnetization current loop;
  • the second embodiment is a modification of the first embodiment, and the working principle is equivalent, and the object of the invention is also achieved.
  • Q2 can be replaced with a P-channel FET. It is necessary to ensure that the body diode inside the P-channel FET is in the same direction as the body diode in Figure 5-1 or Figure 5-2.

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  • Power Engineering (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • Dc-Dc Converters (AREA)

Abstract

L'invention concerne une alimentation électrique de commutateur à transfert indirect, qui, sur la base d'un convertisseur indirect LCL, maintient une extrémité de polarité commune d'un premier enroulement primaire (NP1) d'un transformateur (B) connectée à l'extrémité positive d'une alimentation électrique, et une extrémité de polarité commune d'un second enroulement primaire (NP2) connectée à l'extrémité négative de l'alimentation électrique ; le premier enroulement primaire et le second enroulement primaire sont des enroulements parallèles à double fil, et une extrémité d'un premier condensateur (C1) est connectée à une extrémité de polarité opposée du premier enroulement primaire, tandis qu'une autre extrémité est connectée à une extrémité de polarité opposée du second enroulement primaire ; l'extrémité de polarité opposée du premier enroulement primaire est connectée à l'alimentation électrique au moyen d'un premier transistor à effet de champ (Q1), et l'extrémité de polarité opposée du second enroulement primaire est connectée à l'alimentation électrique au moyen d'un réseau de serrage (400) dans lequel un second transistor à effet de champ (Q2) est connecté en série à un troisième condensateur (C3). Lorsque le premier transistor à effet de champ conduit de manière saturante, le premier enroulement primaire et le second enroulement primaire sont tous deux excités, et lorsque le premier transistor à effet de champ est fermé, un enroulement secondaire (NS) du transformateur délivre de l'énergie, tandis que l'énergie provenant de l'inductance de fuite réalise un serrage actif au moyen d'un passage à travers le réseau de serrage depuis le second enroulement primaire. Après sélection d'une stratégie de commande appropriée, une commutation à tension nulle (ZVS), une absorption sans perte et un cycle de service supérieur à 0,5 peuvent être obtenus, tout en ayant une densité de puissance élevée et une efficacité de conversion élevée.
PCT/CN2018/088025 2017-06-30 2018-05-23 Alimentation électrique de commutateur à transfert indirect Ceased WO2019001179A1 (fr)

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CN107294388B (zh) * 2017-06-30 2020-02-14 广州金升阳科技有限公司 一种反激式开关电源

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US4618919A (en) * 1984-10-04 1986-10-21 Sperry Corporation Topology for miniature power supply with low voltage and low ripple requirements
JPH08168244A (ja) * 1994-12-12 1996-06-25 Nemic Lambda Kk スイッチング電源装置
CN101686015A (zh) * 2008-09-23 2010-03-31 台达电子工业股份有限公司 具有主动箝位电路的正-反激变换器
CN101692595A (zh) * 2009-09-21 2010-04-07 山特电子（深圳）有限公司 一种有源箝位正-反激变换器
CN106655791A (zh) * 2017-03-10 2017-05-10 广州金升阳科技有限公司 一种反激式开关电源
CN107294388A (zh) * 2017-06-30 2017-10-24 广州金升阳科技有限公司 一种反激式开关电源

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