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WO2017138800A1 - Antenne unipolaire - Google Patents

Antenne unipolaire Download PDF

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Publication number
WO2017138800A1
WO2017138800A1 PCT/KR2017/001564 KR2017001564W WO2017138800A1 WO 2017138800 A1 WO2017138800 A1 WO 2017138800A1 KR 2017001564 W KR2017001564 W KR 2017001564W WO 2017138800 A1 WO2017138800 A1 WO 2017138800A1
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Prior art keywords
antenna
quasi
strip
mobius
bridge
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English (en)
Korean (ko)
Inventor
김미정
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Individual
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Individual
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Priority to EP17750492.5A priority Critical patent/EP3416241A4/fr
Priority to JP2018543379A priority patent/JP6583901B2/ja
Priority to US16/076,972 priority patent/US10283856B2/en
Publication of WO2017138800A1 publication Critical patent/WO2017138800A1/fr
Anticipated expiration legal-status Critical
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H01Q1/38—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith formed by a conductive layer on an insulating support
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
    • H01Q3/01—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the shape of the antenna or antenna system
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
    • H01Q3/26—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q7/00—Loop antennas with a substantially uniform current distribution around the loop and having a directional radiation pattern in a plane perpendicular to the plane of the loop
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/16—Resonant antennas with feed intermediate between the extremities of the antenna, e.g. centre-fed dipole
    • H01Q9/26—Resonant antennas with feed intermediate between the extremities of the antenna, e.g. centre-fed dipole with folded element or elements, the folded parts being spaced apart a small fraction of operating wavelength
    • H01Q9/27—Spiral antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/30—Resonant antennas with feed to end of elongated active element, e.g. unipole
    • H01Q9/40—Element having extended radiating surface

Definitions

  • the present invention relates to a monopole antenna, and more particularly to a monopole antenna applied to a wireless communication system.
  • the helical structure is not suitable for the technique of miniaturizing the antenna having a single resonance frequency characteristic as the resonant frequency occurs every rotation of the circumference, and the methods of applying the meta-material and the laminated structure are complicated. And there is a disadvantage that the manufacturing cost increases.
  • a small antenna according to the prior art uses a low cost flat patch antenna composed of a metal ground, a high dielectric constant, and a radiator.
  • a ceramic material having a high dielectric constant reduces the radiation efficiency of the reception antenna and generates an EMI (Electro Magnetic Intereference), thereby reducing the reception sensitivity of the receiver.
  • the small antenna according to the prior art had a limitation that the antenna installation space should be secured.
  • An object of the present invention is to provide a miniaturized monopole antenna capable of operating at low power by using an electrode pattern on a flexible substrate and improving radiation efficiency.
  • Another object of the present invention is to provide a monopole antenna capable of miniaturizing the antenna by using a quasi-Moebius strip structure and realizing high-performance antenna characteristics of broadband and high gain characteristics.
  • the monopole antenna according to the present invention is disposed in the center of the front surface of the dielectric substrate and comprises a plurality of loops formed of a structure in which the Mobius strip is cut along the circumference at least one time, each of the loops And a first bridge connecting one end sequentially and a second bridge connecting via holes formed at one end of the innermost loop and the outermost loop, respectively.
  • the effect of miniaturizing the UWB antenna to which the quasi-Moebius strip and the via hole structure are applied can be obtained.
  • the physical circumference is downsized by 1 / (N + 1) times, and the resonant frequency occurs every time the conventional helical antenna has one rotation of the circumference.
  • the radiation pattern in the Far-Field exhibits non-directional characteristics like a general monopole antenna, and the directivity can be controlled by adjusting the rotation angle of each ring provided in the radiator.
  • the resonance frequency and the reflection coefficient can be optimized by varying three parameters of the quasi-Mobius strip, that is, the thickness of the ring line of the quasi-Mobius strip, the width of the bridge, and the radius and position of the via hole.
  • the simulation results in the effect that the quasi-Moebius strip of the via hole structure can be applied to the matching network.
  • the thickness of the ring line and the size and position of the via hole affect the inductance L
  • the width and the size of the bridge and the size and position of the via hole affect the capacitance C.
  • the resonant frequency of the quasi-Mobius strip antenna is formed in the 2.4 GHz band, the peak value of the reflection coefficient S 11 is 17.3 dB, and the measurement result is 27.65 dB.
  • an optimized quasi-mobius strip is applied to an antenna, but each ring is composed of a plurality of ring lines having different radii, but the effect of having a single resonance frequency characteristic is obtained.
  • the spectral efficiency can be improved as the ECC value representing the correlation between the antennas is 0.02 or less in the frequency range of 3 GHz to 8 GHz. Effect is obtained.
  • the semi-mobius strip having a via hole structure can be applied to not only UWB antenna but also to an RF passive element such as an oscillator and a resonator.
  • FIG. 1 is a block diagram of a circular disk-shaped monopole antenna having coplanar waveguide line feeding implemented on a dielectric substrate;
  • 2a to 2c are conceptual views showing the surface current distribution in the 0 ° feed phase of each operating frequency of the circular disk monopole antenna
  • FIG. 3 is a configuration diagram of a circular double closed loop monopole antenna for miniaturizing a circular disk monopole antenna structure
  • 4A and 4B show input matching characteristics of an optimized simulated circular double closed loop monopole antenna
  • FIG. 5 is a block diagram of a quasi-Mobius monopole antenna
  • 6A and 6B show input matching characteristics of an optimized simulated quasi-Mobius monopole antenna
  • FIG. 7 is a diagram showing surface current distribution at a 90 ° feeding phase of a center frequency of a quasi-Mobius monopole antenna
  • FIG. 10 is a configuration diagram of a double circular ring monopole antenna for miniaturizing a circular disk monopole antenna structure
  • 11 is a graph of input matching characteristics of a circular disk monopole antenna structure and types 1 to 3;
  • 11 is a view illustrating two-dimensional radiation pattern characteristics of each comparison monopole antenna structure
  • FIG. 13 is a comparison of Mobius strips and strips cut along the circumference thereof;
  • 14a and 14b is a block diagram of a flat plate Mobius strip according to the prior art
  • FIG. 16 is a rear view of the monopole antenna shown in FIG. 15;
  • 21 is a configuration diagram of a monopole antenna to which a semi-Moebius strip is applied,
  • 22 is a graph showing a simulation result of resonant frequency and return loss according to the thickness change of a ring line
  • 24 is a graph showing the result of comparing the return loss between the ring 1,2,3 antennas having different radii and the quasi-Moebius strip antennas,
  • 25 is a view showing a via hole structure connecting two microstrip lines
  • 30A-30D illustrate an inductor implemented on a substrate
  • 31 and 32 are schematic views of the first and second quasi-Mobius strips, respectively;
  • 35 and 36 are front and rear views of an antenna to which a quasi-Mobius strip is applied according to a preferred embodiment of the present invention
  • 41 to 43 are views illustrating a state in which the radiators of the UWB antenna shown in FIG. 35 are rotated by 90 °, 180 °, and 330 °, respectively.
  • 49 to 51 are views showing a state in which each ring provided on the radiator of the quasi-Moebius strip rotated by 300 °, 330 °, 350 °, respectively,
  • 58 to 60 are graphs of three-dimensional radiation patterns to which the radiators illustrated in FIGS. 49 to 51 are applied.
  • FIG. 1 is a block diagram of a circular disk type monopole antenna having a coplanar waveguide transmission line (CPW TL) feeding implemented on a dielectric substrate.
  • CPW TL coplanar waveguide transmission line
  • the electrical properties of the dielectric substrate may be expressed as dielectric constant ( ⁇ r ), dielectric thickness (H), copper foil thickness (T), and loss tangent value (tan ⁇ ) of the dielectric substrate.
  • Coplanar waveguide line structure can be seen as, coplanar to the strip transmission line and a center strip line width of the slot in the CPW TL feeding structure having two ground plane on both sides (S g), the center strip line shown in Figure 1
  • the desired characteristic impedance value can be adjusted by changing the value of the width W f .
  • 2A-2C are conceptual diagrams showing the surface current distribution in the 0 ° feed phase of each operating frequency of a circular disk monopole antenna.
  • the surface current mainly flows around the outer edge of the circular disk, and almost no current flows in the center. Therefore, the center part can be partially removed to form an annular monopole antenna structure, which will be used for future miniaturization studies using multiple closed loop monopole antenna structures.
  • the surface current distributions of 3.0 GHz and 4.5 ⁇ form the same direction current distribution within the circular disk structure, but at 6.0 ⁇ the emitter is larger than the operating frequency, forming a higher order current distribution, which is horizontal in the azimuthal radiation pattern. Deteriorate to have a slightly vertical upward offset directivity characteristic in omnidirectional.
  • FIG. 3 is a configuration diagram of a circular double closed loop monopole antenna (hereinafter, referred to as 'type 1') for miniaturizing a circular disk monopole antenna structure.
  • 'type 1' a circular double closed loop monopole antenna
  • FIG. 4 is a diagram illustrating input matching characteristics of an optimized simulated circular double closed loop monopole antenna.
  • the input matching characteristics in the operating band are affected by the size of the outer loop, the line width of the closed loop, the feed match length and the size of the ground plane.
  • the simulated 10 dB reference return loss characteristic operates in the 3.6 to 7.3 dB band.
  • FIG. 5 is a schematic diagram of a quasi-Mobius monopole antenna (hereinafter referred to as “type 2”), and shows the quasi-Mobius monopole antenna structure and design parameters of the antenna for miniaturizing the circular disk monopole antenna structure.
  • type 2 a quasi-Mobius monopole antenna
  • the size of the CPW feed portion affects the input impedance and the radiation pattern, it may be appropriately selected for the quasi-Mobius type monopole antenna structure.
  • Table 2 is a design parameter table of the quasi-Mobius monopole antenna
  • Figure 6 is a view showing the input matching characteristics of the quasi-Mobius monopole antenna optimized.
  • Input matching characteristics in the operating band are affected by the size and line width of the quasi-Moebius loop, the feed match length and the ground plane size. Attempts to downsize the antenna through quasi-Moebius cross-connections resulted in degradation of the low-band antenna input impedance, resulting in a slight reduction in operating bandwidth.
  • the simulated 10 dB reference return loss characteristics operate in the 3.4 to 6.5 dB band.
  • FIG. 7 is a diagram showing the surface current distribution in a 90 ° feeding phase of the center frequency (4.5 GHz) of the quasi-Mobius monopole antenna.
  • 8A, 8B, and 9A to 9C show optimized simulated two-dimensional and three-dimensional radiation patterns at 3.5, 4.5, 5.5 GHz, and three frequencies in an operating band of the quasi-Mobius monopole antenna, respectively.
  • Table 3 is a table of electrical radiation patterns at each frequency.
  • the radiation pattern exhibits good vertical linear polarization characteristics, and the antenna gain within the operating band exhibits omni-directional radiation in the range of approximately 1.9 dBi, and an 8-way radiation characteristic with a 3-kHz beamwidth in the 73.2 to 84.4 ° range in both directions. It can be seen that.
  • FIG. 10 is a configuration diagram of a double circular ring monopole antenna (hereinafter, referred to as 'type 3') for miniaturizing a circular disk monopole antenna structure.
  • 'type 3' a double circular ring monopole antenna
  • Table 4 is an optimized design parameter table.
  • the dual circular ring structure arranged in the longitudinal direction can reduce the antenna size (or length) in the transverse direction, which is a fundamental reason for providing a good omnidirectional radiation pattern in the azimuth direction.
  • the simulated 10dB criterion shows that the input return loss characteristic operates in the band of 2.9 to 6.5GHz.
  • 11 is a graph of input matching characteristics of a circular disk monopole antenna structure and type 1 to type 3;
  • the impedance band is poor in the frequency band lower than 3.5 kHz. This generally indicates a phenomenon that the impedance operating bandwidth deteriorates due to the miniaturization of the antenna.
  • FIG. 11 is a diagram illustrating two-dimensional radiation pattern characteristics of each comparison monopole antenna structure.
  • the miniaturized structures are relatively better than the reference antenna in the center frequency (4.5 GHz) and azimuth direction, and the radiation patterns of the reference antenna structure and the type 1 and type 2 structures in the elevation angle and wave angle direction Similarly, it can be analyzed that the type 3 structure is relatively more directional because of the longer radiation length in the longitudinal direction.
  • Table 5 is a comparison table of electrical and physical characteristics of the reference antenna structure and the type 1 to type 3 structures.
  • Type 1 structure Type 2 structure
  • Type 3 structure Electrical characteristics Operating impedance bandwidth at @ 10 dB 2.4 to 8.0 GHz 3.6 to 7.3 GHz 3.4 to 6.5 GHz 2.9 to 6.5 GHz Partial Bandwidth Rate (%) 107.7% 67.9% 62.6% 76.6%
  • Directional antenna (*) 3.3 dBi 2.7 dBi 2.8 dBi 3.1 dBi
  • Antenna efficiency (*) 86.8% 92.1% 91.7% 92.9%
  • the miniaturization structure of type 1 to type 3 compared to the reference antenna structure has the disadvantage of reducing the operating bandwidth, but it can be analyzed that it shows similar antenna directivity, antenna efficiency and gain characteristics, and 3 dB beam width characteristics. Very effective antenna miniaturization of less than 42% was achieved.
  • Mobius strips have a 180 ° phase difference between the inner space and the outer space.
  • the internal space and the external space are not separated spaces, but have an open space connected thereto.
  • the Mobius strip when cut along the circumference, it is not divided into two strips, but rather a strip that is twice as long as the circumference before cutting.
  • the Mobius strip has no start in phase mathematics and has one side. And Mobius strips resemble cylinders, but are bounded surfaces rather than ordinary surfaces. In addition, the mobius strip is not a three-dimensional closed space but a two-dimensional open space.
  • Table 6 shows the result table according to the number of times the Mobius strip was cut along the circumference.
  • Equation 1 The cos (t) term in Equation 1 and the sin (t) term in Equation 2 generate a 180 ° phase difference.
  • the function M (t, s) indicates that the end of one side is fixed and rotates 180 ° to meet on the opposite side.
  • the total circumferential length l of the Mobius strip can be derived as in Equation 3.
  • FIG. 13 is a comparison of Mobius strips and strips cut along the circumference thereof.
  • the Mobius strip having a phase difference of 180 ° between the inner space and the outer space can be seen to produce the longest strip of the circumference.
  • the present invention can miniaturize RF passive elements such as antennas, oscillators, resonators, etc. by utilizing the characteristics of the Mobius strip.
  • 14a and 14b is a block diagram of a planar Mobius strip according to the prior art.
  • the three-dimensional connecting bridge at the two ring connecting portions is a cause of the coupling coupling effect at low frequencies and is not suitable for application to an RF circuit having a single resonant frequency characteristic.
  • the flat-type Mobius strip according to the prior art is not a perfect two-dimensional structure, but a three-dimensional structure, there is a limit to apply to an integrated circuit such as MMIC.
  • the flat plate-shaped Mobius strip according to the prior art has a three-dimensional structure of a bridge connecting the inner space and the outer space, and as the electromagnetic interference occurs, it is difficult to be applied to an RF passive device having a characteristic of a single resonance frequency. there was.
  • the present invention provides a monopole antenna to which a semi-mobius strip having a via hole structure is applied to minimize the line coupling effect due to the structural characteristics of the flat-type Mobius strip according to the prior art and maximize the miniaturization.
  • the present invention can be miniaturized by increasing the number of cuts along the circumference while maintaining the same physical length of the mobius strip.
  • the total circumference of the mobius strip cut along the circumference is twice the circumference of the general mobius strip.
  • the semi-mobius strips cut along the circumference can be applied in the design of RF circuits, thereby miniaturizing the entire circumference length while maintaining the resonance frequency.
  • FIG. 15 is a front view of the quasi-Mobius strip
  • FIG. 16 is a rear view of the monopole antenna shown in FIG.
  • the quasi-Mobius strip is a radiator, comprising a plurality of loops disposed in the center of the front of the dielectric substrate and formed in a structure in which the Mobius strip is cut along the circumference at least once, as shown in FIGS. 15 and 16, one end of each loop And a second bridge for connecting via holes formed in one end of the innermost loop and the outermost loop, respectively.
  • the present invention physically separates a bridge connecting the inner space and the outer space where the two rings intersect the front and the back in order to minimize the line coupling effect phenomenon, which is a problem of the flat type Mobius strip according to the prior art. Then connect to the via hole.
  • the present invention can minimize the line coupling effect at a low frequency and the electromagnetic interference phenomenon that can occur seriously as the RF circuit becomes smaller at the same resonance frequency.
  • the quasi-Mobius strip can be derived as shown in Equations 4 and 5 when N is an even number.
  • the quasi-Mobius strip can be derived as shown in Equations 6 and 7 when N is an odd number.
  • the quasi-Mobius strip when the phase difference between the inner space and the outer space is 180 °, the quasi-Mobius strip can be miniaturized while increasing the number of times N cut along the circumference of the quasi-Mobius strip.
  • connection bridges of the first ring and the fourth ring are designed in front of the substrate, and the first and second, the second and the third, the third and the fourth. Burn connection bridges can be designed in front of the substrate and connected via via holes.
  • the quasi-Moebius strip having the via hole structure has a problem that it does not resonate with the designed resonant frequency without the optimization of the position of the via hole and the quasi-Mobius strip structure.
  • the impedance matching process optimized for the resonant frequency designed by the parameter sweep according to the position and size of the via hole of the semi-Mobius strip of the via hole structure and the ring thickness and the bridge width of the semi-Mobius strip is described below.
  • Three elements of the parametric sweep for impedance matching of the quasi-Mobius strip are the thickness of the ring line and the bridge width of the quasi-Mobius strip and the position and radius change of the via hole.
  • the thickness of the ring line of the quasi-Moebius strip is changed, and the variation of the resonance frequency and the impedance matching are explained.
  • Table 7 is a result table of the resonant frequency and the return loss according to the thickness change of the ring line of the quasi-Mobius strip
  • FIG. 19 is a front view of the optimized quasi-Mobius strip
  • FIG. 20 shows the parameters of the quasi-Mobius strip. to be.
  • the thickness of the ring line closest to the designed resonant frequency of 2.4 kHz with the best return loss characteristics is 1 mm for the 1st and 3rd ring lines, and 0.6 mm for the 2nd ring line.
  • 21 is a block diagram of a monopole antenna to which a quasi-Mobius strip is applied.
  • the monopole antenna employing the quasi-Mobius strip can be optimized by varying three parameters: the ring line thickness and bridge width of the Mobius strip, and the position and radius of the via hole.
  • 22 is a graph showing simulation results of resonance frequency and return loss according to the thickness change of the ring line.
  • the ring line thickness of the optimized quasi-Mobius strip is 1 mm thick for ring lines 1 and 3, and 0.6 mm thick for ring lines 2.
  • the size and size of the via hole can be varied and optimized under the above conditions.
  • Table 8 is a result table of resonant frequency and return loss according to the change of the position and radius of the via hole of the quasi-Moebius strip.
  • the position and radius of the via hole having the closest resonance frequency of 2.4 GHz and the best return loss characteristics are 0.5 mm above the y axis, and the radius of the via hole is 0.5 mm The radius of the via hole is 1mm.
  • the resonance frequency and return loss can be optimized while varying the width of the bridge of the quasi-Mobius strip under the conditions of the ring line thickness of the optimized quasi-Mobius strip and the position and radius of the via hole.
  • Table 9 is a resonant frequency and return loss table according to the bridge width change
  • Figure 23 is a graph of the measurement results of the resonance frequency and return loss according to the bridge width change.
  • the bridge width that is closest to the designed resonant frequency of 2.4 Hz and has the best reflection loss characteristic is when it is widened by about 0.5 mm, as shown in FIG.
  • the ring lines 1 and 3 have a thickness of 1 mm
  • the ring lines 2 have a thickness of 0.6 mm
  • the radius of the via hole at the top of the y axis is 0.5 mm
  • the radius of the via hole at the bottom It can be seen that the return loss of the quasi-Moebius strip is optimized at 1mm.
  • FIG. 24 is a graph showing a result of comparing return loss between ring 1,2,3 antennas having different radii and quasi-Moebius strip antennas.
  • the resonant frequency of the quasi-Mobius strip antenna is formed in the 2.4 kHz band and the peak loss of the return loss S 11 is 17.3 dB.
  • the antenna using the quasi-Mobius strip optimized in this embodiment is composed of three ring lines with different radii, but has a single resonance frequency.
  • the antenna to which the quasi-Moebius strip is applied to a size of about 1/3.
  • FIG. 25 illustrates a via hole structure connecting two micro strip lines.
  • the via hole may consist of two pads and one cylinder.
  • FIG. 26 is an equivalent circuit diagram of the most accurate but complicated via hole.
  • the equivalent circuit shown in Fig. 26 is composed of three distribution constant elements in a transmission line without loss.
  • the three devices consist of an upper pad, a cylinder and a lower pad, with coupling capacitance between them.
  • the mutual capacitance C between each element is present between the conductor elements.
  • mutual inductance M also exists.
  • the mutual inductance M is generated by the magnetic flux generated by the change of the magnetic field with time between each coupled conductor element.
  • the equivalent circuit can be applied accurately regardless of the frequency, but the circuit is complicated and may take a lot of simulation time.
  • FIG. 27 is a distribution constant via equivalent circuit diagram that may be practically applied as compared to FIG. 26.
  • each element has a distinct influence on each other.
  • the diameter of the cylinder of the via becomes smaller, the upper pad and the lower pad become capacitive, and the cylinder becomes inductive.
  • the distributed constant via equivalent circuit shown in FIG. 27 does not include a coupling phenomenon between each device, and thus it is difficult to apply accurately at high frequencies.
  • the via hole equivalent circuit shown in FIG. 28 is the simplest of the three via equivalent circuits, and can be practically applied when the resonance frequency is less than 3.5 kHz.
  • the via hole equivalent circuit can ignore the energy radiated from the via hole because the via size is relatively small at a low frequency at a low frequency. Thus, via holes are interpreted as lumped elements.
  • the via hole is analyzed by the full wave method or by the ⁇ -type equivalent circuit composed of L (Inductor) and C (Capacitor).
  • the via hole is interpreted as a lumped element.
  • 29 is a via hole equivalent circuit applied in this embodiment.
  • One via hole is connected by one inductance L connected in series and two capacitance C connected in parallel. It can be inferred from the equivalent circuit of the via that the equivalent circuit of the entire quasi-Moebius strip varies depending on the radius and position of the via.
  • the quasi-Mobius strip is composed of an inductance L component and a capacitance C component that increase with the number of cutting.
  • the quasi-Moebius strip consists of n ring lines and bridges.
  • 30A to 30D are exemplary views of an inductor implemented on a substrate, and Table 10 is a coefficient current sheet of inductance L.
  • c 1 , c 2 , c 3 , and c 4 apply 1.00, 2.46, 0.00, and 0.20, respectively.
  • the impedance matching condition is when the relationship between the characteristic impedance (Z 0 ) and the load impedance (Z L ) is a complex conjugate.
  • the total inductance L and the total capacitance C of the quasi-Mobius strip antenna are derived as shown in equations (10) and (11).
  • the equivalent circuit of the quasi-Mobius strip antenna is described using total L and C satisfying the impedance matching condition of the quasi-Mobius strip antenna derived from Equations 10 and 11, and the simulation results are compared and analyzed to compare the differences.
  • Equations 10 and 11 The equivalent circuit of the quasi-Mobius strip antenna is described using total L and C satisfying the impedance matching condition of the quasi-Mobius strip antenna derived from Equations 10 and 11, and the simulation results are compared and analyzed to compare the differences.
  • 31 and 32 are schematic diagrams of first and second quasi-Mobius strips (hereinafter, referred to as 'first type and second type'), respectively.
  • Quasi-Moebius strips of the first type form respective ring lines 1,2,3, created by rotating ring lines with different radii from 0 ° to 325 ° and connecting them from the front and back sides of the substrate,
  • the open strip is connected without any beginning or end, which is characteristic of the strip. That is, the first type quasi-Mobius strip has a structure in which a ring line 1 having a radius of 3 mm is connected to a ring line 2 having a radius of 3.75, and a ring line 3 having a radius of 4.5 mm is connected to the ring line 2 at the front surface of the substrate.
  • ring line 1 is connected to ring line 3 via bridge 1.
  • Quasi-Moebius strips of the second type form respective ring lines 1,2,3, created by rotating ring lines with different radii from 0 ° to 325 °, with the front side of the substrate connecting rings 1 and 3.
  • the back side of the substrate was connected to the ring line 1 and the ring line 2 and the ring line 2 and the ring line 3 to implement an open strip that is connected to the start and end characteristics of the Mobius strip.
  • 33 and 34 are schematic diagrams of the first and second quasi-Mobius strip antennas, respectively.
  • the bridge of the antenna to which the first type quasi-Mobius strip is applied is one, and the number of via holes is two.
  • the number of bridges of the antenna to which the second type quasi-Mobius strip is applied is two, and the number of via holes is four.
  • the present invention is described as applying a quasi-Moebius strip antenna of the first type.
  • 35 and 36 are front and rear views of an antenna to which a quasi-Mobius strip is applied according to a preferred embodiment of the present invention.
  • the distance of the ring 1, ring 2, ring 3 and ring 4 may be set to 1.68mm, which is 0.5 times the radius r, 3.36mm.
  • bridge 1 may be provided on the back of the substrate, and bridge 2, bridge 3, and bridge 4 may be provided on the front of the substrate.
  • the present invention is not necessarily limited thereto, and the bridge 1 may be provided on the front surface of the substrate, and the bridges 2 to 4 may be changed on the rear surface of the substrate.
  • an impedance matching process optimized for the resonant frequency designed through the parameter sweep according to the position and size of the via hole of the quasi-Mobius strip, the thickness of the ring line of the quasi-Mobius strip, and the size of the bridge should be performed.
  • the resonance frequency designed in this embodiment is 2.4 kHz
  • the actual resonant frequency resonates at 2.78 kHz in the measurement result of FIG. 38, so the radiation patterns were measured at 2.4 kHz, 2.5 kHz and 2.78 kHz.
  • the gain of the radiation pattern is as shown in Table 11, and it can be seen that the peak gain and average gain increase at the resonance frequency of 2.78 kHz.
  • the reflection coefficient and the gain can be optimized by varying the thickness of the ring line, the position and radius of the via hole, and the width of the bridge.
  • 41 to 43 are diagrams illustrating a state in which the radiators of the UWB antenna shown in FIG. 35 are rotated by 90 °, 180 °, and 330 °, respectively.
  • FIG. 48 is a graph comparing the reflection loss according to the rotation angle of each rotated radiator.
  • the present invention can control the cue factor and bandwidth by rotating the radiator of the quasi-Mobius strip antenna.
  • the present invention can control the directivity by adjusting the rotation angle of each ring provided on the radiator of the quasi-Mobius strip.
  • FIGS. 49 to 51 are diagrams illustrating a state in which each ring provided in the radiator of the quasi-Mobius strip is rotated by 300 °, 330 °, and 350 °, respectively.
  • 58 to 60 are three-dimensional radiation pattern graphs to which the radiators illustrated in FIGS. 49 to 51 are applied.
  • the present invention can control the directivity of the radiation pattern of the antenna by adjusting the rotation angle of each ring provided in the radiator of the quasi-Mobius strip from 0 ° to 360 °.
  • the radiation pattern of the circular disk monopole antenna is omni-directional in all directions.
  • the reason why the radiation pattern of the circular disk monopole antenna is omnidirectional is formed by electric field and magnetic field distribution due to current feeding.
  • the quasi-Moebius strip antenna according to the present invention is omnidirectional as the rotational angle of the ring provided in the radiator is closer to 360 °, and the directivity is increased as the rotational angle of the ring is smaller than 360 °.
  • MIMO antenna is the core technology of 4G and 5G, and is applied to overcome the limitation of channel capacity due to multipath fading, and improves the limit of channel capacity due to multipath by applying multiple transmit / receive antennas to the ends of the transmit and receive terminals. Let's do it.
  • Performance parameters of such MIMO antennas are largely divided into Diversity performance and MIMO performance.
  • Diversity performance parameters include balanced branch Power Mean Gain (MEG), correlation, diversity gain, and MIMO performance parameters include MIMO capacity and multiplexing efficiency.
  • MEG balanced branch Power Mean Gain
  • correlation correlation
  • diversity gain correlation
  • MIMO performance parameters include MIMO capacity and multiplexing efficiency.
  • the channel capacity in the MIMO system increases in proportion to the number of transmitting and receiving antennas, but there are correlations between the antennas because a plurality of antennas are applied.
  • the correlation between the antennas reduces the channel turbulence of the entire MIMO system.
  • the design of MIMO antennas employs a diversity technique that improves spectral efficiency, with the mutual coupling between antennas degrading the performance of these systems.
  • Correlation between MIMO antennas is one of the most important parameters related to communication system spectral efficiency, and the higher the correlation between antennas, the lower the spectral efficiency of the communication system.
  • Equation 12 The calculation of the correlation envelope coefficient (ECC) between the antennas may be defined by Equation 12 or 13 considering the three-dimensional radiation pattern.
  • F i ( ⁇ , ⁇ ) means a radiation pattern when only I is excited when all other ports are terminated to 50 ⁇ , and means Hermitian operation.
  • Equation 12 may be expressed in a simplified form by using an S-parameter between array antennas. If the environment of the multipath is uniform, Equation 12 may be approximated by Equation 14.
  • Equation 14 derives ECC by applying an S parameter between two antennas, which is simpler than calculating using Equation 13.
  • next generation wireless communication requires a large capacity MIMO antenna technology for high speed data transmission.
  • a large capacity MIMO antenna with high isolation characteristics a number of high isolation antennas must be implemented in a limited radiation space.
  • the signals output to the counter terminal through the long path and the short path satisfy phase conjugation conditions, that is, 180 ° phase difference conditions, at the same amplitude, thereby improving isolation between terminals.
  • the structure of the slot affects the input impedance and the terminal-to-terminal isolation characteristics of the antenna.
  • microstrip feeding was used instead of CPW feeding.
  • each input terminal is implemented at a 1 mm offset position from the slot.
  • the input return loss characteristics and the terminal isolation characteristics are assumed to be 10 dB at the same time, it can operate in the 3.33 to 5.67 GHz band, and especially in the 3.2 to 5.0 GHz band, very good input matching and isolation characteristics with an input reflection coefficient of 0.18 or less are obtained. Can be seen.
  • the two-terminal small MIMO antenna described in this embodiment has been confirmed that the correlation between the antenna is very low in the range of the frequency 3 to 8GHz, it can also be used in the massive MIMO antenna for UWB.
  • the miniaturized UWB antenna according to the present embodiment is applied to a MIMO antenna, it is confirmed that the ECC value representing the correlation between the antennas is 0.02 or less in the frequency range of 3 kHz to 8 kHz, thereby improving spectral efficiency. You can.
  • the UWB monopole antenna has been described, but the present invention is not limited thereto, and the UWB monopole antenna may be applied to not only the UWB monopole antenna, but also to the monopole antenna of various methods, and the unit resonance cell and the wireless power of the meta material. It can be modified to be applied to RF passive elements such as transmission resonators and oscillators.
  • the invention applies to monopole antenna technology employing a quasi-Moebius strip structure.

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  • Variable-Direction Aerials And Aerial Arrays (AREA)
  • Details Of Aerials (AREA)

Abstract

La présente invention concerne une antenne unipolaire qui comprend : un élément rayonnant qui est disposé au centre d'une surface avant d'un substrat diélectrique, et qui comprend une pluralité de boucles formées dans une structure dans laquelle un ruban de Möbius est coupé au moins une fois le long de la circonférence; un premier pont pour connecter séquentiellement une extrémité de chaque boucle; et un second pont pour connecter des trous d'interconnexion formés respectivement à une extrémité de la boucle la plus intérieure et de la boucle la plus extérieure, ainsi obtenant un effet pour permettre à une antenne, sur laquelle un quasi-ruban de Möbius et une structure de trou d'interconnexion sont appliqués, d'être miniaturisée.
PCT/KR2017/001564 2015-02-11 2017-02-13 Antenne unipolaire Ceased WO2017138800A1 (fr)

Priority Applications (3)

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EP17750492.5A EP3416241A4 (fr) 2016-02-11 2017-02-13 Antenne unipolaire
JP2018543379A JP6583901B2 (ja) 2016-02-11 2017-02-13 モノポールアンテナ
US16/076,972 US10283856B2 (en) 2016-02-11 2017-02-13 Monopole antenna

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KR20150021195 2015-02-11
KR1020160015897A KR101729036B1 (ko) 2015-02-11 2016-02-11 모노폴 안테나
KR10-2016-0015897 2016-02-11

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CN109755729A (zh) * 2018-12-11 2019-05-14 上海电力学院 一种柔性的双阻带超宽带mimo天线
US12142828B2 (en) 2021-11-10 2024-11-12 Aisin Corporation Antenna device

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KR102489559B1 (ko) * 2018-02-05 2023-01-18 김미정 그래핀 및 탄소복합소재를 응용한 안테나 및 무선전력 전송 공진기 소형화 방법
KR102641308B1 (ko) * 2022-04-29 2024-02-27 주식회사 에이스테크놀로지 이중 편파 안테나 방사체

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US12142828B2 (en) 2021-11-10 2024-11-12 Aisin Corporation Antenna device

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