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WO2017013467A1 - Systèmes de transmission ofdm avec un rapport de puissance crête-à-moyenne réduit - Google Patents

Systèmes de transmission ofdm avec un rapport de puissance crête-à-moyenne réduit Download PDF

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Publication number
WO2017013467A1
WO2017013467A1 PCT/IB2015/055494 IB2015055494W WO2017013467A1 WO 2017013467 A1 WO2017013467 A1 WO 2017013467A1 IB 2015055494 W IB2015055494 W IB 2015055494W WO 2017013467 A1 WO2017013467 A1 WO 2017013467A1
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symbols
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Inventor
Denis Mestdagh
Joel Cambonie
Loïc LIETAR
Jorge GULFO-MONSALVE
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Greenwaves Technologies
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Greenwaves Technologies
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2614—Peak power aspects
    • G—PHYSICS
    • G06—COMPUTING OR CALCULATING; COUNTING
    • G06F—ELECTRIC DIGITAL DATA PROCESSING
    • G06F17/00—Digital computing or data processing equipment or methods, specially adapted for specific functions
    • G06F17/10—Complex mathematical operations
    • G06F17/14—Fourier, Walsh or analogous domain transformations, e.g. Laplace, Hilbert, Karhunen-Loeve, transforms
    • G06F17/141—Discrete Fourier transforms
    • G06F17/142—Fast Fourier transforms, e.g. using a Cooley-Tukey type algorithm
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2626—Arrangements specific to the transmitter only
    • H04L27/2627—Modulators
    • H04L27/2628—Inverse Fourier transform modulators, e.g. inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators
    • H04L27/2633—Inverse Fourier transform modulators, e.g. inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators using partial FFTs

Definitions

  • the present invention relates to transmission systems, and more particularly to the reduction of the peak-to-average power ratio (PAPR) of the transmitted signal in an orthogonal frequency-division multiplexing (OFDM) system.
  • PAPR peak-to-average power ratio
  • the transmitted signal is formed of a series of symbols, wherein each symbol is the sum of N subcarriers having different amplitude and phase values.
  • N is usually a power of 2, ranging from several tens to several thousands.
  • the amplitude and phase values depend on the conveyed data and may be considered as random.
  • the transmitter is usually designed to handle the average level of the transmission signal and peaks that fall below an accepted threshold. Peaks above the threshold are clipped by the digital- to-analog converter or the linear power amplifier of the transmitter, causing distortion and a temporary increase of the bit error rate.
  • Increasing the accepted threshold may reduce the bit error rate, but increases the power consumption, since the power amplifier of the transmitter is then designed to handle higher amplitudes.
  • the transmission signal may be attenuated, but this reduces the signal-to-noise ratio (SNR) and the communication distance.
  • PCT patent application WO 2015/028684 discloses a technique for reducing the PAPR statistically, leading to a lower power consumption while maintaining the communication distance, or leading to a higher communication distance with a same power consumption.
  • the symbol instead of transmitting a symbol that conveys N subcarriers, the symbol is split, using temporal diversity properties of Fourier transforms, into M consecutive sub-symbols that each conveys only N/M subcarriers.
  • N and M are each a power of two.
  • the PAPR is statistically divided by M.
  • the data to be transmitted is first mapped to N quadrature amplitude modulation (QAM) constellations.
  • QAM quadrature amplitude modulation
  • BPSK 2-QAM
  • 4-QAM 4-QAM
  • 8-QAM 16-QAM
  • Each constellation is represented by a complex number X(i), where i (varying from 0 to N-l) is the input or frequency index of an N-point Inverse Fast Fourier Transform (N-IFFT).
  • N-IFFT Inverse Fast Fourier Transform
  • the N-point IFFT produces N time-domain samples x(0) to x(N-l), each also being a complex number.
  • the N samples x form a symbol that is transmitted over a symbol period Ts.
  • the subcarriers are equally spaced in frequency and the frequency pitch is equal to the inverse of the symbol period.
  • the real and imaginary parts of the symbol are converted into two separate analog signals that are quadrature modulated on a radiofrequency carrier.
  • the samples x of the symbol produced by the IFFT are processed, such as by a digital signal processor DSP, to form M sub-symbols including each N/M samples.
  • Each sub-symbol is built such that it conveys a different set of N/M equally spaced subcarriers.
  • M 2
  • two sub-symbols go and gi are built, corresponding respectively to the even-ranked subcarriers and the odd-ranked subcarriers.
  • a sub-symbol g k conveys the subcarriers of ranks k+Mi, where i varies between 0 and—— 1.
  • a system such a disclosed in the above PCT application requires additional signal processing resources after the IFFT on the transmitter side and reciprocal processing resources before an FFT on the receiver side.
  • the present disclosure generally relates to a data transmission method, comprising the steps of providing N values to transmit, where N is a power of 2; dividing the values in M vectors of N/M values, where M is a power of 2; applying each vector in parallel to two groups of M N/M-point inverse Fourier transforms, producing a first and a second group of M sub-symbols; modifying the input of each inverse Fourier transform by a different pseudo-random number sequence; producing combinations of sums in pairs of sub-symbols from the first group with sub-symbols from the second group; and selecting for transmission the sum having the lowest peak-to-average power ratio (PAPR).
  • PAPR peak-to-average power ratio
  • the method may comprise the further steps of frequency-shifting each z ' -th sub-symbol of each group by a step -i/N, where i varies between 0 and M-l, producing in each group M sub-symbols conveying a different set of subcarriers; frequency-shifting all sub-symbols to a common set of subcarriers; and producing the combinations of sums from the frequency-shifted sub-symbols.
  • the method may comprise the further steps of transmitting the selected sum as a sub- symbol; receiving the transmitted sub-symbol; demodulating the received sub-symbol; and dividing the demodulated sub-symbol by the sum of the pseudo-random sequences used in forming the transmitted sum.
  • the method may alternatively comprise the further steps of transmitting the selected sum as a sub-symbol with zero-padded intervals; receiving the transmitted sub-symbol; inserting a cyclic prefix at the beginning of the received sub-symbol; applying to the cyclic prefix a multiplication factor based on the common subcarrier set; demodulating the received sub-symbol; and dividing the demodulated sub-symbol by the sum of the pseudo-random sequences used in forming the transmitted sum.
  • the method may comprise the further steps of changing the common subcarrier set after each transmitted sum according to a pattern; and sharing the pattern with the receiver.
  • the method may comprise the further steps of changing the pseudo-random sequences after each transmitted sum according to a pattern; and sharing the pattern with the receiver.
  • the pseudo-random sequences applied in the first group of sub-symbols may be real numbers, and the pseudo-random sequences applied in the second group of sub-symbols may be imaginary numbers.
  • FIG. 1 previously described, is a block diagram illustrating the generation of a sub-symbol pair in a reduced peak-to-average power ratio OFDM transmitter;
  • FIG. 2 is a block diagram illustrating an exemplary generation of a sub-symbol pair with an embodiment of a low-complexity 8-point IFFT
  • ⁇ Figure 3 is a block diagram illustrating an exemplary generation of a sub-symbol quadruplet with an embodiment of a low-complexity 8-point IFFT;
  • FIG. 4 is a block diagram of a generic embodiment of a low-complexity N- point IFFT used for generating and transmitting a sub-symbol M-uplet;
  • FIG 5A is a block diagram of a first embodiment of receiver circuitry for processing sub-symbol M-uplets;
  • FIG. 5B illustrates generation of a cyclic prefix on the receiver side in the context of a zero-padding transmission
  • FIG. 6 is a block diagram of another embodiment of receiver circuitry for processing sub-symbol M-uplets
  • FIG. 7 is a block diagram of an embodiment of transmission circuitry using selective mapping
  • ⁇ Figure 8 is a block diagram of an embodiment of transmission circuitry using high-efficiency selective mapping
  • ⁇ Figure 9 is a block diagram of another embodiment of transmission circuitry using high-efficiency selective mapping.
  • a reduced PAPR OFDM system as disclosed in PCT patent application WO 2015/028684 adds complexity to a conventional OFDM system due to constraints such as the reusability of existing FFT and IFFT circuitry. In the present disclosure, it is sought to reduce the complexity of the system while maintaining or improving its performance.
  • the inventors have observed that the sub-symbols built from the output of the inverse FFT in the above PCT application are derivable from intermediate radix stages of the inverse FFT.
  • Figure 2 illustrates an 8-point inverse FFT circuit in more detail, according to a decimation-in-time (DIT) representation.
  • An inverse FFT circuit, or IFFT may be represented as a succession of radix stages.
  • the simplest radix stage is a second order radix stage, denoted radix-2.
  • a radix-2 stage combines the outputs of two half-size IFFTs into one full-size output.
  • x is the N-sample time-domain output
  • si are the sub-symbols respectively output by the two half-size IFFTs
  • i is the sub-symbol sample index (varying between 0 and j— 1)
  • W3 ⁇ 4 is a so-called twiddle factor.
  • the last radix-2 stage combines the outputs of two 4-point IFFTs into the 8-sample output symbol x(0), x(l) ... x(7).
  • the first 4-point IFFT operates on the even-ranked sub-carriers, whereas the second 4-point IFFT operates on the odd-ranked sub-carriers (the sub-carrier ranks are defined here as the indexes of the input values X).
  • each of the 4-point IFFTs may be represented as a radix-2 stage combining the outputs of two 2-point IFFTs, as shown.
  • a set of M sub-symbols sM 0 , sMi... sM M -i may be produced by M intermediate N/M-point IFFTs, where M and N are powers of 2.
  • Each sub-symbol sM k is defined such that it conveys the subcarriers of ranks k+Mi, where & is a constant between 0 and M-l, and i varies from 0 to—— 1.
  • the subcarriers of ranks k+Mi will be denoted hereinafter as the subcarrier set k.
  • sub-symbols can be transmitted individually without modification over sub-symbol periods Ts/M to convey the same information as a full symbol over the symbol period Ts.
  • Such an approach is equivalent to using in parallel M conventional N/M- sub carrier OFDM transmitters, with no apparent added value.
  • the remaining circuitry of the radix-2 stage may be skipped, as shown in gray. In other words, only a partial N-point IFFT is achieved up to the last radix stage.
  • the sub-symbols g2 0 and g2i are tapped from the last radix stage, and a cyclic-prefix CP is pre-pended at 20 to each sub-symbol.
  • the cyclic prefix is not formed as in a conventional OFDM system due to the temporal diversity properties of the sub-symbols, here sub-symbol g2i only.
  • the cyclic prefixes for the sub-symbols g2 0 and g2i are the tail ends of the sub-symbols multiplied respectively by 1 and -1.
  • Each sub-symbol with its cyclic prefix is converted to an analog signal at 22, in fact two signals I and Q corresponding to the real and imaginary parts of the sub-symbol, which are then quadrature modulated and transmitted at 24.
  • Each sub-symbol is thus transmitted over a time Ts/M, where Ts is the full symbol period, equal to the inverse of the frequency pitch of the N sub carriers.
  • the sub-symbols may be zero-padded up to the same length as the cyclic prefix, as conventionally done in a so called ZP-OFDM system. Then, on the receiver side, after finding the sub-symbol boundaries, the zero-padded intervals are filled with the corresponding cyclic prefixes before FFT processing.
  • This technique has the feature of cancelling the transmission power in the zero padded intervals.
  • the cyclic prefixes are subject to the multiplication factors that would have been applied on the transmitter side.
  • Figure 3 illustrates an 8-point IFFT circuit with a last radix-4 stage, according to a DIT representation.
  • a radix-4 stage combines groups of four inputs to produce respective groups of four outputs, according to the matrix equation:
  • i is the sub-symbol sample index varying between 0 and 1
  • s4 0 to s4 3 are
  • the radix-4 stage has not been fully shown in figure 3. Only the links for producing the time-domain samples x(2) and x(5) have been shown.
  • the sub- symbols extracted for transmission g4 0 to g4 3 are the outputs of the four N/4-point IFFTs after application of the twiddle factors W ⁇ kl in the radix-4 stage.
  • the remaining circuitry of the radix-4 stage may be skipped, as shown in gray. In other words, only a partial N-point IFFT is achieved up to the last radix-4 stage.
  • the cyclic prefixes for the sub-symbols g4 0 to g4 3 are the tail ends of the sub-symbols multiplied respectively by 1, -j, -1, and j, as illustrated.
  • M sub-symbols gM 0 to gM M -i may be extracted for transmission from the input of the radix-M stage, after application of the twiddle factors in the radix stage.
  • a sub-symbol gM k having N/M values the applicable twiddle where & is a constant between 0 and M-l and i varies between 0 and
  • twiddle factors apply to the sub-symbol a frequency shift of step -— to produce the sought sub-symbol gM k .
  • the multiplication factors applied in the cyclic prefixes are M-th roots of unity taken in the clockwise direction.
  • Figure 4 is a block diagram of a generic embodiment of transmitter circuitry applying the above teachings.
  • An N/M-value vector X k is applied to an N/M-point IFFT circuit 40 that produces a corresponding symbol sM k .
  • the vector X k may be formed of the series of frequency-domain values X(k+Mi) input to the N-point IFFT, where k is the rank of the vector in a group of M vectors to transmit, and i varies from 0 to—— 1.
  • the structure of figure 4 however does not require that the values of X k be mapped to inputs of an N-point IFFT.
  • the symbol sM k is applied to a programmable frequency-shifter 42, taking the rank k of
  • the incoming signal is quadrature demodulated at 50, and the resulting in-phase and quadrature signals (I, Q) are separately converted to digital at 52.
  • the resulting digital complex samples form the current sub-symbol, with zero-padded intervals preceding and following the sub-symbol.
  • the start-of-symbol is located at 54.
  • a cyclic prefix is inserted at 56, applying the corresponding fourth root of unity, as explained previously.
  • An FFT window is applied at 58 to N/4 samples, usually starting at the start-of-symbol detected in 54.
  • the window thus isolates the current sub-symbol g4 k .
  • g4 k is frequency-shifted by a step— at 60, producing a conventional time-domain sub- symbol s4 k .
  • the sub-symbol s4 k is processed by an N/4-point forward FFT 62 producing frequency-domain values X.
  • the FFT output is usually processed by an N-frequency equalizer 64.
  • the FFT 62 outputs only N/4 values that are mapped to suitable frequency inputs of the equalizer.
  • the mapping may be performed by a demultiplexer 66 that receives the output of the FFT and is controlled by value k.
  • Value k causes the demultiplexer to send the N/4 values to the group of frequencies used by the transmitted sub-symbol g4 k , i.e. the frequencies k+4i.
  • the consecutive values of the FFT output are mapped to inputs 1, 5... l+4i... N-3.
  • Figure 5B illustrates a process of generating the cyclic prefix on the receiver side with a zero-padding transmission, such as performed in boxes 54, 56 and 58 of figure 5A.
  • An OFDM system is usually designed to account for multi-path signal transmissions.
  • an OFDM receiver will receive a main, "Line-Of- Sight" (LOS) symbol, and multiple reflections with various delays and attenuations with respect to the LOS symbol, called “Non-Line-Of- Sight" (NLOS) symbols.
  • Figure 5B depicts the LOS symbol and the NLOS symbol having the largest allowed delay, i.e. the duration of a zero-padded interval.
  • Circuit 54 is designed to find the starting point of the LOS symbol, which also defines the starting point of the FFT window in 58.
  • the FFT window will usually correspond to the length of the LOS symbol, as shown.
  • the shown NLOS symbol is such that it starts in the FFT window after its preceding zero-padded interval ZP - its end portion thus exceeds the end of the FFT window by the duration of the zero-padded interval.
  • the symbols LOS and NLOS, and any other symbols with different delays are received added together to form the signal seen in the FFT window.
  • the cyclic prefix is designed to restore the periodic nature of the signal seen in the FFT window.
  • the cyclic prefix is generated in 56 by adding to the signal at the beginning of the FFT window, the portion of the signal that exceeds the FFT window by the duration of the zero-padded interval.
  • the tail ends of the NLOS symbols are added in the zero-padded interval that precedes the NLOS symbols.
  • the cyclic prefix is obtained differently, as already mentioned.
  • the conventional cyclic prefix is multiplied by -j .
  • the applicable factor is 1, -j, -1 and j respectively for the sub-symbols g4 0 to g4 3 .
  • the factor applicable for a sub- symbol gM k is the k-th M-th root of unity in the clockwise direction.
  • Each sub-symbol g4 k from the FFT window 58 is fed individually into a full N-point FFT 68.
  • Each of the sub-symbols g4 k conveys a different set of subcarriers, i.e. subcarrier set k.
  • the FFT circuit may reflect this by producing significant values only on the N/M outputs corresponding to subcarrier set k. For instance, as shown, feeding sub-symbol g4i to the FFT circuit produces N/4 significant values on outputs 1, 5... l+4i... N-3, the remaining outputs being non-significant in theory, as illustrated by zeros in the figure.
  • each sub-symbol g4 k does not include an integer number of cycles of its subcarriers, because it was frequency-shifted by a step smaller than the pitch of these subcarriers.
  • the zero- padded N-point FFT can thus produce the sought values without applying the frequency-shift corrections (60 in figure 5A) to the sub-symbols g4 k .
  • Each of the M zero-padded sub-symbols thus produces significant values on a different set of outputs of the FFT.
  • the FFT will have produced the sought significant values X(0) to X(N-l) over all N outputs.
  • the significant values may be extracted individually from the corresponding outputs of the FFT as they are produced, and reordered in a memory, if required, for further processing. The extraction is preferably carried out after frequency equalization 64.
  • the resulting values may then be processed further like in a conventional OFDM receiver system. Since the N-point FFT circuitry is designed to always process zero-padded sub- symbols, its structure may be significantly simplified with respect to a full N-point FFT. In particular all the calculation nodes that perform combinations of zero values can be omitted.
  • Each sub-symbol that is thus individually processed by the FFT conveys data on subcarriers that have a frequency pitch M times larger than the pitch between the N original subcarriers. This decreases the sensitivity to Doppler effect by a factor M with respect to a conventional N-subcarrier OFDM system.
  • each sub-symbol conveys only N/M subcarriers
  • the system still uses all N subcarriers and offers the robustness of an N-subcarrier OFDM system. For instance, the larger number of subcarriers offers better redundancy, i.e. the possibility to move data between subcarriers when the transmission channel is degraded for specific subcarriers.
  • Figure 7 is a block diagram of an embodiment of transmission circuitry that may further reduce the peak-to-average power ratio PAPR.
  • the system uses a partial N-IFFT like the one of figure 3 for carrying out a PAPR reduction technique called "selective mapping" or SLM in a cost effective way. Selective mapping was first disclosed in ["A Method to Reduce the Probability of Clipping in DMT -Based Transceivers", Denis J. G. Mestdagh and Paul M. P. Spruyt, IEEE Transactions On Communications, Vol. 44. No 10, October 1996].
  • Selective mapping as disclosed in the above article consists in multiplying a same set of N IFFT input values with successive different pseudo-random sequences, performing the IFFT for each pseudo-random sequence, thus producing a different randomized symbol corresponding to the same initial values, and transmitting the randomized symbol that exhibits the lowest PAPR.
  • an inverse pseudo-random sequence is multiplied with the FFT output values to reestablish the original values. This involves knowledge at the receiver side of the pseudo-random sequence used on the transmitter side.
  • identification may be transmitted by modulating an OFDM pilot tone that usually does not carry information.
  • the input frequency-domain vector X k is applied in parallel to M N/M-point IFFTs 40.
  • Each IFFT is followed by a static frequency shifter 70 that applies a different frequency shift among the ones applied by the programmable shifter 42 of figure 4, i.e. the frequency shifters 70 respectively apply shifts by steps 0, -1/N, -2/N... -(M-l)/N.
  • the first frequency shift by 0 is simply a direct connection.
  • this structure is a "partial N-point IFFT" using a last radix- M stage, as defined in relation to figures 2 and 3.
  • the sub-symbols provided by the frequency-shifters 70 for a same input vector X k are denoted g k ,o to gk,M-i -
  • a first approach for reducing the PAPR could be to select and transmit the sub-symbol g k,q that has the lowest PAPR among the M sub-symbols provided by the frequency shifters, as indicated in 72. This approach is however insufficient, because simulations show that the differences in PAPR between such sub-symbols are insignificant.
  • a preferred approach is to multiply the input vector X k by a different pseudo-random number sequence PRNo to PRNM-I before each IFFT 40.
  • the pseudo-random numbers may be complex and have and absolute value of 1 to preserve the amplitude of the values of vector X k .
  • the first sequence PRNo may be unity, i.e. leave the values unmodified.
  • the differences in PAPR between the thus randomized sub-symbols may be significant, and selecting the randomized sub-symbol g k,q having the lowest PAPR for transmission will yield a significant global PAPR reduction.
  • the selected randomized sub-symbol g k,q follows the same transmission path as in the system of figure 4.
  • the multiplication factor for producing the cyclic prefix CP for this sub-symbol depends on the position q that conditions the subcarrier set used in the transmission.
  • the sub-symbol g k,q selected for transmission does not use the subcarrier set k corresponding to its order of transmission - it uses the subcarrier set q, where q is arbitrary between 0 and M-l . Processing such sub-symbols on the receiver side generally implies knowledge of q, so that the sub-symbol can be correctly demodulated and the correct inverse pseudo-random sequence PRN q _1 be applied after demodulation.
  • the value q may be transmitted, for instance, by modulating an OFDM pilot tone, i.e. a subcarrier that conveys no data and that is used for synchronization purposes and estimating the transmission channel characteristics.
  • an OFDM pilot tone i.e. a subcarrier that conveys no data and that is used for synchronization purposes and estimating the transmission channel characteristics.
  • N-point FFT with zero padding as depicted in figure 6 for processing randomized sub-symbols on the receiver side may not require knowledge of q, provided the cyclic prefixes are added on the transmitter side.
  • a received sub-symbol g k,q is then zero-padded to the right and fed to the full N-point FFT 68.
  • the FFT will produce significant values at the output positions that correspond to the actual subcarriers contained in the sub-symbol, i.e. subcarrier set q and positions q+Mi, for i varying between 0 and—— 1.
  • the value q being unknown, there is an uncertainty as to which set of outputs of the FFT produces the sought values.
  • the value q may however be evaluated by analyzing the power spectrum of the FFT result and locating the outputs providing peak values, normally the outputs corresponding to the subcarriers used in the current sub-symbol. Such analysis may provide better results when performed after the frequency equalization.
  • the transmitter may be designed to systematically frequency-shift the selected sub-symbol g k,q to a fixed subcarrier set r, known to both the transmitter and the receiver, as shown by a frequency shifter 74 applying a frequency-shift step of (r-q)/M.
  • the receiver will systematically insert the cyclic prefixes applying the multiplication factor corresponding to subcarrier set r.
  • the knowledge of q is still required to select the pseudo-random sequence and equalization channels to apply to the demodulated sub- symbol.
  • the value r used in frequency shifter 74 may be modified according to a pattern known to both the transmitter and the receiver, for instance according to a shared secret key, to provide a level of encryption.
  • a receiver that has no knowledge of the pattern will be unable to apply the correct multiplication factors for inserting the cyclic prefixes, and will fail to demodulate the sub-symbols.
  • the pseudo-random sequences PRN may similarly be changed according to a pattern known to both the transmitter and the receiver, thus producing a second level of encryption.
  • Figure 8 is a block diagram of an embodiment of transmission circuitry that may even further reduce the peak-to-average power ratio PAPR. It is based on two partial N-point IFFT circuits 80a and 80b as depicted in figure 7, both receiving the current N/M-value frequency-domain vector X k .
  • the M inputs to the first partial IFFT circuit 80a are modified by pseudo-random sequences PRNA 0 to PRNA M -i, whereas the M inputs to the second partial IFFT circuit 80b are modified by pseudo-random sequences PRNB 0 to PR B M -1-
  • the numbers in the sequences PRNA may be real, for example randomly selected among 1 and -1, whereas the numbers in the sequences PRNB may be imaginary, for example randomly selected among j and -j .
  • the partial IFFTs 80a and 80b thus produce M randomized sub-symbols ak , o to ak,M-i and M randomized sub-symbols bk , o to bk,M-i - All randomized sub-symbols are frequency-shifted at 82 to a same subcarrier set r, i.e. a sub-symbol a k ,i or b k ,i is frequency-shifted by a step (r-i)/M.
  • Value r is arbitrary between 0 and M-l and may be a fixed value or changed according to a pattern known to both the transmitter and the receiver to provide a level of encryption when zero-padding transmission is used, as explained previously in relation to figure 7.
  • the sums may be divided by a normalization factor ⁇ /2 to preserve the average amplitude.
  • a purpose of frequency-shifting all sub-symbols to the same subcarrier set is to ensure that each pair of sub-symbols to add conveys the same subcarriers.
  • the PAPR of each of the M 2 sums o U;V is evaluated in 72 and the sum ⁇ ⁇ having the lowest PAPR is selected for transmission.
  • Such a sum thus uses the subcarrier set r, and has been randomized with sequences PRNA P and PRNB q .
  • value r is known and defines the multiplication factor of the cyclic prefix (when zero-padding transmission is used) and the FFT positions conveying the sought data (when a zero-padded N-point FFT is used, as in figure 6).
  • Values p and q are arbitrary and may be transmitted to the receiver in various know manners, such as by modulating one or several pilot tones. Values p and q may then be used in the receiver to identify the two pseudo-random sequences to use for recovering the original vector X k from the FFT output.
  • the original values of the vector X k may be obtained by the following equation: X r + ⁇ ) ⁇
  • X designates the N-value output vector of the N-point FFT
  • the indexes in parenthesis identify the values of the vectors, for i varying between 0 and N/M-1
  • a is the inverse of the normalization factor used in the transmitter for producing the sums, here equal to 2.
  • the receiver may also be configured as in figure 5A, using an N/M-point FFT producing an N/M-value vector X. Then the term X(r+Mi) in the above equation is replaced by X(i).
  • the original vector X k is recovered by dividing the demodulated sub-symbol by the sum of the pseudo-random sequences used in randomizing the transmitted sub-symbol, considering the division as a term-by-term division between vectors.
  • Figure 9 is a block diagram of a simplified version of the transmitter of figure 8 that may be used in certain applications.
  • the structure uses two groups of M N/M-point IFFTs 40.
  • the sub-symbols a k and b k produced by the IFFTs all use the same N/M subcarriers, whereby no frequency-shifting is required before producing the sums in 84.
  • the sum selected for transmission ⁇ ⁇ conveys the same N/M subcarriers.
  • the receiver side uses an N/M-point FFT and an N/M-frequency equalizer, requiring no frequency shifting and output remapping.
  • the cyclic prefix uses no variable multiplication factor.
  • the system is less robust to transmission channel deterioration, since fewer subcarriers are available for redistributing data transmitted on heavily attenuated subcarriers.

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Abstract

L'invention concerne un procédé de transmission de données, comprenant les étapes consistant à fournir N valeurs (X) à transmettre, où N est une puissance de 2 ; diviser les valeurs dans M vecteurs (Xk) de N/M valeurs, où M est une puissance de 2 ; appliquer chaque vecteur en parallèle à deux groupes (80a, 80b) de transformées de Fourier inverse à M N/M points, produire un premier et un second groupe de M sous-symboles (ak, bk) ; modifier l'entrée de chaque transformée de Fourier inverse par une séquence de nombres pseudo-aléatoires différente (PRNA, PRNB) ; produire (84) des combinaisons de sommes par paires (σu,v) de sous-symboles du premier groupe avec des sous-symboles du second groupe ; et sélectionner (72) pour la transmission la somme (σp,q) ayant le plus faible rapport de puissance de crête-à-moyenne (PAPR).
PCT/IB2015/055494 2015-07-20 2015-07-20 Systèmes de transmission ofdm avec un rapport de puissance crête-à-moyenne réduit Ceased WO2017013467A1 (fr)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US10848359B2 (en) 2016-09-30 2020-11-24 Qualcomm Incorporated Virtual symbol splitting techniques in wireless communications

Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2015028684A2 (fr) 2013-09-02 2015-03-05 Denis Julien Gilles Mestdagh Procédé pour générer un symbole construit pour une transmission de données et procédé pour démoduler un symbole construit reçu

Patent Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2015028684A2 (fr) 2013-09-02 2015-03-05 Denis Julien Gilles Mestdagh Procédé pour générer un symbole construit pour une transmission de données et procédé pour démoduler un symbole construit reçu

Non-Patent Citations (5)

* Cited by examiner, † Cited by third party
Title
CHUNG H ET AL: "A New SLM OFDM Scheme With Low Complexity for PAPR Reduction", IEEE SIGNAL PROCESSING LETTERS, IEEE SERVICE CENTER, PISCATAWAY, NJ, US, vol. 12, no. 2, 1 February 2005 (2005-02-01), pages 93 - 96, XP011124973, ISSN: 1070-9908, DOI: 10.1109/LSP.2004.840915 *
DENIS J.; G. MESTDAGH; PAUL M. P. SPRUYT: "A Method to Reduce the Probability of Clipping in DMT-Based Transceivers", IEEE TRANSACTIONS ON COMMUNICATIONS, vol. 44, no. 10, October 1996 (1996-10-01)
GHASSEMI A ET AL: "Partial Selective Mapping OFDM with Low Complexity IFFTs", IEEE COMMUNICATIONS LETTERS, IEEE SERVICE CENTER, PISCATAWAY, NJ, US, vol. 12, no. 1, 1 January 2008 (2008-01-01), pages 4 - 6, XP011203095, ISSN: 1089-7798 *
KEE-HOON KIM ET AL: "A new selected mapping scheme for PAPR reduction in OFDM systems", INFORMATION THEORY AND ITS APPLICATIONS (ISITA), 2010 INTERNATIONAL SYMPOSIUM ON, IEEE, PISCATAWAY, NJ, USA, 17 October 2010 (2010-10-17), pages 1054 - 1057, XP031811180, ISBN: 978-1-4244-6016-8 *
YUH-REN TSAI ET AL: "A low-complexity SLM approach based on time-domain sub-block conversion matrices for OFDM PAPR reduction", COMPUTERS AND COMMUNICATIONS (ISCC), 2011 IEEE SYMPOSIUM ON, IEEE, 28 June 2011 (2011-06-28), pages 579 - 584, XP032018454, ISBN: 978-1-4577-0680-6, DOI: 10.1109/ISCC.2011.5983900 *

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US10848359B2 (en) 2016-09-30 2020-11-24 Qualcomm Incorporated Virtual symbol splitting techniques in wireless communications

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