WO2013111968A1 - Method for current control pulse width modulation of multiphase full bridge voltage source inverter - Google Patents

Description

Current Controlled Pulse Width Modulation of Multiphase Full Bridge Voltage Source Inverter

The present invention relates to a current control pulse width modulation method of a multiphase full bridge voltage source inverter, and more particularly, to a current control pulse width modulation method of a polyphase full bridge voltage source inverter for driving an independent polyphase permanent magnet synchronous motor.

Independent Multiphase Permanent Magnet Synchronous Motor Introduction

The concept of so-called independent polyphase motors, with no neutral connection in the windings of the phases, was proposed by Non-Patent Document 1 in JAHNS as early as 1980. In 2000, MARTIN et al. Presented non-patent document 2 showing the results of applying an independent polyphase permanent magnet synchronous motor to the electric power propulsion of a ship. The main advantages of the independent polyphase permanent magnet synchronous motor are as follows.

First, since power is distributed in multiple phases, the inverter is easy to manufacture, which is advantageous for high power applications.

Second, since the current of each phase can be controlled independently, it is advantageous for fault tolerance to operate as the remaining phase when some phases fail.

Third, the canceling effect of torque ripples generated in each phase due to non-ideal control such as commutation torque is large, which is advantageous for low noise characteristics.

Motors to which the present invention is applied

An electric machine is a device that converts electrical energy into mechanical energy or mechanical energy into electrical energy, and is also called an electric motor or a generator depending on the purpose of use. A multiphase motor is an electric motor including two or more armature windings (hereinafter referred to as a phase winding), and is divided into two phases, three phases, and four phases according to the number of phase windings. It is further divided into six poles, and further divided into radial magnetic flux type and axial magnetic flux type according to the direction of magnetic flux, and further divided into overlapping and non-redundant winding, total power and disconnection right, centralized right and distributed right according to winding. Depending on the layout, it is further divided into integer slots and fractional slots. Synchronous motors are further divided into electronic type, permanent magnet type, and reluctance type, depending on how the field is formed. Permanent magnet synchronous motors are further divided into surface-attached permanent magnet types and internal permanent magnet types. The permanent magnet synchronous motors are further divided into a pole type and a non-pole type according to the magnetic properties of the field. Among permanent magnet synchronous motors, those with sinusoidal waveforms of electromotive force are called brushless AC motors, and those with trapezoidal waveforms of electromotive force are also called brushless DC motors.

이 다음의 수식으로 된 것이며 반상 진행 방식이라고 부른다.The polyphase method can be divided into two types. The first kind is phase angle of each phase

This is the following formula and is called the reverse phase method.

Equation 1

Where N is the number of phases. 1 is a cross-sectional view showing a 12-phase electric motor of a half-phase running system. The motor of FIG. 1 has the characteristics of two poles, radial magnetic flux, permanent magnet with surface, non-pole type, overlapping winding, total winding, concentrated winding, and integer slot. In the second type, the phase angle of each phase is expressed by the following equation, and it is called the phase shift method. 2 is a cross-sectional view showing a 24-phase electric motor of a full-phase propagation method.

Equation 2

As such, polyphase motors can have various differences. However, even with these differences, other features, such as the phase variable model and the operating principle, are almost identical. The present invention can be applied to all polyphase motors and generators similar in principle of operation to the phase variable model.

Phase Parameter Model of Independent Polyphase Permanent Magnet Synchronous Motor

A phase variable model of a general independent polyphase permanent magnet synchronous motor is known as follows. This model ignores reluctance torque and cogging torque and can be added to the model or included in the reference torque if reflection of reluctance torque and / or cogging torque is required.

Equation 3

는 상 권선의 전압,

는 상 권선의 인덕턴스 행렬 요소,

는 상 권선의 전류, R은 상 권선의 등가 저항, 및

는 상 권선의 기전력이다.In the above formula,

Is the voltage of the phase winding,

Is the inductance matrix element of the phase winding,

Is the current of the phase winding, R is the equivalent resistance of the phase winding, and

Is the electromotive force of the phase winding.

Write the above expression as follows:

Equation 4

는 상권선의 전압 벡터,

은 상 권선의 인덕턴스 행렬,

는 상 권선의 전류 벡터, 는 상 권선의 기전력 벡터이다.In the above formula,

Is the voltage vector of the winding

Inductance matrix of silver phase winding,

Is the current vector of the phase winding, Is the electromotive force vector of the phase winding.

Obtaining a phase variable model in practical applications is possible by analytical methods, measurement methods, and estimation methods using operational data.

Operating Principle and Optimum Current of Independent Polyphase Permanent Magnet Synchronous Motor

The energy conversion model of an independent polyphase permanent magnet synchronous motor is known as follows.

Equation 5

은 코일 변 자속 분포,

은 회전자의 각도,

은 회전자의 각속도, T(t)는 회전자의 토크이다.In the above formula,

Silver coil flux distribution,

Is the angle of the rotor,

Is the angular velocity of the rotor and T (t) is the torque of the rotor.

Given the angle of the rotor and hence the coil flux distribution of each phase, the currents of each phase producing any desired torque are numerous. Among them, the optimum current for minimizing the power loss in the equivalent resistance while generating the desired torque is obtained as follows.

Equation 6

In the case of an open circuit fault, the fault tolerated phase is excluded from the equation, resulting in a fault tolerance optimum current. It is also possible to determine the optimal current of the remaining phases while specifying the current of some phases.

Prior Art of Current Controlled Pulse Width Modulation Method

When the reference current for each phase winding is obtained by the optimization method described above, or by some other method, controlling the output voltage of the inverter so that the actual current follows the reference current is called current control. However, each phase single-phase full bridge voltage source inverter can output only three types of voltages. Therefore, the output voltage is controlled by pulse width modulating the voltage of each phase, which is called a current control pulse width modulation method.

The prior art of the current controlled pulse width modulation method for driving permanent magnet synchronous motors is mostly for three phase Y-connected motors. Basically, the magnetic coupling between phases is small in three phases, but in the Y-connection system, this is canceled out and does not appear in operation. If the linear feedback current control pulse width modulation method or the predictive current control pulse width modulation method applied to a motor without magnetic coupling between phases is extended to an independent polyphase permanent magnet synchronous motor with a large magnetic coupling between phases, It does not show good performance. These points are also mentioned in Non-Patent Document 2, and Non-Patent Document 2 has proposed a hysteresis current control pulse width modulation method as a method for solving this problem. The hysteresis current control pulse width modulation method has the advantages of particularly excellent adaptability and current tracking characteristics, but has the following disadvantages. This disadvantage is particularly problematic in applications where high power and low noise characteristics are required.

First, the switching frequency of a polyphase full bridge voltage source inverter is very irregular.

Second, large switching ripples occur in the torque waveform.

As another prior art, vector control methods are known for neutral point polyphase motors. These methods are characterized by using a synchronous transformation variable model rather than a phase variable model. These methods have the advantages of constant switching frequency characteristics, excellent current tracking characteristics, and adaptability, but basically have a limitation that the synchronous conversion method assumes a polyphase balanced circuit. Therefore, the fault-tolerant operation that results in a polyphase unbalance circuit shows poor current tracking characteristics.

(Patent Document 1) US 7312592 B2 (MASLOV et al.) 2007. 12. 25.

(Non-Patent Document 1) JAHNS, T. M. 'Improved reliability in solid-state ac drives by means of multiple independent phase-drive units', IEEE Transactions on Industry Applications. Vol. IA-16, No. 3, 1980, p 321-331.

(Non-Patent Document 2) MARTIN, J.-P. ; MEIBODY-TABAR, F.; DAVAT, B. permanent magnet synchronous machine supplied by VSIs, working under fault conditions In: Industry Applications Conference, 2000. Conference Record of the 2000 IEEE, 2000, p. 1710-1717.

For applications requiring high power, fault tolerance operation, and / or low noise operation, which are advantages of independent polyphase permanent magnet synchronous motors, the following three performance characteristics for current-controlled pulse width modulation methods can be applied individually or As a whole required.

First, constant switching frequency characteristics for high power applications.

Second, excellent current tracking characteristics for fault tolerance operation.

Third, low switching torque ripple for low noise operation.

The technical problem of the present invention is to achieve the above three performance characteristics individually or entirely.

The technical features of the present invention for achieving the above three performance characteristics are two.

The first technical feature is to achieve a low switching torque ripple characteristic, which staggers the center positions of the output voltage pulses of a multiphase full bridge voltage source inverter.

The second technical feature is to obtain a constant switching frequency and excellent current tracking characteristics. The reference of the multiphase full bridge voltage source inverter using a phase variable model including the inductance matrix of the independent multiphase permanent magnet synchronous motor for normal operation and fault tolerance operation. Find the voltage. The second technical feature is embodied in the linear feedback current control pulse width modulation method and the predictive current control pulse width modulation method.

The above two technical features contribute individually or as a whole to achieve three performance characteristics. Therefore, the present invention will be described as a group of inventions that form one concept of a general invention.

First invention

The prior art is characterized by including a method of equalizing the center positions of the output voltage pulses of a polyphase full bridge voltage source inverter. 3 shows an example in which the preceding pulse width modulation method is implemented in a half-phase running 12-phase motor.

In the current control pulse width modulation method of a multi-phase full bridge voltage source inverter for driving an independent multi-phase permanent magnet synchronous motor, the multi-phase full bridge voltage source inverter to reduce the switching torque ripple generated in the independent multi-phase permanent magnet synchronous motor And staggering the center positions of the output voltage pulses.

The present invention is possible in various embodiments depending on the pulse width modulation method and the method of staggering the center positions of the output voltage pulses. In the present specification, a carrier pulse width modulation method and a method of staggering center positions of output voltage pulses at regular intervals will be taken as a representative example.

By staggering the phase angles of the polyphase carriers at regular intervals, the center positions of the output voltage pulses of all phases are staggered at substantially constant intervals.

In the reversed phase scheme, the constant interval may be a modulation period divided by the number of phases. 4 shows an example in which the first invention is implemented in a half-phase running 12-phase motor.

When the magnetic coupling between the m phase and the (m + N / 2) phase is not large in the phase shift mode, the constant interval may be a modulation period divided by the number of phases. When the magnetic coupling between the m phase and the (m + N / 2) phase is very large in the phase shift mode, the constant interval may be obtained by dividing the modulation period by half the number of phases. This is because it is necessary to synchronize the m-phase and (m + N / 2) -phase voltages. 5 shows an example in which the first invention is implemented in a full-phase 24-phase electric motor.

If the two phases are not synchronized when the magnetic coupling between the m and (m + N / 2) phases is very large, the differential voltage between the two phases causes a very large cyclic current ripple through the magnetic coupling between the phases. do.

In the case of fault-tolerant operation, the number of phases may be used as it is, or the number obtained by subtracting the number of faulty phases may be used. If the magnetic coupling between the m and (m + N / 2) phases is very large in the phase shift mode, the intervals must be adjusted while keeping the m and (m + N / 2) phases synchronized. The circulating current ripple between + N / 2) phases does not increase.

By prior art making the center positions of the output voltage pulses of all phases equal, the output current ripples of all phases become almost in-phase waveforms of the modulation period, and the torque ripples in which these current ripples occur are increased by the reinforcement effect.

By staggering the center positions of the output voltage pulses of all the phases by the present invention, the output current ripples of all the phases become almost polyphase waveforms of the modulation period, and the torque ripples generated by these current ripples are reduced by the cancellation effect.

On the other hand, when the magnetic coupling between phases is large, the present invention has a side effect of slightly increasing the circulating current ripple between phases by staggering the center positions of the output voltage pulses of all the phases.

Second invention

The prior art is a linear feedback current controlled pulse width modulation method, comprising a method of obtaining a reference voltage of a multiphase full bridge voltage source inverter without including the feedforward of a phase variable model including an inductance matrix of an independent polyphase permanent magnet synchronous motor. It is set as (refer patent document 1).

The present invention is a current control pulse width modulation method of a multi-phase full bridge voltage source inverter for driving an independent polyphase permanent magnet synchronous motor, wherein the current control pulse width modulation method is a linear feedback current control pulse width modulation method, and the normal operation and failure And a method of obtaining a reference voltage of the multiphase full bridge voltage source inverter including a feedforward of a phase variable model including an inductance matrix of the independent polyphase permanent magnet synchronous motor for each endurance operation. This is expressed as a formula:

Equation 7

는 기준 전압 벡터,

은 미리 구해놓은 인덕턴스 행렬,

는 입력받은 기준 전류 벡터, R은 미리 구해놓은 등가 저항,

는 입력받은 기전력 벡터, r은 미리 정해 놓은 비례 이득, 및

는 측정한 전류 벡터이다.In the above formula,

Is the reference voltage vector,

Is a previously obtained inductance matrix,

Is the reference current vector received, R is the equivalent resistance,

Is the received electromotive force vector, r is a predetermined proportional gain, and

Is the measured current vector.

The above formula is a representative example, and it is possible to use a somewhat modified formula. For example, you can ignore equivalent resistance or reflect reluctance torque. Linear feedback may use a proportional-integral compensator or the like. The reference current derivative vector can be found numerically using the finite time difference method, or with the derivative of an analog electronic circuit.

The linear feedback current control pulse width modulation method is a method of obtaining a reference voltage of a voltage source inverter through a proportional-integral compensator for a current error signal, and controlling current through pulse width modulation of the reference voltage. Therefore, by adopting the pulse width modulation method having a constant switching frequency characteristics, such as the carrier pulse width modulation method, it is possible to achieve a constant switching frequency characteristics of the current control pulse width modulation method. However, when the prior linear feedback current control pulse width modulation method is applied to an independent multiphase permanent magnet synchronous motor having a large magnetic coupling between phases, the current tracking characteristics are inferior due to the magnetic coupling action between the phases. Inferior current tracking characteristics are particularly problematic in fault tolerant operation.

The present invention feedforwards a phase variable model including an inductance matrix of an independent polyphase permanent magnet synchronous motor for normal operation and fault tolerance operation, thereby compensating for the magnetic coupling action between the phases, and then normal operation and fault tolerance operation. All show excellent current tracking characteristics.

While the present invention shows robust characteristics with respect to the error included in the measured current, it does not with respect to the error included in the phase variable model of the motor. Therefore, it is necessary to obtain a precise phase variable model in advance, and it may be effective to add a method of estimating and reflecting the change in the model even during operation.

Third invention

The prior art is a predictive current control pulse width modulation method, comprising a method of obtaining a reference voltage of a multiphase full bridge voltage source inverter using a phase variable discrete time model that does not include an inductance matrix of an independent polyphase permanent magnet synchronous motor. It is.

The present invention is a current control pulse width modulation method of a multi-phase full bridge voltage source inverter for driving an independent polyphase permanent magnet synchronous motor, wherein the current control pulse width modulation method is a predictive current control pulse width modulation method, and the normal operation and fault tolerance. And a method of obtaining a reference voltage of the multiphase full bridge voltage source inverter using a phase variable discrete time model including an inductance matrix of the independent polyphase permanent magnet synchronous motor for each operation. This is expressed as a formula:

Equation 8

은 현재 표본화 시각;

은 다음 표본화 시각;

는 표본화 주기;

,

, 및

는 각각 첫째, 둘째, 및 셋째 표본화 시각;

는 기준 전압 벡터;

은 미리 구해놓은 인덕턴스 행렬;

는 입력받은 기준 전류 벡터; 는 추정한 전류 벡터; R은 미리 구해놓은 등가 저항; 및

는 입력받은 기전력 벡터이다.In the above formula;

Is the current sampling time;

Is the next sampling time;

Is the sampling cycle;

,

, And

Are the first, second, and third sampling times, respectively;

Is a reference voltage vector;

Is a previously obtained inductance matrix;

Is the received reference current vector; Is an estimated current vector; R is the equivalent resistance obtained in advance; And

Is the received electromotive force vector.

The above formula is a representative example, and it is possible to use a somewhat modified formula. For example, you can ignore equivalent resistance or reflect reluctance torque. It is also possible to use a dead bit controller, a discrete time model with an exponential matrix.

The estimated current vector may be a current vector measured at that time. The estimated current vector may be an average value of a plurality of current vectors measured at times near that time.

The sampling period may be the same as the modulation period or carrier period of the pulse width modulation method, or may be different.

There are several choices for sampling times. For example, you can choose from the following formulas:

Equation 9

The predictive current control pulse width modulation method is a method of obtaining a reference voltage of a voltage source inverter using a discrete time model of an electric motor, and controlling current through pulse width modulation of the reference voltage. Therefore, by adopting the pulse width modulation method having a constant switching frequency characteristics, such as the carrier pulse width modulation method, it is possible to achieve a constant switching frequency characteristics of the current control pulse width modulation method. However, when applying the predictive current control pulse width modulation method without the inductance matrix to an independent multiphase permanent magnet synchronous motor having a large magnetic coupling between phases, the current tracking characteristics are inferior due to the magnetic coupling action between the phases. Inferior current tracking characteristics are particularly problematic in fault tolerant operation.

By using the phase variable discrete time model including the inductance matrix of the independent polyphase permanent magnet synchronous motor for each of the normal operation and the fault tolerance operation according to the present invention, the magnetic coupling action between the phases is compensated, and then the normal operation and fault tolerance Excellent driving performance is shown in both operation.

In contrast to the second invention, the present invention exhibits robust characteristics with respect to the error included in the phase variable model of the motor, while not the error included with the measured current. Therefore, it is necessary to reduce the error or noise included in the measured current through means such as a filter.

가 작을수록 제어기가 기준 전류의 빠른 변화를 잘 추종하지만 민감하다.

가 클수록 덜 민감하지만 기준 전류의 빠른 변화를 추종하는 특성은 떨어진다.At sampling times, generally

The smaller is, the better the controller follows the fast change in reference current, but more sensitive.

The larger the value, the less sensitive it is, but the characteristic of following the fast change of the reference current is inferior.

Binding between inventions

The first invention can be combined with all pulse width modulation methods with constant switching frequency characteristics. The first and second inventions can be combined, and the first and third inventions can be combined.

The second and third inventions can be combined with the preceding pulse width modulation methods, respectively.

It is obvious that the first invention can be combined with direct torque control, scalar control, etc., rather than current control. Such combinations, even if not stated in the appended claims, are to be regarded as equivalent.

It is obvious that the second and third inventions can be applied to other circuit methods such as neutral point wiring. Such application, even if not stated in the appended claims, is to be regarded as an equivalent.

Simulation program

In order to more clearly disclose the present invention, a basic simulation program using Matlab, a computer programming language, is proposed. The program presented is for a typical half-phase motor. The combination of the first invention and the preceding pulse width modulation method, and the second and third inventions can optionally be applied.

In the simulation, the virtual model of the actual motor used the following equation. Since the change in the angular velocity of the rotor is hardly related to the performance of the current control pulse width modulation method, it is kept constant.

Equation 10

In the simulation program, the inductance matrix simulates a motor with a large magnetic coupling between the phases, and the coil flux distribution simulates a motor with a trapezoidal wave shape.

The following is a list of programs written in Matlab language.

function main

clear all; clc; close all

No = 12; N = 12; Ks = N * 50; fs = 4.5e3; Ts = 1 / fs; dt = Ts / Ks;

t1 = 0; t2 = 1/45; NK = round ((t 2 -t 1) / dt);

Vdc = 600; f = 10; w = 2 * pi * f; Tmax = 17e3; Tr = Tmax * 1;

i = zeros (N, NK); v = zeros (N, NK); e = zeros (N, NK); ir = zeros (N, NK);

T = zeros (1, NK); vr = zeros (N, NK); t = t1;

vc = zeros (N, NK); r = 0.1; is = zeros (N, 1);

Ls = 200e-6; Lg = 0.1 * Ls; R = 0.02;

for k = 1: 2 * No; thk = (k−1) / No * pi;

a = 0.05; q = 4; b = 1 / pi-a / (q + 1) * (pi / 2) ^ q;

if thk <pi c (k, 1) = b * pi-2 * b * thk-2 * a / (q + 1) * (thk-pi / 2) ^ (q + 1);

else c (k, 1) =-3 * b * pi + 2 * b * thk + 2 * a / (q + 1) * (thk-3 * pi / 2) ^ (q + 1); end; end

for m = 1: N; for k = 1: N; L (m, k) = Ls * c (mod (k-m, 2 * No) +1); end; end

L 2 = L + Lg * eye (N, N); L2inv = inv (L2);

for Loop = 1: 1

i (:, 1) = i (:, NK); v (:, 1) = v (:, NK); vr (:, 1) = vr (:, NK);

for K = 1: NK; t = t + dt; th = w * t;

e (:, K) = w * PsiH (th, No, N);

e (:, K + Ks / 2) = w * PsiH (th + w * Ts / 2, No, N);

ir (:, K) = Tref (t, Tr) * AlpH (th, No, N);

ir (:, K + 1) = Tref (t + dt, Tr) * AlpH (th + w * dt, No, N);

ir (:, K + Ks) = Tref (t + Ts, Tr) * AlpH (th + w * Ts, No, N);

ir (:, K + Ks / 2) = Tref (t + Ts / 2, Tr) * AlpH (th + w * Ts / 2, No, N);

%%% Second Invention

vr (:, K + 1) = L2 * (ir (:, K + 1) -ir (:, K)) / dt + R * ir (:, K) + e (:, K) + r * ( ir (:, K) -i (:, K));

%%% Third Invention

% Ns = 10; if mod (K, Ks / Ns) == 1 is = is + i (:, K); end

% vr (:, K + 1) = vr (:, K);

% if mod (K, Ks) == 1

% is = is / Ns;

% vr (:, K + 1) = L2 * (ir (:, K + Ks / 2) -is) / Ts + R * ir (:, K + Ks / 2) + e (:, K + Ks / 2);

% is = zeros (N, 1); end

for m = 1: N

vc (m, K) = Vdc * (abs (2-2 * mod (K− (m−1) / N * Ks, 2 * Ks) / Ks) -1); %%% First Invention

% vc (m, K) = Vdc * (abs (2-2 * mod (K, 2 * Ks) / Ks) -1); %%% Leading Pulse Width Modulation Method

if vr (m, K + 1)> vc (m, K) vp = Vdc / 2; else vp = -Vdc / 2; end

if -vr (m, K + 1)> vc (m, K) vn = Vdc / 2; else vn = -Vdc / 2; end

v (m, K + 1) = vp-vn; end

i (:, K + 1) = i (:, K) + L2inv * (v (:, K + 1) -e (:, K) -R * i (:, K)) * dt;

T (K) = e (:, K) '* i (:, K) / w; end;

end

for m = [1: 1]

figure; hold on; grid on; axis ([1 NK-1000 1000]);

plot (ir (m, :), 'r'); plot (i (m, :), 'b'); end

for m = [1: 1]

figure; hold on; grid on; axis ([1 NK-1000 1000]);

plot (e (m, :), 'k'); plot (vr (m, :), 'r'); plot (v (m, :), 'b'); end

figure; hold on; grid on; axis ([1 NK 0 30]);

plot (T / 1000, 'b');

figure; hold on; grid on; axis ([0 * Ks 2 * Ks 0 39])

for m = 1: N; plot (v (m,:) / Vdc + 39-3 * m, 'b'); end

function y = PsiH (th1, No, N)

n = 2; Vmax = 400; wmax = 2 * pi * 45;

for m = 1: N

th = mod (th1- (m−1) / No * pi, 2 * pi);

x = 2 * sin (th);

a = 1; b = 0.005; c = 1.5; d = 0.15; e = 15; f = 6;

y (m, 1) = (sqrt ((x + a) ^ 2 + b) -sqrt ((xa) ^ 2 + b)) / (2 * a)-(c * x) / ((c * x ) ^ f + d) / e;

y (m, 1) = y (m, 1) * n * Vmax / wmax; end

function y = AlpH (th, No, N)

y = PsiH (th, No, N) / norm (PsiH (th, No, N)) ^ 2;

function y = Tref (t, Tr)

y = Tr; if t <1/90 y = Tr / (1/90) * t; end

In the program, No is the number of original phases, N is the number of phases after failure, Ks is the discrete time value of the modulation period, fs is the switching frequency, Ts is the discrete time value of the switching frequency, dt is the finite time difference, t1 starts the simulation. Time, t2 is the end time, NK is the number of times between start and end, Vdc is the DC voltage, f is the frequency of the rotor, w is the angular velocity of the rotor, Tmax is the maximum torque, Tr is the reference torque, and i is Current vector, v is output voltage vector, e is electromotive force vector, T is torque, vr is reference voltage vector, t is current time, vc is carrier voltage vector, r is proportional gain, Ls is common magnetic inductance for each phase, Lg Is the leakage inductance common to each phase, R is the equivalent resistance common to each phase, thk is the phase angle of the kth phase, L2 is the inductance matrix model, L2inv is the inverse of the inductance matrix model, Loop is the number of times to increase the simulation time, and th is The angle of electron, PsiH, is the coil variation flux distribution beck AlpH is the function to find the rotor, AlpH is the function to find the current shape vector

Simulation result

6 is an example of a half-phase progression 12-phase simulation result for the prior hysteresis pulse width modulation method. It shows good current tracking characteristics, but the switching frequency is very irregular and large switching ripples occur in the torque waveform.

7 is an example of a half-phase progression 12-phase simulation result of the combination of the second invention and the preceding pulse width modulation method. Good current tracking and constant switching frequency, but high switching torque ripple.

8 is an example of a reverse phase 12 phase simulation result of the combination of the second invention and the first invention. Although excellent current tracking characteristics, constant switching frequency characteristics, and low switching torque ripple characteristics are all exhibited, the ripple increases slightly in the current waveform.

9 is an example of a reverse phase 12 phase simulation result of the combination of the third invention and the first invention. It shows almost the same characteristics as the result of FIG. 8 for the combination of the second invention and the first invention.

Fig. 10 is an example of simulation results of a reversed-phase progression 12-phase fault tolerance operation for the combination of the third invention and the first invention. The case where the 11th and 12th phases are faulty is simulated. Even in fault tolerance operation, it shows excellent current tracking characteristics, constant switching frequency characteristics, and low switching torque ripple characteristics.

11 is an example of a phase shift 24-phase simulation result of the combination of the third invention and the preceding pulse width modulation method. Good current tracking and constant switching frequency, but high switching torque ripple.

12 is an example of a phase shift 24-phase simulation result of the combination of the third invention and the first invention. Although excellent current tracking characteristics, constant switching frequency characteristics, and low switching torque ripple characteristics are all exhibited, the ripple increases slightly in the current waveform.

As described above, the present invention achieves the following three performance characteristics individually or entirely by two technical features.

First, constant switching frequency characteristics for high power applications.

Second, excellent current tracking characteristics for fault tolerance operation.

Third, low switching torque ripple for low noise operation.

1 is a cross-sectional view showing a half-phase running 12-phase motor.

2 is a cross-sectional view showing a full-phase 24-phase electric motor.

3 shows an example in which the preceding pulse width modulation method is performed on a half-phase running 12-phase motor.

Figure 4 is an example in which the first invention is carried out in a half-phase progress 12-phase electric motor.

5 is an example in which the first invention is implemented in a full-phase 24-phase electric motor.

6 is an example of a half-phase progression 12-phase simulation result for the prior hysteresis pulse width modulation method.

7 is an example of a reversed phase 12-phase simulation result of the combination of the second invention and the preceding pulse width modulation method.

8 is an example of reverse phase 12 phase simulation results for the combination of the second invention and the first invention.

Figure 9 is an example of the reverse phase 12 phase simulation results for the combination of the third invention and the first invention.

Figure 10 is an example of a half-phase progress 12-phase fault endurance operation simulation results of the combination of the third invention and the first invention (when the 11th phase and the 12th phase is a failure).

11 is an example of a phase shift 24-phase simulation result of the combination of the third invention and the preceding pulse width modulation method.

12 is an example of a phase shift 24-phase simulation result of the combination of the third invention and the first invention.

The best mode for carrying out the invention depends on the required performance characteristics and circumstances.

Where low switching torque ripple characteristics are required and an accurate phase variable model is possible, the combination of the first and second inventions may be the best embodiment.

Where low switching torque ripple characteristics are required and accurate phase variable models are difficult, the combination of the first and third inventions may be the best embodiment.

If low switching current ripple is required and an accurate phase variable model is possible, the combination of the preceding pulse width modulation method and the second invention may be the best embodiment.

If low switching current ripple characteristics are required and accurate phase variable models are difficult, the combination of the preceding pulse width modulation method and the third invention may be the best embodiment.

Claims (3)

A current control pulse width modulation method of a multiphase full bridge voltage source inverter for driving an independent multiphase permanent magnet synchronous motor, the method comprising: output voltage of the multiphase full bridge voltage source inverter to reduce switching torque ripple generated in the independent multiphase permanent magnet synchronous motor And a method of staggering the center positions of the pulses. In the current control pulse width modulation method of a multiphase full bridge voltage source inverter for driving an independent polyphase permanent magnet synchronous motor, the current control pulse width modulation method is a linear feedback current control pulse width modulation method, each of a normal operation and a fault tolerance operation. Current control of a multi-phase full bridge voltage source inverter comprising a method of obtaining a reference voltage of the multi-phase full bridge voltage source inverter including a feed forward of a phase variable model including an inductance matrix of the independent multi-phase permanent magnet synchronous motor Pulse width modulation method. In the current control pulse width modulation method of a multiphase full bridge voltage source inverter for driving an independent polyphase permanent magnet synchronous motor, the current control pulse width modulation method is a predictive current control pulse width modulation method, which is used for normal operation and fault tolerance operation, respectively. Current reference pulse width of the multiphase full bridge voltage source inverter using a phase variable discrete time model including an inductance matrix of the independent multiphase permanent magnet synchronous motor for the multiphase full bridge voltage source inverter. Modulation method.

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