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WO1991019993A1 - Circuits de commande pour bobine magnetique d'un systeme d'imagerie par resonance magnetique - Google Patents

Circuits de commande pour bobine magnetique d'un systeme d'imagerie par resonance magnetique Download PDF

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Publication number
WO1991019993A1
WO1991019993A1 PCT/US1991/004238 US9104238W WO9119993A1 WO 1991019993 A1 WO1991019993 A1 WO 1991019993A1 US 9104238 W US9104238 W US 9104238W WO 9119993 A1 WO9119993 A1 WO 9119993A1
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WO
WIPO (PCT)
Prior art keywords
current
coil
capacitive element
voltage
circuit
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
PCT/US1991/004238
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English (en)
Inventor
Stephen Crump
Richard Rzedzian
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Advanced NMR Systems Inc
Original Assignee
Advanced NMR Systems Inc
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Advanced NMR Systems Inc filed Critical Advanced NMR Systems Inc
Publication of WO1991019993A1 publication Critical patent/WO1991019993A1/fr
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

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Classifications

    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R33/00—Arrangements or instruments for measuring magnetic variables
    • G01R33/20—Arrangements or instruments for measuring magnetic variables involving magnetic resonance
    • G01R33/28—Details of apparatus provided for in groups G01R33/44 - G01R33/64
    • G01R33/38—Systems for generation, homogenisation or stabilisation of the main or gradient magnetic field
    • G01R33/385—Systems for generation, homogenisation or stabilisation of the main or gradient magnetic field using gradient magnetic field coils
    • G01R33/3852—Gradient amplifiers; means for controlling the application of a gradient magnetic field to the sample, e.g. a gradient signal synthesizer

Definitions

  • This invention relates to producing modulated gradient fields in a magnetic resonance imaging (MRI) systems.
  • MRI magnetic resonance imaging
  • modulated gradient field waveforms are known to be useful in MRI systems.
  • pulses of constant amplitude are used in certain
  • the invention features feedback control of the resonant circuit formed by the capacitive element and field-generating coil of an MRI system.
  • a current sensing circuit is connected to the resonant circuit, for sensing the current flowing through the coil.
  • Feedback circuitry compares the sensed current to a reference signal (e.g., the desired sinusoidal current waveform), and provides an output signal prescribing the power to be supplied to the resonant circuit to cause the current flowing through the coil to more closely approximate the reference current.
  • a reference signal e.g., the desired sinusoidal current waveform
  • the capacitive element and power source are connected in series; the reference signal comprises a plurality of sinusoidal segments and at least one flat segment; the error between the sensed current and the reference signal is integrated.
  • the invention features a method for charging a storage capacitor connected in series with the magnetic coil of an MRI system.
  • a switch circuit connected in series with the capacitor and coil is opened to isolate the capacitive element from the coil.
  • a charging circuit is connected across the
  • the level of energy stored in the capacitive element is compared to a reference level, and, based on that comparison, additional energy is delivered from the charging circuit to the capacitive element, until the energy stored in the capacitive element
  • the capacitor is charged at two charging rates, to reduce parasistic effects and the errors they induce; the capacitor is first charged to a fraction of the desired level at a relatively higher rate, and then charged to the desired level at a relatively lower rate.
  • the invention provides more accurate control of the current delivered to the coil than has heretofore been achievable.
  • Fig. 1 is a simplified block diagram helpful for understanding the invention.
  • Fig. 2 is a block diagram of a preferred embodiment of the invention.
  • Fig. 3 is a timing diagram of signals produced by the structure depicted in Fig. 1 during the precharging of the capacitor and during the delivery of a sinusoidal pulse of current to the magnetic fieldgenerating coil.
  • Fig. 4 is timing diagram of signals produced by the structure depicted in Fig. 1 during precharging and during the delivery of a constant amplitude pulse to the coil.
  • generating the gradient field are a pulse of sinusoidal oscillations and a DC pulse of constant amplitude.
  • I o I x ⁇ sin (W o t) where W e is the desired angular frequency and I x is the peak current needed to yield the desired gradient field.
  • I o For a DC pulse of constant amplitude, I o should ideally have a perfectly rectangular waveform.
  • FIG. 1 Shown in Fig. 1 is a simplified block diagram of an embodiment of the invention.
  • Capacitor charging circuitry 18 is connected across capacitor 12 to pre-charge the capacitor prior to passing any current through coil 10. Control of coil current I o is handled by control
  • circuitry 20 The load driven by control circuit 20 is represented by load 22, which consists of the series combination of coil 10, capacitor 12, parasitic resistor 14, and shunt resistor 16. Circuitry 20 controls coil current I o by operating run switch 32 and by controlling voltage V p across load 22. The resultant current I o in coil 10 has a waveform determined by voltage V p , the initial capacitor voltage V co , and the impedance
  • V p is chosen to have a sinusoidal waveform.
  • load 22 is designed to have a resonant frequency equal to the desired frequency W o of sinusoidal current I o and to have a high quality factor Q. This allows control circuit 20 to achieve the desired high drive current I o by maintaining V p at the resonant frequency with a relatively low amplitude. More specifically, since the impedance of load 22 at its resonant frequency equals the pure load resistance, the drive voltage V p needed to yield I o is given by:
  • a coil with a relatively large inductance is required.
  • a gradient coil with a measured inductance L of 1,025 microHenries is used. Accordingly, for load 22 to have a resonant frequency equal to a desired current frequency of 1.00 KHz, capacitor 12 must have a capacitance C of 24.7 microfarads.
  • the series resonant system of Fig. 1 uses a feedback architecture which monitors coil current I o and adjusts driving voltage V p to maintain the desired coil current I o .
  • Current sensing amplifier 24 detects voltage V s across a small, precisely known, shunt resistor 16 and amplifies V s with gain G a to produce feedback voltage V f ,, which is proportional to coil current I o .
  • Difference node 26 subtracts voltage V f , from sinusoidal reference voltage V R provided by signal generator 28.
  • the gain G a of current sensing amplifier 24 places feedback voltage V f on the same scale as reference voltage V R so that any difference between voltages V f and V R represents undesired error V a in the magnitude of coil current I o .
  • This scale is chosen to maximize the dynamic range of the system.
  • Difference node 26 provides error signal V a to the input of run controller 30, which operates on the error signal with a phase compensating transfer function designed to provide appropriate gain and phase
  • the phase compensation is provided with a transfer function of (s + w o ) /s, where s is the Laplace operator. This amounts to an integration of all error signal frequency components from DC to w o .
  • the resultant output control voltage V b is applied to the input of power cell 34.
  • the power cell 34 drives load 22 with voltage V p , which tracks control voltage V b .
  • This feedback architecture extends the high precision of reference voltage V R to coil current I o .
  • Rzedzian discloses precharging a capacitor in the tuned circuit to a voltage corresponding to the peak capacitor voltage in the desired steady state.
  • charging circuitry 18 precharges capacitor 12 to an initial voltage V co equal to the peak capacitor voltage V x generated when load 22 resonates with the desired coil current I o .
  • the capacitor voltage signal V c is given by:
  • charging switch 36 is closed to connect current source 38 across capacitor 12.
  • a feedback architecture is used to control current source 38 to closely match the magnitude of the initial voltage V co to voltage V x calculated by the above equation.
  • voltage sensing amplifier 40 reads the voltage on capacitor 12 to provide a
  • measured capacitor voltage V cf The difference between measured capacitor voltage V cf and a reference voltage V L which prescribes the desired capacitor voltage) is fed as error signal V e . to controller 42, which, in turn, supplies charge control voltage V d to current source 38.
  • Fig. 3 illustrates the sequence in which the feedback architecture adjusts capacitor voltage V c until the measured capacitor voltage V cf precisely corresponds to reference voltage V L .
  • Controller 42 (Fig. 1)
  • a dual-rate charging scheme is used, in which the capacitor is first charged to within a few percent of the desired level at a high rate H (V d at a relatively high
  • run controller 30 initiates a sinusoidal pulse of current through gradient coil 10 by asserting control signal C2 at time t 3 . This closes run switch 32, and allows capacitor 12 to discharge into coil 10.
  • controller 30 provides energy, in the form of voltage V p , to compensate for resistive losses.
  • Run controller 30 terminates at time t ⁇ the sinusoidal pulse by opening run switch 32 at the precise instant when capacitor 12 is fully charged and coil current I o is zero, thus avoiding the need to recharge the capacitor for subsequent pulses.
  • the series configured drive circuitry of Fig. 1 is also capable of providing DC pulses of drive
  • capacitor 12 must be initialized prior to commencement of a run operation. In the DC mode, capacitor 12 is precharged as for a sinusoidal pulse to the V co required to generate the desired coil current I x equal to I DC .
  • controller 30 asserts run control signal C2 to close run switch 32, allowing capacitor 12 to discharge into coil 10 with the same resonant current flow which produced the sinusoidal cycles described above; but the tuned circuit is only allowed to oscillate for a quarter cycle.
  • run control signal C2 to close run switch 32, allowing capacitor 12 to discharge into coil 10 with the same resonant current flow which produced the sinusoidal cycles described above; but the tuned circuit is only allowed to oscillate for a quarter cycle.
  • T 1 when capacitor voltage V c has reached zero and coil current I o has reached I De' run controller 30 asserts capacitor bypass control signal C3, thereby closing bypass switch 46 to effectively short capacitor 12. Once capacitor 12 is bypassed, power cell 34 maintains the coil current at level I De .
  • run controller 30 turns off bypass control signal C3, thereby opening switch 46. This initiates another sinusoidal pulse segment, in which capacitor 12 charges through a quarter cycle of resonant current flow, until capacitor voltage V c reaches its maximum and coil current I o reaches zero. At this point (T 3 ), run controller 30 turns off control signal C2, thereby opening run switch 32 and terminating the run operation.
  • a positive pulse having three segments: a constant amplitude segment bounded by two sinusoidal segments each consisting of a quarter cycle of a resonant sinusoid.
  • Other waveforms can also be constructed.
  • the sinusoidal segments can be larger than a quarter cycle; an integral number of quarter cycles can be used.
  • the inverse of the above described positive pulse can be achieved by
  • Rectifiers and bridge circuits such as disclosed in Rzedzian U.S. Patent No. 4,628,264, may be employed to achieve half and full wave rectification of the
  • a separate precharge feedback architecture comprising capacitor charging circuitry 18 must control the initial capacitor voltage V co to within a small allowable error.
  • the degree of precision required relates in part to the fact that load 22 is designed to have a high quality factor Q. This results in high voltages across capacitor 12 and coil 10 using only a relatively low voltage V p . As a result, a small error in initializing capacitor 12 can
  • Charging circuit 18 is accordingly designed to precharge capacitor 12 to within a fraction of 1% of the ideal level. This is achieved by using the feedback architecture described above. To assure precision in the feedback circuitry, voltage sensing amplifier 40 is a high quality differential amplifier with common-mode rejection. This assures that the feedback signal V cf accurately represents the capacitor voltage. The series inductances in charging circuit 38 are also carefully minimized to avoid overcharging. To minimize any
  • a dual-rate charging scheme is used in which the charging rate drops to a relatively slow rate as the capacitor voltage approaches the desired level.
  • the temperature of the capacitor is kept constant by being positioned in the incoming airstream of the power cell chassis. Further, the capacitors themselves have a low dissipation factor and low temperature coefficient of capacitance. A bank of twelve 2.0 microfarad capacitors with a 2KV peak rating are combined with sufficient trim capacitors (all of General Electric series 28F5600). Steps are also taken to assure stability in control circuitry 20. Run controller 30, difference node 26, and reference signal generator 28 are implemented with a digital computer clocked by a highly stable crystal oscillator.
  • run controller 30 integrates all low frequency components of error signal V a . Since run controller 30 cannot be
  • current sensing amplifier 24 is a high quality differential amplifier chosen to have DC stability. Further, as with capacitor 12, airflow is provided over amplifier 24 to minimize temperature variations and further reduce drift.
  • Fig. 2 shows the preferred embodiment.
  • a single, bipolar power cell 110 is used to provide energy for both precharge and run operations.
  • controller 112 controls the operation of power cell 110 during both operations.
  • a digital control signal on line 111 prescribing the output of the power cell is delivered by controller 112, and converted by DAC 114 into a corresponding analog voltage V B .
  • Differential output terminals 116, 118 are connected across the series resonant load, which consists of capacitor 122, shunt resistor 124, and coil 126.
  • Terminals 116, 118 are also connected to capacitor charge circuit 120 to supply energy during the precharge
  • the multiplexing of the output of power cell 110 between the two operations of the precharging and running is controlled by digital controller 112.
  • the controller activates capacitor charge circuit 120 by asserting control signals CS1, CS2, CS3 to close charging switches 130, 132, 134 (SCR1, SCR2, SCF3, SCR4).
  • controller 112 asserts control signal CS4 to close run switch 136 (SCR5, SCR6).
  • controller 112 supplies a 15 KHz sinusoidal input to the power cell input line 111 causing the power cell to produce a corresponding 15 KHz sinusoidal voltage V p across
  • Controller 112 next asserts control signal CSl to close charging switch 130.
  • the 15 KHz current flows in primary coil 138 of step-up transformer 139 and provides a 15 KHz voltage at the output of each of secondary coils 140, 142.
  • Low inductance connections are used on the primary side of transformer 139 (e.g., for lines 116, 118) to minimize parasitic inductance.
  • the voltage generated by secondary coil 140 is applied to high-voltage rectifier 144 which produces a corresponding positive DC voltage across terminals 146, 148.
  • the voltage generated by secondary coil 142 is applied to high-voltage rectifier 150 to produce a negative DC voltage across terminals 148, 152.
  • Controller 112 selects a positive precharge voltage by asserting control signal CS2 to close charging switch 132.
  • controller 112 instead asserts control signal CS3, thereby causing the capacitor voltage to reach the negative level set by the output of rectifier 150.
  • feedback architecture is used to precisely control the magnitude of the capacitor precharge voltage.
  • the voltage across capacitor 122 is read by voltage sensing amplifier 156 and converted to digital form by ADC 158.
  • Controller 112 compares the sensed capacitor voltage with a predetermined digital reference voltage stored in the controller's memory. Based on the result of that comparison, controller 112 adjusts the power cell output. Using a dual rate charging scheme, controller 112 brings the capacitor voltage to the
  • controller 112 begins a run operation by asserting control signal CS4 to close run switch 136 and thereby initiate current flow into coil 126.
  • Voltage sensing amplifier 160 reads the voltage across shunt resistor 124 to provide a
  • the voltage is compared with a preset digital signal stored in the controller.
  • Controller 112 internally calculates an error signal (V a , Fig. 1) and an appropriate correction signal (V b , Fig. 1). The appropriate correction signal is then supplied to power cell 110 on input line 111. In this manner, controller 112 implements the operation of run controller 30, charge control circuitry 42, voltage references V R and V L , and difference nodes 26, 44 (Fig. 1) . Status monitor 180 provides an operator interface. To minimize the DC offset problems discussed above, both sensing amplifier 160 and ADC 162 should be chosen to minimize DC offset. Further, proper airflow over these components should be provided to minimize temperature variation.

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  • Physics & Mathematics (AREA)
  • Condensed Matter Physics & Semiconductors (AREA)
  • General Physics & Mathematics (AREA)
  • Magnetic Resonance Imaging Apparatus (AREA)

Abstract

Commande de réaction pour un circuit résonnant (22) formé par un élément capacitif (12) et une bobine génératrice de champ (10) d'un système d'imagerie par résonance magnétique. Un circuit détecteur de courant (24) est connecté au circuit résonnant pour détecter le courant qui passe par la bobine (10). Des circuits de réaction comparent le courant détecté à un signal de référence (28) (par exemple, la forme d'onde de courant sinusoïdale souhaitée), et fournissent un signal de sortie correspondant au courant devant être appliqué au circuit résonnant (22) afin que le courant traversant la bobine s'approche davantage du courant de référence.
PCT/US1991/004238 1990-06-13 1991-06-13 Circuits de commande pour bobine magnetique d'un systeme d'imagerie par resonance magnetique Ceased WO1991019993A1 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US53802190A 1990-06-13 1990-06-13
US538,021 1990-06-13

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WO1991019993A1 true WO1991019993A1 (fr) 1991-12-26

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2005124382A1 (fr) * 2004-06-18 2005-12-29 Siemens Aktiengesellschaft Amplificateur electrique et procede de commande associe
CN116667888A (zh) * 2023-08-02 2023-08-29 沈阳仪表科学研究院有限公司 一种极低频电磁信号发射机

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4628264A (en) * 1984-03-14 1986-12-09 Advanced Nmr Systems, Inc. NMR gradient field modulation
US4644282A (en) * 1983-10-05 1987-02-17 Siemens Aktiengesellschaft Apparatus for the formation of images of an examination subject with nuclear magnetic resonance

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4644282A (en) * 1983-10-05 1987-02-17 Siemens Aktiengesellschaft Apparatus for the formation of images of an examination subject with nuclear magnetic resonance
US4628264A (en) * 1984-03-14 1986-12-09 Advanced Nmr Systems, Inc. NMR gradient field modulation

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2005124382A1 (fr) * 2004-06-18 2005-12-29 Siemens Aktiengesellschaft Amplificateur electrique et procede de commande associe
US7468607B2 (en) 2004-06-18 2008-12-23 Siemens Aktiengesellschaft Electric amplifier and method for the control thereof
CN116667888A (zh) * 2023-08-02 2023-08-29 沈阳仪表科学研究院有限公司 一种极低频电磁信号发射机
CN116667888B (zh) * 2023-08-02 2023-10-20 沈阳仪表科学研究院有限公司 一种极低频电磁信号发射机

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