RU2608569C2 - System of decametric radio communication with high-speed data transmission - Google Patents

Abstract

FIELD: radio engineering and communications.

SUBSTANCE: invention relates to radio communication and can be used in wide application decameter range radio networks. For this purpose a system of decametric radio communication with high-speed data transmission includes in the transmitting complex series-connected: an additional unit of channel manipulators, an additional radio transmitter and an additional transmitting antenna, and in the receiving complex it includes an additional unit of N channel demodulators and 2N units of coherent signals addition (CSA), each CSA comprises two phasing units, each of which comprises serially connected: a channel filter, a normalizing amplifier, the first multiplier, a measuring filter and the second multiplier.

EFFECT: technical result is improvement of noise immunity of receiving data at the disturbing effect of concentrated in the spectrum sinusoidal and fluctuation interference.

1 cl, 3 dwg, 1 tbl

Description

The invention relates to radio communications and can be used in decameter radio networks of widespread use, designed to transmit high-speed data (discrete messages) using signals with angular manipulation.

A known decameter radio communication system with high-speed data transmission using a single-channel (sequential) method of transmitting discrete messages, comprising a transmitting complex comprising a series-connected encoder, modulator, radio transmitting device and a transmitting antenna, as well as a receiving complex containing a series-connected receiving antenna, a radio receiving device, demodulator and decoder [1], p. 107.

In this system, the initial information binary sequence (data) from the output of the encoder with a speed V _and = 1 / T _and (bit / s), determined by the duration of the binary element T _{and the} information sequence, manipulates a single carrier frequency in the modulator. Depending on the multiplicity of compression of the transmitted signal k [2], p. 573, the modulator can generate signals with angular manipulation, for example, such as for k = 1 - signals of a single relative phase telegraphy (OFT) or frequency telegraphy (CT) signals with a manipulation speed V _m (bit / s) equal to the binary information transfer rate V _m = V _and = 1 / T _e (T _e is the duration of the element of the signal (package) generated by the manipulator transmitted via the radio transmitting device and transmitting antenna, in this case T _e = T _and ); when k = 2 - signals twice OBT (DOFT) or double frequency telegraphy (DCHT) at a speed of manipulation is 2 times lower information rate V _m = 1 / T _e = 1 / 2T _and etc.

Demodulation of these signals can be carried out by traditional methods, for example, given in [2], and determining the structure of constructing a demodulator that implements the operation of restoring the transmitted binary sequence, the inverse of the modulation operation.

However, single-channel (sequential) data transmission systems with traditional methods of processing received signals [2] and more complex methods of processing signals [3] have the following disadvantages:

1. When using conventional techniques demodulate received signals [2] decameter radio transmission of data at high speed V _and is associated with serious difficulties due to the occurrence at the receiving side intersymbol interference due to the presence of delayed rays. If you do not take special measures (to reduce or even eliminate the harmful effects of the multipath effect), then the duration T _{e of the} element of the transmitted signal (sending) cannot be selected less than 2-3 ms, which limits the maximum manipulation speed of the transmitted signal V _m of the order of 300-500 bit / s [3].

2. An increase in the data transfer rate by increasing the multiplexing factor k while maintaining the necessary duration T _{e of the} element of the transmitted signal leads to a decrease in the noise immunity of receiving discrete information [2], p. 615.

3. Increasing the data transfer rate by reducing the duration T _{e of the} element of the transmitted signal when implementing more complex algorithms for processing the received signals, allowing to overcome the consequences of intersymbol interference, for example, as in a communication system with a test pulse and prediction (SIIP) [3], reduces interference immunity due to a decrease in the energy of the transmitted signal element [2] and the deterioration of the electromagnetic compatibility of the radio communication system due to the expansion of the transmission spectrum input signal.

In addition, the expansion of the spectrum of the transmitted signal requires a corresponding increase in the bandwidth when receiving this signal, which further reduces the noise immunity of the reception due to the increased likelihood that the spectral components of the station or spectrum-focused (interference) radio interference get into the wider reception band.

A known system of decameter radio communication with multi-channel (parallel) transmission of discrete messages, containing a transmitting complex containing a serially connected message source, an encoder and a serial-parallel converter, the outputs of which are connected to the corresponding inputs of the block N channel manipulators, the output of which is connected to a series-connected radio transmitting device and a transmitting antenna, as well as a receiving complex containing a series-connected receiving antenna, a radio A receiving device and a block of N channel demodulators, the outputs of which are connected to the corresponding inputs of the parallel-serial converter, the output of which is connected to the decoder and the receiver of messages in series [4].

In transmitting complex the system the transmitted binary sequence output from the encoder to the original data rate V _u = 1 / T _and (bit / s) converted to serial-to-parallel converter in kN parallel channel sequences with the channel rate following binary symbols in each V _CAD = V _and / kN (bits / s), where k is the multiplicity of signal compression generated by each channel manipulator as part of a block of N channel manipulators.

In block N channel manipulators, consisting of the N identical channel manipulators, e.g., phase or frequency, each arm at the initial time of each clock period of duration T _ch = kNT _and fed in parallel and synchronously k symbols corresponding k channel sequences to form a feature of the transmitted channel signal (channel parcels) of duration T _kan.

The output signals of all channel manipulators are summed up at the output of a block of N channel manipulators and a group N - channel (frequency) signal is broadcasted using a radio transmitting device and a transmitting antenna.

In the receiving complex of this system, the received group signal from the output of the radio receiver is fed to a block of N channel demodulators, consisting of N of the same type of demodulators, each of which demodulates the channel signal at the corresponding channel frequency by the selected traditional method [2].

As a result, at the outputs of the block of N channel demodulators, kN binary channel sequences are formed in parallel, which are converted by a parallel-serial converter into one binary sequence similar to the sequence at the output of the encoder of the transmitting complex, which, after decoding in the decoder, is sent to the information receiver.

The duration T of the transmitted signal _kan element in each channel becomes kN times the initial duration T _and the binary signal at the encoder output element that allows to overcome the negative effects of multipath reception signal at a relatively high data rate group.

However, the noise immunity of this decameter radio communication system is insufficient to conduct radio communications with a higher information transfer rate:

1. Increasing the data rate by increasing the k seal the multiplicity of each channel while maintaining the necessary duration T _kan element channel signal transmitted reduces the noise immunity of the reception of digital information in each channel [2].

2. Increasing the data rate by reducing the duration T of the transmitted signal _kan element in each frequency channel or by increasing the number N of frequency channels results in a corresponding expansion of the spectrum of the transmitted baseband signal and the corresponding bandwidth extension at its reception, which leads, as mentioned above, to reduce the noise immunity of receiving discrete information and the deterioration of the electromagnetic compatibility of the communication system.

Of the known decameter radio communication systems, the closest in essence to the tasks to be solved and to the majority of the essential features coinciding is the decameter radio communication system with high-speed data transmission, given in [5], p. 7.

The structure of this system with multichannel (parallel) transmission of discrete messages basically corresponds to the above communication system [4] except that in the receiving complex of this system not only can one receive signals on one antenna, but also more noise-resistant dual reception on two antennas spaced apart in space or polarization when a group multi-frequency signal is emitted in a transmitting complex in the upper or lower sideband.

Consider in more detail the operation of the receiving complex of this system.

The voltages of two samples of a group signal received at spatial or polarized antenna antennas from the outputs of the linear paths of the corresponding two radio receivers simultaneously arrive at each of N = 20 channel blocks (KB), which filter out each of the samples of the group signal to its channel signals and values proportional to the cosines and sines of the phase difference between the received sample of the channel signal and the corresponding reference oscillation are calculated.

Each channel block provides dual reception and consists of two identical active filters, which receive the corresponding group signal samples from the outputs of the linear paths of two radio receivers and the same reference oscillation from the frequency grid generator.

Each of the 2N active filters, in turn, consists of two identical correlators, characterized in that the reference oscillations supplied to them are 90 ° out of phase. The correlator contains a multiplier and an integrator, built on the basis of a DC amplifier with a large gain and an RC feedback circuit.

The output signals of the integrators of each channel block are the results of converting the corresponding two samples of the channel signal to zero frequency with the decomposition of each channel signal into two quadrature components, the voltage values of which are recorded and stored in the corresponding cells of the storage device (memory). Information on the quadrature components of each package of each channel signal in the form of analog voltage levels is stored in memory cells (on capacitors) for the duration of two adjacent packages. Moreover, from the values of the levels of each pair of quadrature components, one can calculate the amplitude and phase of the carrier wave of the corresponding channel signal.

In the subsequent phase difference calculation unit (BVRF), linear incoherent addition of each of the N pairs of samples of channel signals received to the corresponding two spaced apart, for example, in the antenna space is performed by summing the corresponding analog levels of the quadrature components recorded in the memory.

Since the decameter radio communication system under consideration uses phase difference manipulation (relative phase telegraphy) signals with multiplicity of multiplexing k = 1, or k = 2, or k = 3 to transmit information, to determine the true values of binary symbols when demodulating each of the N resulting channel signals (after linear addition), it is required to determine the phase difference between every two adjacent time-related bursts of the resulting channel signal. In BVRF, this operation is performed by calculating the values of trigonometric functions according to the data recorded in the memory.

The output binary information, depending on the multiplicity of compaction k, is issued from the output (s) of each channel demodulating device to the input (s) of the subsequent output device (WU) via one, two, or three buses (outputs), i.e. by the number of binary subchannels (corresponding to multiplication factor k) in one channel of the system.

The VU is intended for folding binary information arriving at it via kN outputs into one binary sequence, similar to the transmitted sequence at the output of the encoder of the transmitting complex.

Thus, in the BVRF, a linear incoherent addition of each of N pairs of identical samples of channel signals is performed and the phase difference of the signal sendings across all channels is calculated for all manipulation factors, i.e. in fact, the BVRF performs the function of a block of N channel demodulators, which, unlike the block of channel demodulators of the above system [4], provides demodulation of each result of the linear addition of two samples of the channel signal.

WU here actually performs the function of a parallel-serial converter, as in the above radio communication system [4].

The double diversity reception is an effective means of increasing the noise immunity in radio channels with fading signals [2] and in this case to some extent compensates for the reduction in noise immunity of the reception due to an increase in the data transfer rate.

However, the data transfer rate of this system is insufficient. In addition, in this system, linear incoherent summation of the output levels of channel signal quadrature correlators is performed.

From [6], p. 183, it is known that with diversity reception, linear signal addition provides a smaller gain in noise immunity (in terms of signal-to-noise power ratio) with respect to the optimal coherent addition of these same signals, since in-phase addition of oscillations, the signals are added algebraically, while the noises are added geometrically.

The disadvantage of linear addition is that its implementation puts forward stringent requirements for ensuring equal gain in the diversity branches. A significant difference in the gain in the branches in the limit turns the dual reception into a single one. The allowable spread of gain should provide a spread of no more than 1-2 dB in the entire dynamic range of the linear receiving path, taking into account the influence on the characteristics of the devices destabilizing factors [6], p. 180.

Moreover, a significant drawback of linear addition in this case is the insecurity of each result of adding two samples of any channel signal from exposure to at least one of the radio receiving devices of the diversity diversity branches of additive concentrated (sinusoidal) interference [6], p. 7, within the frequency band occupied by the spectrum of any channel signal.

In this case, the interference voltage will add up (without reducing (suppressing) its level) with the voltages of the channel signal samples, distorting the result of the summation. Upon reaching the level of interference voltage commensurate with the resulting level of the summed samples of the channel signal, the demodulation of the result of the summation by the corresponding channel demodulator can be blocked, i.e. when regenerated binary sequence at the output of the channel demodulator (for k = 1) or in each of the k regenerated binary subsequences corresponding to k outputs of the channel demodulator (for k> 1) begin to appear erroneously received symbols with probability P _err ≈0,5.

As a result, if one of the N channel signals is concentrated by the interference concentrated in the spectrum, the output binary sequence at the output of the parallel-serial converter (to the decoder) may be distorted, for example, for k = 1, each N-th binary symbol. When N = 20, error probability (before the decoder) could reach value P _OJJJ ≤0,05, which may be critical for correcting capability of the selected correction code data communication system and not acceptable to the recipient information (after the decoder).

Under the influence of two or more concentrated interference of a certain level, radio communication obviously becomes unsuitable.

Under the influence of a comparatively broadband spectrum of additive interference, for example, station noise, the occupied frequency band of which covers the frequency band occupied by more than one channel signal, then when a certain level of interference is reached, two or more parallel channels may be affected, which is also obviously not suitable for communication.

The disadvantage is that in the known system [5] it is possible to transmit and receive only signals with phase difference modulation at k = 1, 2, 3, which limits its functionality.

The objectives of which the present invention is directed, a decameter radio communication system with high-speed data transmission, is a 2-fold increase in the data transmission rate and an increase in the noise immunity of binary information reception when exposed to additive interference and fluctuation noise concentrated in the spectrum.

In addition, an additional object of the invention is to expand the functionality of the proposed system by ensuring the transmission and reception of signals with any angular manipulation of the carrier channel signals both in phase (OFT, DOPT, etc.) and frequency (CT, DCT).

The solution of the tasks is achieved by the fact that in a decameter radio communication system with high-speed data transmission containing a transmitting complex containing a serially connected message source, an encoder and a serial-parallel converter, the outputs of which are connected to the corresponding inputs of the block N channel manipulators, the output of which is connected to series-connected a radio transmitting device and a transmitting antenna, as well as a receiving complex containing two receiving antennas, the output of each of a unit connected to the input of the corresponding radio receiver, and a block of N channel demodulators, the outputs of which are connected to the corresponding inputs of a parallel-serial converter, the output of which is connected to a series-connected decoder and a message receiver, and an additional block of N channel manipulators, the inputs of which are connected in series to the transmitting complex connected to the corresponding additional outputs of the serial-parallel converter, additional glad an transmitting device and an additional transmitting antenna, and an additional block of N channel demodulators and 2N coherent signal combining (BCS) blocks are introduced into the receiving complex, one input of each of which is combined with the output of one radio receiver and the other input of each BCS is combined with the output of another radio receiver moreover, the output of each BCS from the first group of N BCS with serial numbers from 1 to N is connected to the corresponding input of the block N channel demodulators, and the output of each BCS from the second N th group of other BCS with serial numbers from 1 to N connected to the corresponding input of the additional channel unit N demodulators whose outputs are connected to corresponding additional inputs of parallel-serial converter.

Each BCS contains two phasing nodes, each of which contains a channel filter connected in series, the input of which is a corresponding BCS input, a normalizing amplifier, a first multiplier, a measuring filter and a second multiplier, the other input of which is connected to the input of the first multiplier, the output of the second multiplier of each phasing node connected to the corresponding input of the adder, the output of which is connected to the input of the filter of the resulting oscillation, whose output, which is the output of the BCS, is connected through Without a normalizing amplifier of the resulting oscillation with a different input of the first multiplier of each phasing unit.

In FIG. 1 is an electrical structural diagram of the proposed decameter radio communication system; in FIG. 2 is a schematic representation of system signals.

A decameter radio communication system with high-speed data transmission comprises a transmitting complex 1, comprising a message source 2 connected in series, an encoder 3 and a serial-parallel converter 4, the outputs of which are connected to the corresponding inputs of the channel manipulator unit N 5 ₁ , the output of which is connected to the radio-transmitting device 6 connected in series ₁ and transmitting antenna 7 ₁ , the additional outputs of the serial-parallel converter 4 are connected to the corresponding inputs the unit block N of channel manipulators 5 ₂ , the output of which is connected to the additional radio transmitting device 6 ₂ and the additional transmitting antenna 7 ₂ in series, as well as the receiving complex 8, which contains two receiving antennas 9 ₁ and 9 ₂ , the output of each of which is connected to the input of the corresponding a radio receiving device 10 ₁ (10 ₂ ), and a block N of channel demodulators 11 ₁ , the outputs of which are connected to the corresponding inputs of the parallel-serial converter 12, the output of which is connected to the series-connected decoder 13 and the recipient of messages 14.

One input of each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _{2-N is} combined with the output of one radio receiver 10 ₁ , and the other input of each BCS is combined with the output of another radio receiver 10 ₂ , moreover, the output of each BCS from the first group of N BCS 15 _1-1 , ..., 15 _1-N with serial numbers from 1 to N is connected to the corresponding input of the block N channel demodulators 11 ₁ , and the output of each BCS from the second group of N other BCS 15 _2-1 , ..., 15 _2-N with serial numbers from 1 to N connected to the corresponding input of an additional block of N channel demodulators 11 ₂ the outputs of which are connected to the corresponding additional inputs of the parallel-serial Converter 12.

Each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N contains two phasing nodes 16 ₁ and 16 ₂ , each of which contains a channel filter 17 connected in series, the input of which is a corresponding input BCS, normalizing amplifier 18, the first multiplier 19, the measuring filter 20 and the second multiplier 21, the other input of which is connected to the input of the first multiplier 19.

The output of the second multiplier 21 of each phasing unit 16 ₁ (16 ₂ ) is connected to the corresponding input of the adder 22, the output of which is connected to the input of the filter of the resulting oscillation 23, the output of which, which is the output of the BCS, is connected through the normalizing amplifier of the resulting oscillation 24 to the other input of the first multiplier 19 each phasing node is 16 ₁ (16 ₂ ).

The decameter radio communication system with high-speed data transmission operates as follows.

In the transmitting complex 1, the transmitted information data stream from the message source 2 enters the encoder 3, the task of which is to increase the noise immunity of data transmission. Coding, as a rule, is accompanied by two effective procedures — scrambling and interleaving. Scrambling converts a digital signal into a quasi-random one in order to obtain a more uniform energy spectrum of the emitted radio signal. Simple interleaving (temporal permutation) of message symbols allows to decorrelate errors in the channel, i.e. convert long-duration error packets into a series of single ones. The last operation significantly increases the coding efficiency [4].

From the output of the encoder 3, a binary sequence with a speed of V _and = 1 / T _and (bit / s), where T _and is the duration of the binary element of the transmitted sequence, is fed to the input of the serial-parallel converter 4, which provides its conversion into 2kN parallel channel sequences with channel repetition rate of binary symbols each equal to V _ch = 1 / 2kNT _and (bit / s). Here N is the number of parallel orthogonal channel signals in each of the two transmitted group signals, k is the multiplicity of multiplexing of each channel signal.

The first kN parallel channel sequences with kN outputs of the serial-parallel converter 4 are supplied to the corresponding inputs of the block N channel manipulators 5 ₁ for generating channel signals of the first group signal, the second kN parallel subsequences from the additional kN outputs of the serial-parallel converter 4 are supplied to the corresponding inputs of the additional block N channel manipulators 5 ₂ for the formation of channel signals of the second group signal.

, а каждый i-ый (i=1,…,N) манипулятор дополнительного блока N канальных манипуляторов 5₂ формирует элементарный сигнал (посылку) на соответствующей частоте

, отличающейся от частоты

на величину

, где

- частотный интервал между соседними канальными частотами [5].Each block of N channel manipulators January ₅ (5 _2), which consists of N similar channel manipulators, e.g., phase or frequency, at k inputs of each link arm at the initial time of each clock period of duration equal to the duration of the channel signal element T _ch = 2kNT _and , parallel and synchronous k symbols of the corresponding k channel sequences are supplied. Moreover, in each clock interval, every i-th (i = 1, ..., N) manipulator of the block N of channel manipulators 5 ₁ generates an elementary signal (package) at the corresponding channel frequency

and each i-th (i = 1, ..., N) manipulator of an additional block of N channel manipulators 5 ₂ generates an elementary signal (package) at the corresponding frequency

different from frequency

by the amount

where

- frequency interval between adjacent channel frequencies [5].

The output signals of the channel manipulators in each block of N channel manipulators 5 ₁ (5 ₂ ) are summed, forming at the output of each block a group N-channel (frequency) group signal. The first group signal generated by the block N of channel manipulators 5 ₁ is radiated using the radio transmitting device 6 ₁ and the transmitting antenna 7 ₁ . The second group signal generated by an additional block N of channel manipulators 5 ₂ is radiated by means of an additional radio transmitting device 6 ₂ and an additional transmitting antenna 7 ₂ .

The antenna 7 ₁ emitting the first group signal is a transmitting antenna emitting an electromagnetic field of horizontal polarization, and the additional antenna 7 ₂ emitting a second group signal is a transmitting antenna emitting an electromagnetic field of vertical polarization [7].

. Здесь же указаны частоты, отстоящие от несущего колебания

излучаемого однополосного группового сигнала (в верхней боковой полосе шириной F=3100 Гц) на 300 и 3400 Гц, ограничивающие полосу пропускания телефонного канала. Эти граничные частоты отстоят от крайних канальных частот на величину ΔF. Частотный интервал между соседними канальными частотами, как указывалось выше, равен

.The first baseband signal emitted in the air in an upper sideband (e.g., radiation class J3E or R3E), represents in each transmission interval of duration T = 2kTN amount _kan N harmonic oscillations with frequencies located schematically depicted in FIG. 2 a. These frequencies, called channel frequencies, are marked with dots on the frequency axis

. Frequencies far from the carrier wave are also shown here.

radiated single-band group signal (in the upper sideband with a width of F = 3100 Hz) at 300 and 3400 Hz, limiting the bandwidth of the telephone channel. These boundary frequencies are separated from the extreme channel frequencies by ΔF. The frequency interval between adjacent channel frequencies, as indicated above, is equal to

.

, где

- частота фазовой манипуляции при скачкообразном изменении фазы круговой частоты манипуляции Ω_м от 0 до π/2 [8], с. 129.In addition, the dashed lines conventionally indicate the frequency spectra of individual channel signals, and within the frequency band occupied by the spectrum of each channel signal, the amplitudes of the main frequency components of the spectrum of the signal, spaced apart from the channel frequency of each channel signal by the value

where

- the frequency of phase manipulation with an abrupt change in the phase of the circular frequency of manipulation Ω _m from 0 to π / 2 [8], p. 129.

однополосного сигнала смещена по частоте относительно

на величину

. На эту же величину смещены и канальные частоты

относительно канальных частот

.In FIG. 2b schematically shows a second single-band group signal emitted by an additional transmitting antenna 7 ₂ , similar in structure to that emitted by an antenna 7 ₁ , except that the carrier frequency

a single-band signal is offset in frequency relative to

by the amount

. Channel frequencies are shifted by the same amount

relative to channel frequencies

.

, которая может составлять порядка нескольких десятков герц [5]), то электромагнитные волны передаваемых групповых сигналов отражаются от одних и тех же областей ионосферы и приходят к местам приема по одним и тем же траекториям [9].It is assumed here that the power and directivity of the radiation of antennas 7 ₁ and 7 _{2 are the} same. Since the two group signals emitted by these antennas actually occupy the frequency band of the same telephone channel with a width of F = 3100 Hz (due to the relatively small value

, which can be on the order of several tens of hertz [5]), then the electromagnetic waves of the transmitted group signals are reflected from the same areas of the ionosphere and arrive at the receiving sites along the same trajectories [9].

It is known that with diversity reception, each “diversity” branch receives its own “sample” of a signal reflected from the ionosphere with a certain implementation of interference [2], in our case, to each diversity branch, consisting of a series-connected receiving antenna 9 ₁ (9 ₂ ) and a radio receiver device (RPU) 10 ₁ (10 ₂ ), receives its own sample of the total signal, consisting of samples of the first and second group signals, as well as its own sample of interference (fluctuation, concentrated in the spectrum, etc.).

Further, for convenience of presentation, the sample of the total signal at the output of the first RPU 10 ₁ (the first diversity branch) will be called the “first sample of the total signal”, and the components of this signal will be called:

- the first sample of the first group signal;

- the first sample of the second group signal;

с порядковым номером i);- the first sample of the i-th channel signal of the first group signal (at the channel frequency

with serial number i);

с порядковым номером i).- the first sample of the i-th channel signal of the second group signal (at the channel frequency

with serial number i).

Another sample of the total signal at the output of the second RPU 10 ₂ (second diversity branch) will be called the "second sample of the total signal", and the components of this signal will be called:

- the second sample of the first group signal;

- the second sample of the second group signal;

с порядковым номером i);- the second sample of the i-th channel signal of the first group signal (at the channel frequency

with serial number i);

с порядковым номером i).- the second sample of the i-th channel signal of the second group signal (at the channel frequency

with serial number i).

In the receiving complex 8, the antenna 9 ₁ is a receiving antenna receiving an electromagnetic field of horizontal polarization, and antenna 9 ₂ is a receiving antenna receiving an electromagnetic field of vertical polarization [7].

The samples of the total signal and additive interference received by antennas 9 ₁ and 9 ₂ from the outputs of the linear reception paths of the corresponding RPUs 10 ₁ and 10 ₂ simultaneously arrive at the corresponding two inputs of each of 2N BCS 15 _1-1 , ..., 15 _1-N and 15 _{2- 1} ..., 15 _2-N .

первого группового сигнала после его приема РПУ 10₁ и 10₂, а во второй группе из N других БКС 15_2-1,…,15_2-N центральные частоты полос пропускания канальных фильтров 17 узлов фазирования 16₁ и 16₂ соответствуют канальным частотам

второго группового сигнала после его приема РПУ 10₁ и 10₂.In each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N channel filters 17 phasing nodes 16 ₁ and 16 _{2 are} identical, and in the first group of N BCS 15 _1-1 , ..., 15 _{1-N the} center frequencies of the passbands of the channel filters 17 phasing units 16 ₁ and 16 ₂ correspond to the channel frequencies

the first group signal after its reception RPU 10 ₁ and 10 ₂ , and in the second group of N other BCS 15 _2-1 , ..., 15 _{2-N the} center frequencies of the passbands of the channel filters 17 phasing nodes 16 ₁ and 16 ₂ correspond to channel frequencies

the second group signal after its reception RPU 10 ₁ and 10 ₂ .

The frequency response of each of the identical channel filters 17 phasing nodes 16 ₁ and 16 _{2 of} each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _{2-N is} consistent with the frequency spectrum of the corresponding channel signal of the first or second received group signals.

Schematically, the frequency response of each of the identical channel filters 17 phasing nodes 16 ₁ and 16 _{2 of} each BCS from the first group of N BCS 15 _1-1 , ..., 15 _1-N can be represented similarly to the frequency spectrum of the corresponding channel signal of the first group signal shown on FIG. 2a, and the frequency response of each of the identical channel filters 17 phasing nodes 16 ₁ and 16 _{2 of} each BCS from the second group of N BCS 15 _2-1 , ..., 15 _2-N is similar to the frequency spectrum of the corresponding channel signal of the second group signal shown in FIG. 2, b.

первого группового сигнала и напряжение первого образца i-ой аддитивной межканальной помехи, создаваемой колебаниями отдельных спектральных составляющих двух соседних канальных сигналов на канальных частотах

и

первого образца второго группового сигнала.The channel filter 17 of the first phasing node 16 _{1 of} any i-th BCS 15 _1-i with serial number i (i = 1, ..., N) from the first group of N BCS 15 _1-1 , ..., 15 _1-N filters ( passes the input of the subsequent normalizing amplifier 18) the voltage of the first sample of the i-th channel signal at the channel frequency

the first group signal and the voltage of the first sample of the i-th additive inter-channel interference created by the oscillations of the individual spectral components of two adjacent channel signals at channel frequencies

and

the first sample of the second group signal.

первого группового сигнала и напряжение второго образца i-ой аддитивной межканальной помехи, создаваемой колебаниями отдельных спектральных составляющих двух соседних канальных сигналов на канальных частотах

и

второго образца второго группового сигнала.Channel filter 17 of the second phasing unit 16 _{2 of} any i-th BCS 15 _1-i with serial number i (i = 1, ..., N) from the first group of N BCS 15 _1-1 , ..., 15 _1-N filters the voltage second sample of the i-th channel signal at the channel frequency

the first group signal and the voltage of the second sample of the i-th additive inter-channel interference created by the oscillations of the individual spectral components of two adjacent channel signals at channel frequencies

and

the second sample of the second group signal.

с порядковым номером i второго группового сигнала и напряжение первого образца i-ой аддитивной межканальной помехи, создаваемой колебаниями отдельных спектральных составляющих двух соседних канальных сигналов на частотах

и

первого образца первого группового сигнала.Channel filter 17 of the first phasing unit 16 _{1 of} any i-th BCS 15 _2-i with serial number i (i = 1, ..., N) from the second group of N BCS 15 _2-1 , ..., 15 _2-N filters the voltage the first sample of the i-th channel signal at the channel frequency

with serial number i of the second group signal and the voltage of the first sample of the i-th additive inter-channel interference created by the oscillations of the individual spectral components of two adjacent channel signals at frequencies

and

the first sample of the first group signal.

с порядковым номером i второго группового сигнала и напряжение второго образца i-ой аддитивной межканальной помехи, создаваемой колебаниями отдельных спектральных составляющих двух соседних канальных сигналов на канальных частотах

и

второго образца первого группового сигнала.Channel filter 17 of the second phasing unit 16 _{2 of} any i-th BCS 15 _2-i with serial number i (i = 1, ... N) from the second group of N BCS 15 _2-1 , ..., 15 _2-N filters the voltage of the second sample of the i-th channel signal at the channel frequency

with serial number i of the second group signal and the voltage of the second sample of the i-th additive inter-channel interference created by the oscillations of the individual spectral components of two adjacent channel signals at channel frequencies

and

second sample of the first group signal.

In addition, the voltage of the corresponding sample concentrated along the channel interference spectrum, as well as the voltage of the fluctuation interference, can fall into the passband of the channel filter 17 of the first and second phasing nodes 16 ₁ and 16 _{2 of} any BCS.

For a more detailed analysis of the operation of the receiving complex 8 of the proposed communication system, we consider the process of changing the direction of polarization of the radio waves emitted by the antennas 7 ₁ and 7 _{2 of the} transmitting complex 1, after reflection from the ionosphere.

From [9], p. 276-279, it is known that a plane-polarized wave incident on the ionosphere (in our case, an electromagnetic field (EMF) with horizontal polarization, radiated by an antenna 7 ₁ or electromagnetic field with vertical polarization, radiated by an antenna 7 ₂ ) is split in general by the Earth’s magnetic field case with two elliptically polarized beams, and the resulting field at the receiving site acquires the character of an elliptically polarized field with a very elongated polarization ellipse, which is clearly illustrated in [9], p. 277 (Fig. 5.24).

Fluctuation changes in the electron concentration along the path of propagation of radio waves manifest themselves in a continuous change in the direction of the large axis of the polarization ellipse.

Moreover, it was experimentally established [9] that if one simultaneously receives such a field on two antennas with different polarizations (in our case, antennas 9 ₁ and 9 ₂ ), then fluctuations in the direction of the plane of polarization will lead to independent fading of the signal at the outputs of the corresponding two RPUs connected to these antennas (in our case, RPU 10 ₁ and 10 ₂ ).

It is noted that an increase in the signal level when receiving a vertical antenna is accompanied by a decrease in the signal level on a horizontal antenna and vice versa, which clearly indicates the existence of continuous fluctuations in the orientation of the major axis of the polarization ellipse [9], p. 227. Such fading is called polarization. A sample recording of signal levels from the outputs of two RPUs that receive on the vertical and horizontal dipoles is given in [9], p. 278 (Fig. 5.25).

In view of the foregoing, let us consider the processes of changes in the voltage levels of samples of each channel signals in the composition of the samples of the total signals at the outputs of the RPU 10 ₁ and 10 ₂ .

The polarization planes of the electromagnetic fields emitted by the antennas 7 ₁ and 7 _{2 of the} transmitting complex 1 are mutually perpendicular. At the receiving site, as mentioned above, each of the two electromagnetic fields becomes elliptically polarized, and the small and large axes of the two polarization ellipses will also be mutually orthogonal, since the channel frequencies of the first and second emitted group signals differ insignificantly (as indicated above, by a few tens of hertz) )

первого группового сигнала, примет направление, близкое к горизонтали (при этом большая ось эллипса поляризации поля i-го канального сигнала на частоте

второго группового сигнала, примет направление, близкое к вертикали), то электродвижущая сила (эдс), наводимая в этой антенне полем i-го канального сигнала на частоте

первого группового сигнала будет максимальной, а эдс, наводимое полем i-го канального сигнала на частоте

второго группового сигнала, будет минимальной (около нуля). Соответственно на выходе РПУ 10₁ уровень напряжения первого образца i-го канального сигнала на частоте

первого группового сигнала будет максимальным, а уровень первого образца i-го канального сигнала на частоте

второго группового сигнала будет минимальной (около нуля).When receiving these signals on the antenna 9 ₁ (receiving EMF of horizontal polarization), when in the process of continuous changes the large axis of the polarization ellipse ([9], p. 227), for example, the field of the ith channel signal at a frequency

the first group signal, will take a direction close to the horizontal (in this case, the large axis of the ellipse of polarization of the field of the i-th channel signal at a frequency

of the second group signal, will take a direction close to the vertical), then the electromotive force (emf) induced in this antenna by the field of the i-th channel signal at a frequency

the first group signal will be maximum, and the emf induced by the field of the i-th channel signal at a frequency

the second group signal will be minimal (near zero). Accordingly, at the output of the RPU 10 ₁ the voltage level of the first sample of the i-th channel signal at a frequency

the first group signal will be maximum, and the level of the first sample of the i-th channel signal at a frequency

the second group signal will be minimal (near zero).

первого группового сигнала будет минимальным (около нуля), а уровень напряжения второго образца i-го канального сигнала на частоте

второго группового сигнала будет максимальным.In this case, in the process of receiving signals to the antenna 9 ₂ (receiving the electromagnetic field of vertical polarization), the opposite picture will be observed at the output of the RPU 10 ₂ : the voltage level of the second sample of the i-th channel signal at a frequency

the first group signal will be minimal (near zero), and the voltage level of the second sample of the i-th channel signal at a frequency

the second group signal will be maximum.

Thus, due to polarization fading, in the proposed communication system, the levels of the received first samples of channel signals of the first and second group signals at the output of the RPU 10 ₁ and the levels of the received second samples of the same channel signals of the first and second group signals at the output of the RPU 10 ₂ in the following way:

первого группового сигнала уменьшается уровень напряжения (до минимального значения - около нуля) первого образца i-го канального сигнала на частоте

второго группового сигнала, одновременно уменьшается уровень напряжения (до минимального значения - около нуля) второго образца i-го канального сигнала на частоте

первого группового сигнала и возрастает уровень напряжения (до максимального значения) второго образца i-го канального сигнала на частоте

второго группового сигнала.- with increasing voltage level of the first sample of the i-th channel signal at a frequency

of the first group signal, the voltage level decreases (to a minimum value of about zero) of the first sample of the i-th channel signal at a frequency

of the second group signal, the voltage level decreases simultaneously (to the minimum value - about zero) of the second sample of the i-th channel signal at a frequency

the first group signal and the voltage level increases (up to the maximum value) of the second sample of the i-th channel signal at a frequency

second group signal.

These changes in the levels of channel signals can be represented in another form:

второго группового сигнала уменьшается уровень напряжения (до минимального значения - около нуля) первого образца i-го канального сигнала на частоте

первого группового сигнала, одновременно уменьшается уровень напряжения (до минимального значения - около нуля) второго образца 16₁ и 16₂ канального сигнала на частоте

второго группового сигнала и увеличивается уровень напряжения (до максимального значения) второго образца i-го канального сигнала на частоте

первого группового сигнала.- as the voltage level increases (up to the maximum value) of the first sample of the i-th channel signal at a frequency

of the second group signal, the voltage level decreases (to a minimum value of about zero) of the first sample of the i-th channel signal at a frequency

of the first group signal, the voltage level decreases simultaneously (to the minimum value - about zero) of the second sample 16 ₁ and 16 ₂ channel signal at a frequency

the second group signal and the voltage level increases (up to the maximum value) of the second sample of the i-th channel signal at a frequency

first group signal.

первого группового сигнала, а на фиг. 3,б - модель характера замираний амплитуд U'_1Ci и U'_2Ci напряжений образцов i-го канального сигнала на частоте

второго группового сигнала (также по упрощенным линейным законам).To illustrate the nature of the polarization fading of the voltage samples of the i-th channel signal (as part of the samples of the total signal) received RPU 10 ₁ and 10 ₂ and recorded at the outputs of the channel filters 17 phasing nodes 16 ₁ and 16 _{2 of the} i-th BCS 15 _1-ii from the composition of the first group of N BCS and by analogy with the experimental data given in [9], p. 278 (Fig. 5.25), in FIG. _3a , a model of the character of fading of the amplitudes U _1Ci and U _2Ci (according to simplified linear laws) of the voltage of samples of the ith channel signal at a frequency is presented

the first group signal, and in FIG. _3b is a model of the character of fading of the amplitudes U ' _1Ci and U' _{2Ci of the} voltage of samples of the i-th channel signal at a frequency

second group signal (also according to simplified linear laws).

Changes in the amplitudes of the voltages are presented within a certain period of time, which can be called a “half-period” of fading T _P3 (0≤t≤t ₃ ) with a duration equal to 4 conditional gradations of time.

и

первого образца второго группового сигнала при максимальном значении их амплитуд, здесь принято равным 10% от максимального значения амплитуды U_1Cimax (фиг. 3,а), что практически может несколько превышать истинное максимальное значение величины U_1МПimах, определяемое экспертным путем или экспериментально.The voltage amplitudes of the samples of channel signals are given in the form of normalized (relative) quantities - U (t) / U _1max . It also shows the nature of the fading of the average values of the stress amplitudes of the inter-channel interference samples in relative units. The maximum average value of the voltage amplitude of the first sample of the i-th inter-channel interference U _1MPimax at the output of the channel filter 17 of the first phasing unit 16 _{1 of the} i-th BCS 15 _1-i created by the voltages of two adjacent channel signals at frequencies

and

the first sample of the second group signal at the maximum value of their amplitudes, here it is assumed to be 10% of the maximum value of the amplitude U _1Cimax (Fig. 3, a), which can slightly exceed the true maximum value of the value U _1MPimax , determined by experts or experimentally.

и

второго образца второго группового сигнала при максимальном значении их амплитуд, принято также равным 10% от максимального значения амплитуды U_2Cimах (фиг. 3,а).The maximum average voltage amplitude of the second sample of the i-th inter-channel interference U _2MPimax at the output of the channel filter 17 of the second phasing unit 16 _{2 of the} i-BCS 15 _1-i created by the voltages of two adjacent channel signals at frequencies

and

the second sample of the second group signal at the maximum value of their amplitudes, is also taken equal to 10% of the maximum value of the amplitude U _2Cimax (Fig. 3, a).

и

первого группового сигнала.Similarly in FIG. _3b shows the nature of the change in the average values of the amplitudes U ' _1MPi and U' _{2MPi of the} voltages of the first and second samples of the i-th inter-channel interference at the outputs of the channel filters 17 phasing nodes 16 ₁ and 16 _{2 of the} i-th BCS 15 _2-i from the second group from N BCS and created by voltage samples of two adjacent channel signals at frequencies

and

first group signal.

Consider the process of isolating the voltage of channel signals, for example, the first group signal from the voltages of the samples of the total signal at the outputs of the RPU 10 ₁ and 10 ₂ using BCS from the first group of N BCS 15 _1-1 , ..., 15 _1-N .

с помощью соответствующего i-го БКС 15_1-i из состава первой группы из N БКС 15_1-1,…,15_1-N.Since the channel signals of each group signal are orthogonal, i.e. independent of each other, it is enough to consider the process of selecting one of the N channel signals of the first group signal using the example of selecting, for example, the ith channel signal with a channel frequency

using the corresponding i-th BCS 15 _1-i from the first group of N BCS 15 _1-1 , ..., 15 _1-N .

Let us analyze the operation of any i-th BCS 15 _1-i from the composition of the first group of N BCS 15 _1-1 , ..., 15 _1-N , which provides selection and optimal coherent addition of voltages of two samples of the i-th channel signal of the first group signal. We will consider the operation of the BCS when receiving at any stationarity interval of duration Δt selected in accordance with FIG. 3, a, b within the duration of the conditional average “half-period” of fading samples of the channel signal T _P3 (0≤t≤t ₃ ).

We assume that the voltage amplitudes of the samples of channel signals within each stationarity interval of duration Δt do not change. Moreover, the duration of each selected interval Δt should be greater than the time constant of the measuring filter 20 and more than the time constant of the AGC circuit of the normalizing amplifiers 18 BCS 15 _1-i , but much less than the value of T _P3 .

Under such conditions, it is natural to assume that at the end of any arbitrarily chosen stationary interval of duration Δt, all transients in each i-th BCS 15 _{1-i are} completed with the voltage levels of the channel signal samples and additive noise at the outputs of the channel filters 17 corresponding to this interval Δt.

Let from the outputs of the linear paths of the RPU 10 ₁ and 10 ₂ to the first and second inputs of the BCS 15 _1-i come within a stationary interval of duration Δt, ending, for example, at a conditional time t = 1 (Fig. 3, a, b), samples of the first group signal in the form of:

- at the first entrance -

- to the second entrance -

Here U _1Gy (t) and U _2Gy (t) are the voltages of the first and second samples of the first group signal, respectively;

U _1Сi (t) is the voltage of the first sample of the i-th channel signal;

U _2Ci (t) is the voltage of the second sample of the i-th channel signal;

U _1Ci and ϕ _1Ci - the amplitude and phase of the voltage of the first sample of the i-th channel signal;

U _2Ci and ϕ _2Ci - the amplitude and phase of the voltage of the second sample of the i-th channel signal;

;ω _Ci is the angular channel (carrier) voltage frequency of the i-th channel signal

;

θ _Ci (t) is a function that determines the type of angular manipulation of the voltage of the i-th channel signal;

N is the number of channel signals in the received group signal.

Simultaneously with the outputs of the linear paths of receiving the RPU 10 ₁ and 10 _2, the first and second inputs of the BCS 15 _1-i receive voltage samples of the second group signal. By analogy with expressions (1) and (2), samples of the second group signal can be represented as:

- at the first entrance -

- at the second entrance -

Here U ' _1Gy (t) and U' _2Gy (t) are the voltages of the first and second samples of the second group signal, respectively;

U ' _1Ci (t) is the voltage of the first sample of the i-th channel signal;

U ' _2Ci (t) is the voltage of the second sample of the i-th channel signal;

U ' _1Ci and ϕ' _1Ci - the amplitude and phase of the voltage of the first sample of the i-th channel signal;

U ' _2Ci and ϕ' _2Ci - the amplitude and phase of the voltage of the second sample of the i-th channel signal;

;ω ' _Ci is the angular channel (carrier) voltage frequency of the i-th channel signal

;

θ ' _Ci (t) is a function that determines the type of angular manipulation of the i-th channel signal;

N is the number of channel signals in the second group signal.

To simplify the analysis of the operation of BCS 15 _1-i, we assume that the transmission coefficient of any of the filters (17, 20, 24), as well as the adder 22 in each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N is equal to one. In addition, due to the fact that the structure of each BCS is a closed loop system of self-regulation with feedback, signal delays or changes in their initial phases when passing through these BCS filters will not be taken into account.

We also assume that at the output of the channel filter 17 of each phasing unit 16 ₁ (16 ₂ ), the maximum voltage amplitude of the sample of the i-th channel signal is much higher than the maximum average voltage amplitude of the corresponding inter-channel interference sample, i.e. U _1Cimax >> U _1MPimax and U _2Cimax >> U _2MPimax . Such amplitude ratios make it possible not to take into account the negative effect of the voltage of the inter-channel interference samples created by the second group signal, which can occur in the following cases:

1. For example, when using OFT or DOPT signals for data transmission, when in each channel signal the average number of abrupt changes in the phase of the carrier wave occurring per unit time is much less than the maximum possible number of abrupt changes in phase determined by the speed of channel signal manipulation. In this case, when the code combinations transmitted by each channel signal, consisting of a different number of symbols of the same type (in the form of long “taps” and long “depressions”), alternate at a fairly low average speed, the spectral components of the channel signal are grouped near the carrier wave of the channel signal .

(фиг. 2,а,б), и негативное действие напряжений образцов межканальной помехи можно не учитывать (даже при отсутствии поляризационных замираний, обеспечивающих существенное уменьшение негативного действия напряжений образцов межканальной помехи за счет реализации весового когерентного сложения напряжений образцов канального сигнала, при котором происходит подавление напряжения одного из образцов этой помехи в том узле фазирования 16₁ (16₂), на выходе канального фильтра 17 которого напряжение образца межканальной помехи превышает уровень напряжения соответствующего образца канального сигнала).The bandwidth occupied by the spectrum of such a channel signal is less than the maximum possible value of the bandwidth occupied by the channel signal, equal to

(Fig. 2, a, b), and the negative effect of stresses of inter-channel interference samples can be ignored (even in the absence of polarization fading, which significantly reduce the negative effect of stresses of inter-channel interference samples due to the implementation of weighted coherent summation of the voltage of the channel signal samples at which suppression of the voltage of one of the samples of this interference in that phasing unit 16 ₁ (16 ₂ ), at the output of the channel filter 17 of which the voltage of the inter-channel interference sample exceeds the level voltage of the corresponding channel signal sample).

.2. With an increase in the multiplicity k of the channel signal of the transmitted group signal, for example, when using phase-difference manipulation, which allows to reduce the speed of channel signal manipulation to a certain value at which the channel signal bandwidth becomes significantly less than

.

In view of the foregoing, samples of the i-th channel signal of the first group signal filtered at the inputs of the normalizing amplifiers 18 of the i-th BCS 15 _1-i, filtered by channel filters 17 of the phasing nodes 16 ₁ and 16 _2, can be represented as follows:

- for the first node 16 ₁ -

- for the second node 16 ₂ -

It should be noted that in the proposed communication system, RPUs 10 ₁ and 10 ₂ should operate in the off mode of their own automatic gain control (AGC) system, since the RPU AGC can only adjust the level of a group signal received in the corresponding wide band, and not each channel signal with the reception band in N of the smaller reception band of the group signal.

The AGC system of each normalizing amplifier 18 and 24 of any BCS can be characterized by the coefficient of regulation of the AGC system. The AGC control coefficient shows how many times the range of the signal at the output of the normalizing amplifier is less than at its input [6]:

where U _{IN MIN} and U _{OUT MIN} are the minimum input and minimum output voltages, which are limited by the value of the real sensitivity of the normalizing amplifier 18 of the phasing unit 16 ₁ (16 ₂ ), and U _{IN MAX} and U _{OUT MAX} are limited by the maximum value of input oscillations at which the level of combination components at the output of the normalizing amplifier 18 does not exceed the permissible.

For each of the identical normalizing amplifiers 18 and 24 of any BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N, we consider acceptable, for example, a change in the signal at the input of the normalizing filtered by the corresponding channel filter 17 amplifier 18 to 100 dB when the signal at its output changes by no more than 3 dB. AGC systems with such parameters are implemented in modern RPUs [10].

At the output of the normalizing amplifiers 18 of each phasing unit 16 ₁ and 16 _{2, the} filtered samples of the i-th channel signal are equalized in level and fed to the inputs of the first multipliers 19, the other inputs of which come from the normalizing amplifier of the resulting oscillation 24, the resulting signal:

where U _Pi , ω _Pi ϕ _Pi are the amplitude, angular frequency, and phase of the resulting signal, respectively.

The output product of the first multiplier 19 of the first phasing unit 16 ₁ , to one input of which the filtered and normalized voltage of the first sample of the i-th channel signal is supplied, and to its other input, the resulting signal, can be represented as:

where K ₁ is the transmission coefficient of the normalizing amplifier 18 of the first phasing unit 16 ₁ , at which normalization of the first sample of the input channel signal with amplitude U _{1Ci is provided} .

The first term in braces is easily eliminated by the measuring filter 20 of the first phasing unit 16 ₁ , because its spectrum is much higher than the spectrum of the second term.

The second term in curly brackets of expression (7) is harmonic oscillation (without manipulation) at the difference circular frequency ω _Фi = ω _Ci -ω _Pi , which coincides with the central frequency of the measuring filter 20 phasing nodes 16 ₁ and 16 ₂ . Since this oscillation is directly proportional to the amplitude of the received signal U _1Ci , then in the absence of interference at the inputs of the i-th BCS 15 _{1-i considered} , at the output of the measuring filter 20, the amplitude of this oscillation will be maximum and correspond to the maximum “weight” of the voltage of the received first sample channel signal in the normalized oscillation at the output of the normalizing amplifier 18.

The output voltage of the measuring filter 20 of the first phasing unit 16 ₁ , taking into account the foregoing, can be represented as:

For a more accurate assessment in the phasing unit 16 ₁ (16 ₂ ) of the level or "weight" of the channel signal sample in the normalized mixture of signal and noise at the output of the normalizing amplifier 18, the passband of the measuring filter 20 of each phasing unit, on the one hand, should be extremely small and, on the other hand, it is necessary that this band provides the ability to "track" the signal level during its fading and changes in the frequency of the channel signal during its reception. In the practical implementation of the communication system, this band can be selected on the order of (20-25) Hz.

Similarly to (8), one can imagine the output voltage of the measuring filter 20 of the second phasing unit 16 ₂ , which in this case also corresponds to the maximum “weight” of the voltage of the received second sample of the ith channel signal:

where K ₂ is the value of the transfer coefficient of the normalizing amplifier 18 of the second phasing unit 16 ₂ , which ensures normalization of the second sample of the input channel signal.

The output of the second multiplier 21 of the first phasing site 16 ₁ will be:

Similarly, the output of the second multiplier 21 of the second phasing unit 16 _{2 is} recorded:

The first terms in braces (10) and (11) are eliminated during further filtering of the output product of the adder 22 by the filter of the resulting oscillation 23 and can be ignored. Therefore, the voltage of the first sample of the channel signal at the output of the filter of the resulting oscillation 23, which must be taken into account when summing in the adder 22 (at its first input), can be represented as:

Similarly, you can imagine the voltage of the second sample of the channel signal at the output of the filter of the resulting oscillation 23, which must be taken into account when summing in the adder 22 (at its second input):

In this case, the output voltage of the filter of the resulting oscillation 23 is written in the form:

Given that in the phasing nodes 16 ₁ and 16 _2, the voltage amplitudes of the corresponding samples of the i-th channel signal are equalized by each of the normalizing amplifiers 18 to a certain normalized value U _CH , the maximum range of which does not exceed 3 dB when the input voltage amplitude changes to 100 dB , then the magnitude of the amplitude of the output normalized oscillation U _CH with a limited, for example, up to 40 dB range of changes in the amplitude of the input oscillations, can be considered constant:

In view of (15), expression (14) can be represented as:

Using the normalizing amplifier 24, the resulting voltage U _{P CiФ} (t) is normalized by level, i.e. reduced to the form (6).

с порядковым номером i второго группового сигнала с помощью i-го БКС 15_2-i из состава второй группы N других БКС 15_2-1,…,15_2-N.Similarly, the selection and "weight" addition of two samples of each channel signal at a frequency

with serial number i of the second group signal using the i-th BCS 15 _2-i from the second group N of other BCS 15 _2-1 , ..., 15 _2-N .

In this case, the 17 phasing nodes 16 ₁ and 16 _{2 of the} i-BCS 15 filtered by channel filters _{2 2-i} samples of the i-th channel signal of the second group signal will have the form:

- for the first node 16 ₁ -

- for the second node 16 ₂ -

The demodulation of each i-th resulting channel signal of the first group signal from the output of the i-th BCS15 _1-i (from the first group of N BCS) is performed by the corresponding channel demodulator of the block N channel demodulators 11 ₁ using one of the known methods [2], and demodulation each i-th resulting channel signal of the second group signal from the output of the i-th BCS 15 _2-i from the second group of N BCS is produced by the corresponding channel demodulator of an additional block of N channel demodulators 11 _{2 in a} similar way m

As a result, the first kN binary channel sequences are formed at the outputs of the block of N channel demodulators 11 ₁ , and the second kN binary channel sequences are formed at the outputs of the additional block of N channel demodulators 11 ₂ .

Using a parallel-serial converter 12, all 2kN channel sequences are converted into one binary sequence similar to that transmitted from the output of encoder 3, which, after decoding in decoder 13, is then transmitted to the information receiver 14.

Thus, in the proposed communication system, the voltage amplitudes of samples of any i-th channel signal, both the first and second group signals in the corresponding i-th BCS 15 _1-i (15 _2-i ) are squared and summed in phase at the output of this BCS or at the input of the corresponding i-th demodulator of the block of N channel demodulators 11 ₁ (11 ₂ ). In this case, the data transfer rate increases by 2 times in relation to the prototype [5].

Based on the above analysis method of the operation of any BCS, it can be proved that the amplitudes of the received RPU 10 ₁ and 10 ₂ voltages of two samples of the additive i-th concentrated in the interference spectrum, which is (when broadcasting) harmonic oscillation at a fixed frequency falling into the frequency band occupied by the i-th channel signal, for example, the first group signal, in the adder 22 of the i-th BCS 15 _{1-i are} added geometrically.

Moreover, the larger the voltage amplitude of the sample concentrated by the interference spectrum at the output of the channel filter 17, for example, the first sample U _1Pi (the first phasing unit 16 ₁ ) with respect to the voltage amplitude of the first sample of the channel signal U _1Ci at the output of this filter, the smaller weight ”the converted voltage of this sample is concentrated on the interference spectrum from the output of the second multiplier 21 to the first input of the adder 22.

As a result, the proposed communication system provides an increase in the signal-to-noise voltage ratio at the output of any i-th BCS 15 _1-i (at the input of the i-th demodulator of the block N channel demodulators 11 ₁ ) and at the output of any i-th BCS 15 _{2- i} (at the input of the i-th demodulator of an additional block of N channel demodulators 11 ₂ ) with respect to the value of the same signal to noise ratio at the input of the i-th channel demodulator of the prototype [5] under the same communication conditions.

To prove the advantages of the proposed communication system - to achieve higher noise immunity of reception, both the first and second group signals, we will conduct a comparative assessment of its noise immunity with respect to noise immunity of the known communication system [5] - prototype.

, занимаемой спектром передаваемого i-го канального сигнала первого группового сигнала (фиг. 2,а).To do this, we analyze the work of the i-th BCS 15 _1-i from the composition of the first group of N BCS 15 _1-1 , ..., 15 _1-N when receiving RPU 10 ₁ and RPU 10 ₂ voltages of the corresponding samples of the total signal and samples of the additive i-th concentrated along the spectrum of channel interference [6], p. 7 radiated into the air by an external (interfering) radio transmitting device and representing a sinusoidal oscillation at a fixed frequency within the frequency band

occupied by the spectrum of the transmitted i-th channel signal of the first group signal (Fig. 2, a).

канальных фильтров 17 i-го БКС 15_1-i).Sinusoidal oscillation at a fixed frequency within the frequency band of the signal reception, simulating a real channel-centered channel interference, is often used to check the noise immunity of the RPU as a whole and its component, the demodulator. This kind of i-th interference concentrated along the spectrum will be called the i-th in-band channel interference (falling into the passband

channel filters 17 i-th BCS 15 _1-i ).

The voltages of the first and second samples of such an i-th in-band channel interference can act on the inputs of the normalizing amplifiers 18 of the i-th BCS 15 _1-i at any fixed frequency within the passband of the channel filters 17.

The voltage of the first sample of the i-th in-band channel interference at the input of the normalizing amplifier 18 of the first phasing unit 16 _{1 of the} i-th BCS 15 _1-i , operating within the stationary interval of duration Δt, can be represented as:

where U _1Пi and ϕ _1Пi are the amplitude and phase of the first interference sample, which are constant values within the interval Δt, ω _Пi is the angular frequency of the interference.

Similarly, you can imagine the voltage of the second sample of the i-th in-band channel interference at the input of the normalizing amplifier 18 of the second phasing unit 16 _{2 of the} i-th BCS 15 _1-i :

where U _2Pi and ϕ _2Pi are the amplitude and phase of the voltage of the second interference sample.

When considering various relationships of voltage amplitudes of samples of channel signals of the first and second group signals at the inputs of channel filters 17 phasing nodes 16 ₁ and 16 _{2, the} magnitudes of the voltage amplitudes U _1Pi and U _{2Pi of} samples of in-band channel noise at the outputs of these filters will be chosen obviously significantly larger _than real average values voltage amplitudes of inter-channel interference samples U _1MPi and U _2MPi i.e. U _1Pi >> U _1MPi and U _2Pi >> U _2MPi .

Гц, в зависимости от средней скорости скачкообразных изменений манипулируемого параметра канального сигнала при угловой манипуляции, определяемого функцией θ_Ci(t) из (1), например, фазы - при ОФТ, ДОФТ или частоты - при ЧТ, ДЧТ.It should be noted here that in the process of receiving the total signal over any stationarity interval of duration Δt, the values of the average values of the voltage amplitudes of the inter-channel interference samples at the outputs of the channel filters 17 of the phasing nodes 16 ₁ and 16 ₂ depend both on the levels of the corresponding samples of channel signals (in this case, the second group signal), and also on the spectrum width of each of these channel signals, which, as noted above, can occupy a frequency band from units of hertz to a maximum value -

Hz, depending on the average rate of step-wise changes in the manipulated parameter of the channel signal during angular manipulation, determined by the function θ _Ci (t) from (1), for example, the phase during the OFT, DOPT, or the frequency during BH, CT.

The real average values of the voltage amplitudes of the inter-channel interference samples at the output of each channel filter 17 can only be determined experimentally or expertly, since such interference in the analytical form has not been determined in the technical literature.

Analysis of the operation of the i-th BCS 15 _1-i under the action of the voltage of samples of the i-th in-band channel interference, the amplitudes of which significantly exceed the average voltage amplitudes of the corresponding samples of inter-channel interference, will allow us to evaluate the signal-to-noise voltage ratio at the output of each i-th BCS 15 _1-i (which determines the noise immunity of the reception of each i-th channel signal) under more severe communication conditions than under the action of real voltage values of inter-channel interference samples. Accordingly, at such values of U _1Pi and U _{2Pi, the} influence of inter-channel interference on the reception noise immunity can be ignored, and reception noise immunity can only be considered from the influence of the external i-th in-band channel interference.

Thus, at the output of the channel filter 17 of the first phasing unit 16 _{1 of the} i-th BCS 15 _1-i , two oscillations will act - the first sample of the i-th channel signal U _1Ci (t) and the first sample of the i-th in-band channel interference U _1Pi ( t) defined by expressions (3) and (19), respectively:

At the output of the channel filter 17 of the second phasing unit 16 ₂ BCS 15 _1-i , two oscillations will also act - the second sample of the i-th channel signal U _2Ci (t) and the second sample of the i-th in-band channel interference U _2Pi (t), defined by the expressions (4) and (20), respectively:

An analytical expression describing with high accuracy the sum of two oscillations of type (21) and (22) was not found in the technical literature. However, taking into account that in expressions (21) and (22), the detuning | Δω | = | ω _{Ci -ω Пi} | is a small quantity compared to the average circular frequency (ω _Ci + ω _Пi ) / 2, and also, with a certain probability, situations are possible when the function θ _Ci (t), which determines the type of angular manipulation of the ith channel signal, changes stepwise its value within the stationary interval of duration Δt is relatively rare, or remains constant (which can occur when the first group signal is transmitted via the i-th frequency channel, for example, in the OFT or CHT mode, the signal is “pressed” as a series of binary symbols “1 ", Ne edavaemyh within Δt interval), then each resulting oscillation can be considered narrowband process [11].

To assess the noise immunity of the reception of the i-th channel signal, which is characterized by the value of the signal-to-noise voltage ratio at the output of the i-BCS 15 _1-i for various voltage ratios of the samples of the channel signal and in-band interference, we consider the process of receiving the total signal and in-band channel interference within each of those indicated in FIG. 3, a, b time intervals.

1. Reception on any stationarity interval of duration Δt, selected within the time interval T1 (0 <t <t ₁ ).

Within this interval, at the output of the channel filter 17 of the first phasing unit 16 _{1 of the} i-th BCS 15 _1-i, the voltage amplitude of the first sample of in-band channel interference U _1Pi exceeds the voltage amplitude of the first sample of the i-th channel signal U _1Ci and significantly exceeds the average voltage amplitude of the first inter-channel interference sample U _1MPi , i.e. U _1Ci <U _1Pi >> U _1MPi .

In this case, the voltage amplitude of the second sample of the i-th channel signal U _2Ci within the time interval T1 (0 <t <t ₁ ) in accordance with FIG. _3a exceeds the voltage amplitude of the second sample of in-band channel interference U _2Pi , which, in turn, significantly exceeds the average voltage amplitude of the second sample of inter-channel interference U _2MPi , i.e. U _2Ci > U _2Pi >> U _2MPi .

The envelope U _1T1 (t) of the resulting oscillation (21), which is a narrow-band process in the time interval T1, can be represented with a certain degree of accuracy according to [11], p. 119, in the form:

where M _1iT1 = U _1Ci / U _1Пi is the i-th channel ratio of signal amplitudes of the signal to noise at the output of the channel filter 17 of the first phasing unit 16 ₁ on any stationary interval of duration Δt ending at the considered time t and chosen within the first (in in accordance with Fig. 3, a, b) of the time interval T1, for M _1iT1 <1 and ω _{Ci −ω 1Pi} = Δω> 0.

In this case, the resulting oscillation (21) at the input of the normalizing amplifier 18 of the first phasing unit 16 ₁ taking into account (23) will have the form [11]:

where θ (t) is the function that describes the phase changes of the resulting oscillation with the center frequency of the in-band channel interference ω _Pi at U _1Pi > U _1Ci [11].

For the second phasing unit 16 _{2, the} envelope U _2iT1 (t) of the narrow-band process (22) at the input of the normalizing amplifier 18 of the second phasing unit 16 _2, similarly to (23), can be represented as:

where R _2iT1 = U _2Pi / U _Ci is the i-oe channel ratio of the amplitudes of the interference voltage / signal at the output of the channel filter 17 of the second phasing unit 16 ₂ on the stationary interval of duration Δt for R _2iT1 <1.

The resulting oscillation (22) at the input of the normalizing amplifier 18 of the second phasing unit 16 ₂ taking into account (25) will have the form [11]:

where θ '(t) is the function that describes the phase changes of the resulting oscillation with the central frequency of the signal ω _Ci for U _2Ci > U _2Пi [11].

The time constant of the AGC system of normalizing amplifiers 18 should be greater than the period of change of the envelope of the resulting oscillations (beats) with a frequency Δω and, at the same time, less than the interval Δt.

From (24) and (26) it follows that, within a specific interval of stationarity of duration Δt, the average values of the amplitudes of the resulting oscillations at the inputs of the normalizing amplifiers 18 of the phasing nodes 16 ₁ and 16 ₂ taking into account the maximum and minimum values of the variable quantity cosΔωt will have the form:

Define in expression (27) the value of the second factor of the numerator (in brackets):

It is taken into account that when extracting the square root, each of the two terms must be greater than zero for M _1iT1 <1.

Accordingly, in the expression (28), the value of the second factor of the numerator will also be equal to:

Thus, expressions (27) and (28) take the following form:

Given that the AGC of the normalizing amplifier 18 of each of the phasing nodes 16 ₁ and 16 ₂ responds to the average value of the amplitude of the input oscillation, the average value of the amplitude of the normalized resulting oscillation at the output of the normalizing amplifier 18 of the first phasing node 16 ₁ taking into account (15) and (27 ') will be equal to:

Here K ₁ is the value of the transfer coefficient of the normalizing amplifier 18 of the first phasing unit 16 ₁ , at which its output level is normalized by the AGC system to a known value U _CH determined by expression (15).

From (29) we determine the value of K ₁ on a specific interval of stationarity Δt within the first (in accordance with Fig. 3, a, b) time interval T1, on which the value M _1iT1 <1 takes the corresponding value:

Using the principle of superposition on a selected stationarity interval of duration Δt, the output product of the first multiplier 19 of the first phasing unit 16 ₁ in this case will be:

Here, the first term characterizing the voltage of the first sample of the channel signal at the output of the first multiplier 19 of the first phasing node 16 ₁ can be represented similarly to expression (7):

In this case, the measuring filter 20 of the first phasing unit 16 ₁ will isolate from the sum of two oscillations only harmonic oscillation with the central frequency of this filter ω _φi = ω _Ci -ω _Pi , which, by analogy with (8) and taking into account (30), can be represented as :

The second term in (31), which characterizes the voltage of the first sample of in-band channel interference at the output of the first multiplier 19 of the first phasing unit 16 ₁ , will have the form:

From consideration (34) it follows that in the first multiplier 19, the harmonic oscillation of the interference U _1Пi (t) is converted into two oscillations with angular manipulation: at the upper carrier frequency ω _в = ω _Пi + ω _Pi (the first term in parentheses) and at the bottom carrier frequency ω _n = ω _Pi -ω _Pi (the second term in parentheses).

The spectral components of the first oscillation with angular manipulation at the carrier frequency ω _are much higher than the center frequency ω _{Фi of the} measuring filter 20 and are easily eliminated by them, and individual spectral components of the second phase-manipulated signal at the carrier frequency ω _n can fall into the passband of the measuring filter 20.

However, taking into account that the passband of the measuring filter 20 is much smaller than the passband of the channel filter 17 corresponding to the spectrum of the received signal, the response of the narrow-band measuring filter 20 to the above individual spectral components of interference with angular manipulation can be ignored due to the low levels of these components.

Thus, the output voltage of the measuring filter 20 when receiving a mixture of signal and in-band channel interference with a certain degree of accuracy will correspond to expression (33), i.e. the output voltage of this filter will determine the "weight" of the channel signal in the mixture of signal and interference:

Using the principle of superposition, the output of the second multiplier 21 of the first phasing unit 16 ₁ , taking into account (35), will have the form:

The first term in (36) determines the voltage of the first channel signal sample at the output of the second multiplier 21 of the first phasing branch 16 ₁ . By analogy with (10) and taking into account (30) and (35), this term can be written in the form:

Considering that the first term in (37) is eliminated during further filtering, we will take into account, as before, only the second term at the first input of adder 22, which characterizes the voltage of the first sample of the channel signal:

with average amplitude

The second term in (36) determines the voltage of the in-band channel interference at the first input of the adder 22. Taking into account (19), (30) and (35), this term will have the form:

By analogy with (37) in (39), we will take into account only the second term at the first input of adder 22, which characterizes the voltage of the first sample of in-band channel interference:

where ω ' _Пi = ω _Ci- ω _{Pi -ω Пi} , ψ ₁ = ϕ _1Ci -ϕ _Pi -ϕ _1Пi .

We perform similar operations to determine the resulting oscillation at the output of the second multiplier of the second phasing unit 16 ₂ , which is fed to the second input of the adder 22.

In this case, on the selected stationarity interval of duration Δt, the average value of the amplitude of the normalized resulting oscillation at the output of the normalizing amplifier 18 of the second phasing unit 16 ₂ taking into account (15) and (28 ') will be equal to:

Here, K ₂ is the value of the transfer coefficient of the normalizing amplifier 18 of the second phasing unit 16 ₂ , at which its average output level is normalized by the AGC system to a known value of U _CH (15).

From (41) we determine the value of K ₂ , which is constant within the stationarity interval Δt ending at the considered time t:

Using the principle of superposition, the output product of the first multiplier 19 of the second phasing unit 16 ₂ in this case will be:

Here, the first term characterizing the voltage of the second sample of the i-th channel signal of the first group signal at the output of the first multiplier 19 of the second phasing node 16 ₂ , which, by analogy with (32), can be represented as:

In this case, the measuring filter 20 of the second phasing unit 16 ₁ will isolate from the sum of two oscillations only harmonic oscillation with the central frequency of this filter ω _φi = ω _Ci -ω _Pi , which, by analogy with (9) and taking into account (42), can be represented as :

The second term in (43), which characterizes the interference voltage at the output of the first multiplier 19 of the second phasing node 16 _2, will have the form:

Similarly to the previously considered, in the first multiplier of the second phasing unit 16 _{2, the} harmonic oscillation of the interference U _2Пi (t) is converted into two signals with angular manipulation: at the upper carrier frequency ω _в = ω _Пi + ω _Pi (the first term in parentheses) and at the bottom carrier frequency ω _n = ω _Pi -ω _Pi (the second term in parentheses).

The spectral components of the first signal with angular manipulation at the carrier frequency ω _are much higher than the center frequency ω _{Фi of the} measuring filter 20 and are easily eliminated by it, and individual spectral components of the second phase-manipulated signal at the carrier frequency ω _n can fall into the passband of the measuring filter 20.

However, taking into account, as in the first phasing unit 16 ₁ , that the passband of the measuring filter 20 is much smaller than the passband of the channel filter 17, the response of the narrow-band measuring filter 20 to the above individual spectral components of the signal with angular manipulation can be ignored.

Thus, the output voltage of the measuring filter 20 when receiving a mixture of the channel signal and in-band interference with a certain degree of accuracy will correspond to the expression (45):

Using the principle of superposition on the selected interval of stationarity Δt, the output product of the second multiplier 21 of the second phasing node 16 ₂ will have the form:

The first term in (48) determines the voltage of the second sample of the i-th channel signal of the first group signal at the output of the second multiplier 21 of the second phasing unit 16 ₂ . By analogy with (37) and taking into account (42) and (47), this term can be written in the form:

Considering that the first term in (49) is eliminated during further filtering, we will take into account, as before, only the second term at the second input of adder 22, which characterizes the voltage of the second sample of the channel signal:

with average amplitude voltage

The second term in (48) determines the interference voltage at the second input of the adder 22. Taking into account (42) and (47), this term will have the form:

By analogy with (39) in (51), we will take into account only the second term at the second input of adder 22, which characterizes the interference voltage:

where ω ' _Пi = ω _Ci -ω _{Pi -ω Пi} , ψ ₂ = ϕ _2Ci -ϕ _Pi -ϕ _2Пi .

In the adder 22, the amplitudes of the first and second samples of the channel signal of the first group signal from the outputs of the second multipliers of the phasing nodes 16 ₁ , 16 ₂ and determined by expressions (38) and (50) add up algebraically as in-phase oscillations. The resulting voltage of the channel signal at the output of the i-BCS 15 _1-i , which implements the optimal addition of samples of this signal, will look like:

with the average value of the amplitude of the resulting voltage of the channel signal in the form:

The voltage amplitudes of the samples of in-band channel interference defined by expressions (40) and (52) in the adder 22 are added geometrically, like oscillations with the same frequency ω ' _Пi = ω _Ci- ω _{Pi -ω Пi} and different phases: ψ ₁ = ϕ _1Ci- φ _Pi -ϕ _1Пi and ψ ₂ = ϕ _2Ci -ϕ _Pi -ϕ _2Пi .

The resulting voltage of the in-band channel interference at the output of the i-th BCS 15 _1-i will be equal to [12], p. 185:

- амплитуда результирующего напряжения внутриполосной канальной помехи;Where

- the amplitude of the resulting voltage in-band channel interference;

Δψ = ψ ₂ -ψ ₁ ;

We determine the average value of the amplitude U _POPiT1 ( _Δψ ) of the resulting in-band channel noise voltage.

Taking into account the maximum and minimum values of cos (ψ ₂ -ψ ₁ ), which characterizes the phase difference between the voltages of two samples of in-band channel interference, which can vary arbitrarily, including within any of the selected stationary intervals of duration Δt, the average value of U _POPiT1 ( _Δψ ) will be equal to:

provided that M _1iT1 > R _2iT1 , or

provided that R _2iT1 > M _1iT1 .

The resulting channel ratio of the average values of the resulting amplitudes of the signal / noise voltages at the output of the i-th BCS 15 _1-i (at the input of the i-th channel demodulator of the channel demodulator block 11 ₁ ) with optimal (weighted) addition of spaced samples of the i-th channel signal at any a stationary interval of duration Δt selected within the time interval T1 (0 <t <t ₁ ) in accordance with FIG. 3a, b, taking into account (53 ') and (55) it will be equal to:

Or considering (56):

2. Reception at any stationarity interval of duration Δt, selected within the time interval T2 (t ₁ <t <t ₂ ).

In this case, the following inequalities occur within the time interval T2: 1) U _1Pi / U _1Ci = R _1iT2 <1 and 2) U _2Pi / U _2Ci = R _2iT2 <1.

Therefore, the voltage of the first sample of the channel signal at the output of the second multiplier 21 of the first phasing unit 16 ₁ or at the first input of the adder 22, taking into account the identity of the phasing units 16 ₁ and 16 ₂ and in accordance with expressions (50), (50 '), is written as:

with average amplitude voltage

Similarly, the voltage of the second sample of the channel signal at the second input of the adder 22 is recorded:

with average amplitude voltage

In adder 22, the average amplitudes of the first and second samples of the channel signal of the first group signal, defined by expressions (59 ') and (60'), are added algebraically as in-phase oscillations.

The voltage of the resulting channel signal at the output of the i-BCS 15 _1-i , which implements the optimal addition of samples of this signal, taking into account (59 ') and (60'), will look like:

with the average value of the amplitude of the resulting voltage of the channel signal

It is not difficult to show that the voltage of the first sample of in-band channel interference at the first input of adder 22, by analogy with expression (52), will look like:

where ω ' _Пi = ω _Ci -ω _{Pi -ω Пi} , ψ' ₁ = ϕ ' _1Ci -ϕ' _Pi -ϕ ' _1Пi .

Here, as before, it is _understood that each of the values of R _1iT2 , ψ ' ₁ , ϕ' _1Ci , ϕ ' _Pi , ϕ' _1Pi takes a specific numerical value on each selectable stationary interval of duration Δt within the time interval T2 (t ₁ <t <t ₂ ).

The voltage of the second sample of the in-band channel interference at the second input of the adder 22 will have a similar form:

where ψ ' ₂ = ϕ' _2Ci -ϕ ' _Pi -ϕ' _2Пi .

The voltage amplitudes of the samples of in-band channel interference defined by expressions (62) and (63) in the adder 22 are added geometrically, like oscillations with the same frequency ω ' _Пi = ω _Ci- ω _{Pi -ω Пi} and different phases.

The resulting voltage of the in-band channel interference at the output of the i-th BCS 15 _1-i, similar to expression (54), will be equal to:

-Where

-

- the amplitude of the resulting voltage in-band channel interference; Δψ '= ψ' ₂ -ψ '₁;

Let us determine the average value of the amplitude U _POPiT2 ( _Δψ ') of the resulting voltage of the in-band channel interference (64).

Given the maximum and minimum values of cos (ψ ' ₂ -ψ' ₁ ), which characterizes the phase difference between the voltages of two samples of in-band channel interference, which can vary arbitrarily, including within any of the selected stationarity intervals Δt in the time interval T2, the average value of U _POПiT2 ( _Δψ ') will be similar to expression (56) equal to:

provided that R _1iT2 > R _2iT2 , or:

Thus, when conducting radio communications on any stationarity interval Δt within the time interval T2, the resulting channel ratio of the average signal / noise amplitudes at the output of the i-th BCS 15 _1-i or at the input of the i-th channel demodulator of the channel demodulator block 11 ₁ at optimal ( weight) addition of spaced samples of the i-th channel signal taking into account (61 ') and (65) will be equal to:

Or considering (66):

3. Reception at any stationarity interval of duration Δt, selected within the time interval T3 (t ₂ <t <t ₃ ).

In this case, the following inequalities occur within the time interval T3: 1) U _1Pi / U _1Ci = R _1iT3 <1 and 2) U _2Ci / U _2Pi = M _2iT3 <1.

Thus, the conditions for receiving the total signal in the interval T3 are actually similar to the conditions for receiving in the interval T1 and differ mainly in that at the inputs of the phasing nodes 16 ₁ and 16 ₂ the signal-to-noise and interference / signal ratios, each of which is less than unity, have changed in places. Accordingly, taking into account the identity of the phasing nodes, the resulting channel relationship of the average signal amplitudes of the signal / noise at the output of the i-BCS 15 _1-i or at the input of the i-channel demodulator of the channel demodulator block 11 ₁ with optimal (weighted) addition of spaced samples of the i-channel signal by analogy with (57) will be equal to:

Or similarly (58):

Let us evaluate the noise immunity of the reception of the i-th channel signal of the first group signal (Fig. 2, a) in the form of the signal-to-noise ratio at the output of the corresponding linear addition device (at the input of the i-th demodulator) by the known prototype communication system [5].

Given that the receiving complexes of the compared communication systems should work under the same conditions, i.e. to conduct diversity reception on two antennas receiving electromagnetic waves polarized in mutually perpendicular directions, then the fading of two samples of the received group signal at the outputs of two RPUs of both the proposed communication system and the prototype should be the same.

As noted above, when linearly adding signals of two diversity branches, stringent requirements are put forward to ensure equal gain in the diversity branches, i.e. antennas, as well as RPUs of diversity branches should be strictly identical in gain [6]. Assume that this condition is met.

Similarly to the foregoing, we will consider the operation of the well-known communication system - the prototype, when receiving two samples of each i-th channel signal subject to polarization fading in accordance with the model of the nature of fading shown in Fig. 2 a.

Let, as before, the samples of the group signal U _1C (t) and U _2C (t) of the form (3) and (4), as well as the samples of the in-band sinusoidal channel noise U _1Пi (t) and U _2Пi (t) of the form (17 ) and (19) from the outputs of the linear paths of the corresponding RPMs of the receiving complex of the known communication system, they arrive within any of the stationary time intervals of duration Δt for N channel blocks (KB), each of which is designed for dual reception and consists of two identical active filters [5 ].

The voltage of the samples of the i-th channel signal and the i-th in-band channel noise at the inputs of the linear addition device (ULS) as part of the phase difference calculation unit (BVRF) allocated by the filters of the i-channel channel block can be represented similarly to (21) and (22) in the form :

- at the first entrance -

- at the second entrance -

Hereinafter, we will assume that the transmission coefficient of each active filter and each ULS is 1.

It should be noted that for any type of linear signal addition, the frequency components of the spectrum of each of the samples of the i-th channel signal of the diversity branches should not be subject to selective distortion during their addition. From this it follows that the linear addition of the voltage of the samples of the i-th in-band channel interference should be carried out according to the same law as the linear addition of the voltage of the samples of the i-th channel signal.

Using the principle of superposition, the result of incoherent linear addition of voltages of two samples of the i-th channel signal and taking into account (71) and (72) can be represented as:

As noted above, no analytical expression describing with high accuracy the sum of two oscillations of type (73) was found in the technical literature. However, if we take into account that in expressions (73) the pattern | Δω | = | ω _{Ci -ω Пi} | is a small quantity compared to the average circular frequency (ω _Ci + ω _Пi ) / 2, and also, with a certain probability, situations are possible when the function θ _Ci (t), which determines the type of angular manipulation of the ith channel signal, changes stepwise its value within the stationary interval of duration Δt is relatively rare, or remains constant (which can occur when the first group signal is transmitted via the i-th frequency channel, for example, in the OFT or CHT mode, the signal is “pressed” as a series of binary symbols “1 ", Ne edavaemyh within Δt interval), the resulting oscillation can be considered narrowband process [11].

With sufficient accuracy for the necessary calculations, expression (73) in accordance with [11], p. 185 can be written as:

where the envelope U _РЛCi and phase ϕ _{РЛCi of the} resulting voltage of the channel signal at any of the stationary intervals of duration Δt can be determined in the form:

Let us determine the average value of the amplitude U _РЛCi (74) of the resulting voltage of the channel signal after linear addition.

Taking into account the maximum and minimum values of cos (ϕ _2Ci -ϕ _1Ci ), which characterizes the phase difference between the voltages of two samples of the channel signal, which can vary arbitrarily, including within any of the selected stationarity intervals Δt at any time interval T , the average value of U _РЛCi, similar to expression (55), will be equal to:

provided that U _1Ci > U _2Ci , or:

Similarly, taking into account (71) and (72), we can obtain the result of incoherent linear addition of voltages of two samples of the i-th in-band channel noise:

where the resulting value of the voltage amplitude U _RLPi and the resulting value of the phase ϕ _{RLPi of the in-} band channel interference at any of the stationary intervals Δt in accordance with [12], p. 185 are defined as:

The average value of the amplitude of the resulting voltage of the in-band channel interference after linear addition of the voltages of its two samples can be written similarly to (76) and (77) in the form:

either in the form:

Accordingly, when receiving samples of the first group signal and in-band channel interference operating within the frequency band of the i-th channel signal, the resulting channel ratio of the average values of the signal / noise voltage amplitudes at the output of the i-th linear addition device or at the input of the i-th channel demodulator known communication system [5] will be equal to:

1. When receiving at any stationarity interval of duration Δt within the first time interval T1 (0 <t <t ₁ ) and the first half of the time interval T2 (t ₁ <t <2), where U _2Ci > U _1Ci and U _1Пi > U _2Pi and taking into account (77) and (81):

2. When receiving at any stationarity interval of duration Δt within the second half of the second time interval T2 (2 <t <t ₂ ) and the third time interval T3 (t ₂ <t <t ₃ ), where U _1Ci > U _2Ci and U _2Пi > U _1Pi and taking into account (76) and (82):

Thus, the above expressions allow us to estimate the resulting channel signal-to-noise ratios at the output of any i-th addition device (optimal and linear) of spaced samples of the i-th channel signal (or at the input of the i-th channel demodulator) of each of the compared communication systems depending on the initial channel signal-to-noise ratio at the output of the linear path of each of the RPU 10 ₁ and 10 ₂ communication systems.

However, when comparing the values of H _Piopt and H _Pilin, it is necessary to know the critical value of the signal-to-noise ratio (in-band) at the input of each demodulator H _cr. = U _C / U _P , below which its operation fails. Since we previously agreed that the demodulators of the compared communication systems are identical, the value of N _cr. for the compared communication systems should be the same, i.e. N _cr.lin = N _cr.opt .

The numerical value of N _cr for an industrial demodulator of telegraph signals, for example, for a demodulator (block B5-72) from the 3rd generation RPU R-160P [10], is N _cr <1.66 for operating modes of CT and H _cr <2 for OFT modes.

The calculated data of the relations H _Pilin and H _Piopt calculated according to formulas (57), (67), (68), (69), (83), (84) for different values of U _1Ci , U _1Pi , U _2Ci , U _2Pi fixed at relative times t in accordance with FIG. 2a are shown in the table.

From the analysis of the table it follows that the proposed high-speed decameter radio communication system significantly exceeds the well-known radio communication system - the prototype, in noise immunity of receiving any i-th channel signal while simultaneously acting on the input (output) of the linear path of each of the two RPUs 10 ₁ and 10 _{2 of the} corresponding in-band sample channel interference with a frequency within the frequency band occupied by the spectrum of the i-th channel signal.

In addition, since when calculating the noise immunity of the reception of any i-th channel signal, it was assumed that the output amplitudes of the channel filters 17 of the phasing nodes 16 ₁ and 16 ₂ of the voltage amplitudes U _1Pi and U _{2Pi of the} samples of the in-band channel noise were chosen to be significantly larger _{than the} actual average amplitudes voltage of inter-channel interference samples U _1MPi and U _2MPi i.e. U _1Pi >> U _1MPi and U _2Pi >> U _2MPi , then the noise immunity of reception of any i-th channel signal, characterized by the ratio of H _Piopt = U _{PO cp CiTX} / U _POcpPiTX at a specific time interval TX (T1, T2, T3) in accordance with FIG. 3a, it is mainly determined by the magnitudes of the amplitudes of the stresses of the samples of the in-band channel interference and practically does not depend on the action of the stresses of the samples of the inter-channel interference.

It should also be noted that in the proposed communication system when the same duration T _ch channel sending signal as that of the known connection system (prior art) [5], the total bit rate is 2 times higher. Therefore, to compare the signal-to-noise _ratios H _Piopt and H _{Pilin of the} proposed communication system and the prototype when they work under absolutely identical conditions, it is necessary to increase the transmission speed in the prototype by 2 times, for example, by increasing the multiplexing factor k of the signal in each frequency channel of the group signal (Fig. 2, a).

For example, when comparing the proposed communication system in each of the 2N frequency channels, you can use OFT or CT signals at k = 1, and in the prototype (at the same speed of channel signal manipulation), you can use DOFT or DCT signals at k = 2, providing speed binary information transmission is 2 times higher than OFT or CT signals. In this case, the data of calculating the signal-to-noise _ratios H _Pilin and H _Piopt at the input of each i-th demodulator of the compared systems (given in the table) will not change. Only the value N _cr.lin for each i-th demodulator of the known system will change towards higher values when using DOPT or DCT signals, i.e. N _cr.lin > N _cr.opt or N _cr.lin = L N _cr.opt , where the coefficient L> 1. Accordingly, the gain in noise immunity of the proposed communication system in relation to the prototype should actually increase by an additional L times, i.e. the gain Q in noise immunity can be determined in the form:

where the values of H _Pipt and H _Pilin for various relative units of time t are given in the table.

From the above analysis it follows that the advantage of the proposed communication system with respect to the prototype for the noise immunity of reception of each i-th channel signal under the action of the i-th interference concentrated in the spectrum is achieved due to the fact that when the group data transfer rate is 2 times higher data on each of the 2N frequency channels does not increase, in addition, when performing dual diversity reception on two antennas 9 ₁ and 9 ₂ , receiving waves polarized in mutually perpendicular directions, the optimal (weighted) summation of the voltages of two samples of each of the 2N channel signals is used, which, in turn, provides an increase in the channel signal-to-noise ratio at the input of the corresponding channel demodulator of the channel demodulator block 11 ₁ or 11 ₂ .

, занимаемую одним из 2N канальных сигналов первого и второго групповых сигналов.Since the channel signals of each group signal are orthogonal, i.e. are independent of each other, then the noise immunity of receiving any i-th channel signal of the first and second group signals does not depend on the noise immunity of receiving any other channel signal. Accordingly, a gain is also achieved in the noise immunity of receiving the total signal as a whole under interfering effects of one channel interference concentrated in the spectrum of interference, and with the simultaneous action of G channel interference concentrated in the spectrum (G≤N), each of which falls into the frequency band

occupied by one of the 2N channel signals of the first and second group signals.

The gain in noise immunity of the reception will also be achieved if there is an interfering effect of fluctuation interference acting within the frequency bands occupied by the channel signals, due to the fact that in each BCS the samples of the channel signal are added algebraically as in-phase oscillations, and the fluctuation interference (noise) is added geometrically .

From a practical point of view, all the components of a decameter radio communication system with high-speed data transmission are realizable, moreover, most of the components of the system (with the exception of transmitting antennas 7 ₁ , 7 ₂ , receiving antennas 9 ₁ , 9 ₂ and power amplifiers as part of radio transmitting devices 6 ₁ , 6 ₂ ) can be implemented using methods of digital signal processing [10].

In conclusion, it should be noted that the implementation of the present invention is a decameter radio communication system with high-speed data transmission will achieve the following advantages in relation to the known systems [1], [4], [5]:

1. To increase the group data transfer rate by 2 times without actually expanding the frequency band when transmitting and receiving group signals.

2. To increase the noise immunity of the reception of binary information both under the interfering effect of spectrum-focused (sinusoidal) noise and fluctuation noise.

3. To expand the functionality with respect to the known system [5] in terms of transmission and reception of a wider class of channel signals with angular manipulation: both with phase difference and frequency manipulation for any multiplex k of channel signal multiplexing.

Information sources

1. Klovsky D.D. Theory of signal transmission. Textbook for high schools. M .: Communication, 1973, 376 p.

2. Fink L.M. Theory of discrete message transmission. M .: Soviet Radio, 1970, 728 p.

3. Klovsky D. D., Nikolaev B. I. Engineering implementation of radio circuits (in discrete message transmission systems under conditions of intersymbol interference). M .: Communication, 1975, 200 p.

4. Kiselev A.M., Mahotin V.V., Ryzhov N.Yu., Shatalova G.V. A method for implementing a high-speed parallel modem // Radio engineering. 2006. Issue. 11. S. 5-15.

5. Ginsburg VV, Girshov B.C., Zayezny A.M., Kagan BD, Kustov OV, Okunev Yu.B. and other equipment for the transmission of discrete information MS-5 / Edited by Zaezdny A.M. and Okuneva Yu.B. - M .: Communication, 1970, 152 p.

6. N.A. Sartasov, V.M. Edvabny, V.V. Mushroom. Short-wave trunk radio receivers. M .: Communication, 1971, 288 p.

7. G.Z. Eisenberg, S.P. Belousov, E.M. Zhurbenko, G.A. Kliger, A.G. Kurashov. Shortwave antennas / Edited by G.Z. Eisenberg. - 2nd, rev. and add. - M.: Radio and Communications, 1985, 536 p.

8. M.V. Nazarov, B.I. Kuvshinov, O.V. Popov. Theory of signal transmission. M.: Communication, 1970, 368.

9. M.P. Dolukhanov. Propagation of radio waves. M .: Communication, 1972.336 s.

10. Berezovsky V.A., Dulkeit I.V., Savitsky O.K. Modern decameter radio communication: equipment, systems and complexes / Ed. V.A. Berezovsky. - M.: Radio Engineering, 2011, 444 p.

11. I.S. Honorovsky. Radio circuits and signals. Textbook for high schools. Ed. 3rd, rev. and add. M .: Sov. Radio 1977, 608 pp.

12. Bronstein I.N., Semendyaev K.A. Math reference. M .: Fiz.-mat. literature, 1962, 608 p.

Claims (1)

A decameter radio communication system with high-speed data transmission, comprising a transmitting complex comprising a serially connected message source, an encoder and a serial-parallel converter, the outputs of which are connected to the corresponding inputs of the block N channel manipulators, the output of which is connected to a serially connected radio transmitting device and transmitting antenna, as well as a receiving complex containing two receiving antennas, the output of each of which is connected to the input of the corresponding radio a removable device, and a block Ν channel demodulators, the outputs of which are connected to the corresponding inputs of a parallel-serial converter, the output of which is connected to a series-connected decoder and message receiver, characterized in that an additional block N channel manipulators, the inputs of which are connected to the transmitting complex, are connected in series with corresponding additional outputs of the serial-parallel converter, additional radio transmitting device and an additional transmitting antenna, and an additional block of N channel demodulators and 2Ν blocks of coherent signal addition (BCS) are introduced into the receiving complex, one input of each of which is combined with the output of one radio receiver, and the other input of each BCS is combined with the output of another radio receiver, and the output of each BCS from the composition of the first group of N BCS with serial numbers from 1 to N is connected to the corresponding input of the block of N channel demodulators, and the output of each BCS from the composition of the second group of Ν other BCS with p by serial numbers from 1 to N it is connected to the corresponding input of an additional block of N channel demodulators, the outputs of which are connected to the corresponding additional inputs of the parallel-serial converter, each BCS contains two phasing nodes, each of which contains a channel filter connected in series, the input of which is the corresponding input of the BCS , a normalizing amplifier, a first multiplier, a measuring filter and a second multiplier, the other input of which is connected to the input of the first of the multiplier, the output of the second multiplier of each phasing unit is connected to the corresponding input of the adder, the output of which is connected to the input of the filter of the resulting oscillation, the output of which is the output of the BCS, connected through the normalizing amplifier of the resulting oscillation to another input of the first multiplier of each phasing unit.

Images