RU2652455C2 - Device for impedances matching - Google Patents

Abstract

FIELD: physics.

SUBSTANCE: invention relates to radio communication means for power line matching. Device provides the possibility of an accurate electrodynamic calculation of the elements of the scattering matrix of the named compounds without using the equivalent circuit method by using tees, matched by the input arm when being connected to the other two arms of the matched loads of tees with two symmetrical arms. Symmetrical arms of one tee across two transmission lines are connected to the symmetrical arms of the second tee, to the output arm of which the matched load is connected. Example of matching with a power feeder of an antenna, having a vibrator size of 1/10 wavelength in free space, is given.

EFFECT: technical result is to reduce the reflections from the input impedance of the load.

3 cl, 3 dwg

Description

The invention relates to the field of radio communications and can be used to match the transmission line with the impedance (complex load resistance) at one frequency, and with a given tolerance for the deviation of the reflection coefficient from zero in the frequency band. The most promising application of the proposed technical solution for radio communication with mobile devices or communication on radio frequencies, when the antenna dimensions are much less than the wavelength in free space.

Known standard devices for matching an antenna, which has in general a complex input impedance, with a feed path. Quarter-wave transformers, single-loop, two- and three-loop impedance transformers, double-plate dielectric transformers are used [1, p. 216]. In some developments, devices such as a “glass”, a U-elbow, an exciting gap, a current compensator of the outer shell of a coaxial line are used [2, p. 259]. A design of an impedance transformer with smooth tuning for ease of use is proposed. [3].

Known impedance transformers contain T-shaped connections of transmission lines. Studies have shown that such compounds cannot be fully coordinated [4, p. 149]. Comparison with experiment also gives approximate values of the matching effect calculated from the scattering matrix, the elements of which were obtained not by a purely electrodynamic method, but using the equivalent circuit method [1, p. 168; 5, p. 89].

The main disadvantage of the above analogues is the impossibility of using them to coordinate with the supply line a load having an arbitrary complex input impedance.

In recent years, matching circuits on discrete elements have become widespread, allowing you to use the capabilities of computer technology to optimize their ratings. Mostly such schemes are used for mobile devices. The latter have small-sized antennas, short in comparison with the wavelength, an indicator of the received signal and non-linear elements for fine tuning the circuit with a control signal. Such technical solutions of recent years are given below.

The matching circuit for adaptive impedance matching in radio devices [6] includes a set of matching elements that are automatically included in the circuit to obtain the maximum received signal power by the indicator. The need for such a adjustment is dictated by a change in the input impedance of the antenna depending on the position of the mobile device. This device can be in the user's hand, in a pocket, etc. When a radio device operates on two carrier frequencies, there are two sets of matching elements.

Known methods for matching complex resistances and devices for its implementation [7, 8]. Matching devices are made of reactive and resistive lumped elements, and a nonlinear element is introduced to provide frequency-tunable matching. The values of the elements are determined by calculation from a system of equations that reflects the conditions of full coordination at several frequencies of the working frequency band.

In the operational calibration device for matching the antenna in the radio reception system [9], matching circuit elements are provided that are adjusted to match the input impedance of the antenna.

The lack of matching devices in the form of a set of discrete elements is identical to that noted above - it is impossible to coordinate a load having an arbitrary complex input impedance with the supply line of the transmission. We can note one more drawback, which consists in the use of resistors in a set of discrete elements. The latter can contribute to the reduction of reflections from the matching device, but cannot ensure the complete transmission of signal power to the load. This follows from the law of conservation of energy. In [7, 8], the reflection coefficient from the input of the matching device is determined by the element S _{11 of} the scattering matrix, which is not such. For this reason, doubts arise about the correctness of optimizing the values of a set of discrete elements.

As a prototype, the matching device described in [1, p. 223], which uses T-shaped connections of two transmission lines forming a tee. A signal source 5 is connected to the input arm of tee 1, the other arm is connected to the input arm of tee 2. A matched load Ζ _{H is} connected to the output arm of this tee. The remaining shoulders of the tees are loaded with line segments with movable short circuits 3 and 4. For matching, it is sufficient to determine the sizes L ₁ and L ₂ . Previously, this was done in a pie chart, now it is easier to use a simple program.

In [1], the disadvantages of the prototype were also noted — the impossibility of matching at any values of the input load impedance. The pie chart has an “unreachable zone” for some values of this impedance. The disadvantages noted above for T-shaped compounds remain significant.

The technical result of the claimed invention is the possibility of more complete use of the signal transmitted along the line without its reflections or with a significant reduction in reflections from the input load impedance. At a fixed frequency, almost complete load balancing is achieved with an arbitrary input impedance. The term “practically” should be understood in such an aspect that at this stage in the development of such devices, the absorption of electromagnetic energy in a metal or dielectric is not taken into account. This refers to the walls of the hollow waveguide, the outer shield, the central conductor and the dielectric of the coaxial cable, the strip and the dielectric of the microstrip or strip line. Within certain acceptable values of the reflection coefficient, a working frequency band is formed near the frequency of full agreement.

The specified technical result is achieved by the fact that in the device for matching impedances, including the first and second tees, according to the invention, the tees are made in the form of self-consistent tees, a signal source is connected to the input of the first tee, the symmetrical shoulders of this tee are connected to the symmetrical shoulders of the second tee via two transmission lines , to the output arm of which a load is connected, the input impedance of which must be coordinated with the input transmission line, and in one of the transmission lines there is a phase rotation spruce.

In addition, this technical result is achieved in that the tees can be made in the form of a segment of a rectangular waveguide, divided into parts of its length by a thin metal partition parallel to a wide wall.

In addition, the specified technical result is achieved in that the tees can be made in the form of a Y connection of coaxial or strip lines with a wave impedance of one of the arms, half the wave impedance of the other two arms.

Self-consistent tees are those tee connections of transmission lines that are obtained consistent along the input arm when connected to the other two arms of the agreed loads. It turned out that the elements of the scattering matrix of at least some of these tees can be calculated strictly electrodynamically, without resorting to the equivalent circuit method. A similar calculation has not been performed before. For this reason, the elements of the scattering matrix are determined exactly, without any simplifying assumptions. The use of such tees for coordination allows in practice to obtain results that correspond to the calculated ones.

A diagram of the proposed matching device for transmission lines in the form of a rectangular waveguide is shown in FIG. 1. A segment of the main rectangular waveguide 1 is divided by a thin metal plate 2 into two partial waveguides. In one of the partial waveguides placed phase shifter in the form of two plates 3, pressed against narrow walls. The design of the phase shifter may be different, for example, in the form of a dielectric plate parallel to the narrow wall of the waveguide. The left end of the waveguide in the figure is the input end, a matched load with a known complex reflection coefficient is connected to the right one.

To calculate the matching effect, the scattering matrix apparatus was used [10]. In FIG. 1 shows the complex amplitudes of the waves propagating to the right P ₁ , P _2B , P _2H , P _3B , P _3H , P ₄ and to the left L ₁ , L _2B , L _2H , L _3B , P _3H , L ₄ . The amplitudes are counted in the planes of the cross section N ₁ , N ₂ , N ₃ . The distances d ₁₂ , d ₂₃ , d ₃₄ between the sections are multiples of the wavelength of type H ₀₁ in the waveguides. The latter is due to the simplification of the calculation, but can be easily performed if there are no strict restrictions on the dimensions. The ends of the plate 2 should be located approximately in the middle or slightly closer to the phase shifter in order to form the field H ₀₁ in the reference planes of the main waveguide without strong influence of higher types of fields arising near the edges of the plate.

When the above conditions are met in all reference planes, only the main wave field of the rectangular waveguide is formed. The design of the parts of the device between the planes N ₁ -N ₂ and N ₃ -N ₄ is a self-consistent tees. The relations between the wave amplitudes for the tee N ₁ -N ₂ have the form:

Similar relations can be written for the tee N ₃ -N ₄ . A rigorous electrodynamic calculation of the elements of the scattering matrix A _ik has not yet been reflected in the literature.

The design of the part of the device between the planes N ₂ -N ₃ is a four-terminal network. As a result, we have 11 equations for 12 complex amplitudes.

;

; L_2B=e^jϕL_3B P_3H=P_2H;

;

; L _2B = e ^jϕ L _3B P _3H = P _2H ;

;

; L_3B=e^jϕL_2B P₄=L₄/K;

;

; L _3B = e ^jϕ L _2B P ₄ = L ₄ / K;

;

; L_2H=L_3H.

;

; L _2H = L _3H .

In this system of equations, K has the meaning of a complex reflection coefficient from a matched load in the N ₄ plane, which we consider known. Some abbreviations were not carried out to highlight the numerical values of the elements of the scattering matrices. It is possible to arbitrarily select the module and phase of the wave fed to the input of the device. Therefore, we put P ₁ = 1. Then we get the completely correct problem of determining 11 unknown amplitudes connected by 11 algebraic equations. Omitting the mathematical transformations, we write out the resulting solution.

;

;

;

;

;

; P_3H=P_2H;

;

; P _3H = P _2H ;

;

; L_3B=е^-jϕL_2B;

;

; L _3B = e ^-jϕ L _2B ;

P _3B = e ^jϕ P _2B ; P ₄ = L ₄ / K; L _3H = L _2H .

и фазой γ коэффициента отражения в плоскости N₄

и фазой ϕ, вносимой фазовращателем. При полном согласовании

, что приводит к уравнениюThe wave amplitudes in individual segments of the device transmission lines are determined by the module

and phase γ of the reflection coefficient in the plane N ₄

and phase ϕ introduced by the phase shifter. With full agreement

, which leads to the equation

which determines the values of the phases ϕ and γ. When the phase of the reflection coefficient of the coordinated load ψ does not coincide with γ, it is impossible to connect the load directly to the output of the device in section N ₄ , but it can be connected through a line segment of length l, determined from the condition

,

,

in which β is the wave number of the wave in the line.

Based on the obtained calculation results, the application algorithm of the proposed device can be described in the following sequence.

и фаза ψ коэффициента отражения согласуемой нагрузки.1. Measured (or calculated) module

and phase ψ of the reflection coefficient of the matched load.

решается комплексное уравнение (1) относительно ϕ и γ.2. With a known value

the complex equation (1) is solved with respect to ϕ and γ.

3. The length l of the segment of the transmission line is found, through which a coordinated load should be connected to the output of the device for matching impedances.

либо

or

According to the second formula, the length l is determined with a negative value obtained by the first formula.

4. The calculated value of the phase shift ϕ of the phase shifter for the waveguide device is set. For coaxial, microstrip or strip lines, the length l _{1 of the} additional line segment is calculated by the formula

5. The agreed load is connected to the device through a segment of the transmission line of the calculated length l.

After performing these steps, the load is matched at the selected frequency, and with some tolerance for the deviation of the reflection coefficient from zero, matching also occurs in the frequency band. This band depends not only on the tolerance mentioned, but also on the frequency dispersion of the input load impedance and the wave impedance of the line. For this reason, the determination of the matching band is more complex and is not the purpose of the invention.

The design of the matching device and the algorithm for its application are not tied to the type of transmission line and the specific frequency of full coordination. All calculations can be carried out in terms of the modulus of the reflection coefficient, phase shifts and wave impedances of the used transmission lines. Therefore, it is possible to cite the design of the proposed device for other transmission lines that differ from a rectangular waveguide. It is only important that self-consistent tees exist for the transmission line used. A two-wire line is not suitable due to noticeable radiation in the free space. The coaxial and strip devices are shown in FIG. 2.

The self-consistency condition for coaxial and strip tees is easily obtained from the continuity of the potential difference between the signal conductors and the screen and the continuity of currents in the signal conductors for the TEM wave [10, p. 53]. This condition is that the wave impedance of one arm should be 2 times less than the wave impedance of the other two arms. In FIG. 2, this ratio is qualitatively reflected by the thickness of the signal conductor or strip. The scattering matrices coincide with those described above for the waveguide matching device. The difference from the latter is only in the design of the phase shifter. For coaxial lines (Fig. 2a), an additional segment of a coaxial line of length l ₁ that can be laid in the form of a ring is convenient. For strip lines (FIG. 2b), an additional piece of strip line may be included. Coordinated rotations of the strip line have sections, the geometry of which has been worked out for a 90 ° rotation [11, p. 178].

The calculation of the proposed device is illustrated below by the example of matching an asymmetric vibrator with a length of 1/10 wavelength in free space, the parameters of which are taken from [2, p. 267]. The input impedance calculated by the Stretton and Chu method is 5-i150 Ohms, the reflection coefficient for the 75 Ohm path is 0.585-i0.778. Without coordination, the power fraction of 0.948 is reflected. It is necessary to substitute the modulus of the reflection coefficient 0.974 in equation 1 and find the real values of ϕ and γ. We obtain ϕ = 2.91 and γ = 3.37. According to the author, the easiest way to do this is numerically. Attempts to analytically solve this complex equation lead to the need to separate complex roots that correspond to either paths with active losses or paths with an active medium. The lengths of lines l and l _{1 are} determined by formulas (2) and (3). For coaxial line

where c is the speed of light in free space, f is the signal frequency, and ε is the relative permittivity of the dielectric at this frequency. For a frequency of 1 GHz and ε = 3, we have l = 59 mm and l ₁ = 80 mm. The dimensions of the matching device are determined by the wavelength in the coaxial line 2π / β = 173 mm, which gives the distance between the reference planes of the field amplitudes. For a microstrip or strip line ε in the formula (4), it makes sense the effective value of the dielectric constant of the dielectric.

отраженной волны от входа, выравниваются интенсивности волн P_2H, L_2H и L_2B, а разность квадратов модулей P_2B и L_2B в точках 1 и 2, определяющая поток энергии через согласующее устройство, обращается в 1. Последнее соответствует полному согласованию. Проходящая в нагрузку мощность определяется проникшим через сечение N₄ полем

.The physical mechanism of the matching effect is illustrated in FIG. 3, which shows the change in the wave intensity depending on the variation of the phase ϕ for the given example. At the optimum phase, as a result of wave interference, the intensity decreases to zero

of the reflected wave from the entrance, the intensities of the waves P _2H , L _2H and L _{2B are} aligned, and the difference between the squares of the modules P _2B and L _2B at points 1 and 2, which determines the energy flow through the matching device, turns to 1. The latter corresponds to full agreement. The power passing into the load is determined by the field penetrated through section N ₄

.

The advantages of the proposed device for matching impedances in comparison with known analogues are as follows.

1) An accurate and simple calculation of design parameters is possible with a known value (calculated or measured) of the impedance of the coordinated load.

2) The design and use of the device is possible in a wide frequency range from about 300 MHz to 30 GHz when using transmission lines of the appropriate type.

3) The use of microelectronics technology allows the device to be used at higher frequencies than those specified in paragraph 2).

4) In the absence of restrictions on the dimensions, it is possible to use the device at lower frequencies compared to those specified in paragraph 2).

5) It is possible to calculate the effect of increasing the field intensity in individual lines of the device compared to the input field. This allows an assessment of the maximum input power.

6) The highest field intensity occurs in the line with the connected load (| P ₄ | in Fig. 3). This allows us to consider the proposed device as optimal for matching the impedance of the load with the supply line at maximum power.

Literature.

1. Lebedev I.V. Microwave equipment and devices. T. 1, M .: Higher school, 1970.

2. Eisenberg G.Z. Antennas of ultrashort waves. M .: State. Publishing House for Communications and Radio, 1967.

3. Dolzhenkov A.A., Polyanskiy M.Yu. Matching device. AC No. 579670, Н01Р 3 | 08, 1977.

4. Harvey A.F. The technique of superhigh frequencies. Translation from English, ed. B.I. Sushkevich. T. 1. M.: Sov. radio, 1965.

5. Megla G. Technique of decimeter waves. Translation from it. Owls Radio, M., 1958.

6. NGUEN Hein, ZHOU Yi, PARPIA Vijay. Matching circuit for adaptive impedance matching in radio devices. RF patent No. 2497306, H04W 4/00, 2010.

7. Golovkov Α.Α., Malyutina I.A. The method of coordination of complex resistances and the device for its implementation. RF patent No. 2486667, H03J 3/00, 2013.

8. Golovkov Α.Α., Malyutina I.A. The method of coordination of complex resistances and the device for its implementation. RF patent No. 2486667, H03J 3/00, 2013.

9. Ali Shirook M., Warden James. Dynamic real-time calibration for antenna matching in a radio frequency receiver system. Publication info: US 2014210686 (A1), H01Q 1/50, 2014.

10. Altman, J. L. Devices of superhigh frequencies. Translation from English M.: Mir, 1968.

11. Nefedov E.I., Fialkovsky A.T. Strip transmission lines. M .: Nauka, 1980.

Claims (3)

1. Device for matching impedances, including the first and second tees of transmission lines, characterized in that the tees are made in the form of self-consistent, matched along the input arm when connected to the other two arms of the coordinated loads, tees with two symmetrical arms, the source is connected to the input of the first tee signal, the symmetrical shoulders of this tee are connected through two transmission lines to the symmetrical shoulders of the second tee, to the output arm of which a matched load is connected, and in one of the lines The first gear has a phase shifter. 2. The device according to claim 1, characterized in that the tees are made in the form of a segment of a rectangular waveguide divided into parts of its length by a thin metal partition parallel to a wide wall. 3. The device according to claim 1, characterized in that the tees are made in the form of a Y connection of coaxial or strip lines with a wave impedance of one of the arms, half the wave impedance of the other arms.

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