RU2528405C1 - Clock synchronisation method and device therefor - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: device which carries out the disclosed method comprises a relay artificial earth satellite and two spaced-apart ground-based stations A and B. Each ground-based station comprises a frequency and time standard 1, heterodynes 2.1 and 2.2, a pseudo-noise signal generator 3, a switch 4, mixers 5, 13 and 19, a first intermediate frequency amplifier 6, power amplifiers 7 and 12, a duplexer 8, a transceiving antenna 9, clippers 10 and 15, buffer storage devices 11 and 16, a delay and delay derivative measuring device 17, a +90° phase changer 18, second intermediate frequency amplifiers 14 and 20, a -90° phase changer 21, an adder 22, multipliers 24, 30 and 31, amplitude detectors 25, 32 and 33, a switch 26, a calibrator 27, controlled phase changers 28 and 29, a subtractor unit 34, low-pass filters 35 and 38, inverted amplifiers 36 and 39.

EFFECT: high noise-immunity and accuracy of synchronising remote time scales through complete suppression of spurious signals received over an image channel by identifying receiving channels.

2 cl, 4 dwg

Description

The proposed method and device relate to communication technology and can be used in radio interferometry with extra-long bases (VLBI), as well as in the service of a single time and frequencies.

Known methods and devices for clock synchronization (ed. Certificate of the USSR No. 591.799, 614.416, 970.300, 1.180.835, 1.244.632, 1.278.800; RF patents No. 2.001423, 2.003.157, 2.040.035, 2.146.833, 2.177.167, 2.292.574; German patent No. 3.278.943; EP patent No. 0.564.220; Gubanov BC, Finkelstein AM, Friedmann P.A. Introduction to radio astrometry. - M., 1983, and others).

Of the known methods, the closest to the proposed one is the “Clock synchronization method (RF patent No. 2.292.574, G04C 11/02, 2005), which is selected as a prototype.

The specified method provides a comparison of time scales spaced over long distances, and is based on the use of the duplex method of communication through a geostationary satellite repeater.

The main advantage of the duplex communication method is that it eliminates the length of the signal path. Therefore, its accuracy mainly depends on the parameters of the onboard transponder, the type of signal used, and the technique for measuring time intervals.

The known method also provides the suppression of false signals (interference) received through the mirror and Raman channels.

It should be noted that the conversion of the frequency of the false signal (interference) received on the mirror channel occurs with the same conversion coefficient K _ol as on the main reception channel. Therefore, it most significantly affects the selectivity and noise immunity of the method.

However, the complete suppression of false signals (interference) received on the mirror channel is possible only with the identification of the receiving channels. Real amplifiers of the second intermediate frequency and other elements that are part of the receiving channels have different characteristics. Therefore, the complete suppression of false signals (interference) received via the mirror channel does not occur, and thus the noise immunity and accuracy of synchronization of remote time scales are reduced.

An object of the invention is to increase the noise immunity and accuracy of synchronization of remote time scales by completely suppressing false signals (interference) received on the mirror channel by identifying the receiving channels.

The problem is solved in that the clock synchronization method, based, in accordance with the closest analogue, on the simultaneous reception of microwave noise signals from an artificial Earth satellite by spaced ground points, coherently converting them to a video frequency, digitally recording the received signals and determining the time delay of one of the same signal to synchronization points by the method of correlation processing of registered signals, the magnitude of which compares time scales, while at the beginning ny ₁ at time t by the clock of the first item with a noise-like code sequence form the microwave signal, it is recorded at the same point, the conditioned signal is converted to a frequency ω _1, increase its power emit the amplified signal in a direction of the artificial satellite land- repeater in the same time t _1, the clock of the second paragraph with the same code sequence form the same noise-like microwave signal, it is recorded in the second paragraph, taking onboard equipment artificial cn pendulum Earth relay signal at the frequency ω _1, re-emit it to the first and second receiving points at frequency ω ₂ while preserving phase relationships, at an arbitrary time t ₃ by the clock of the second paragraph similarly formed and retransmit noise-like microwave signal generated signal is converted to frequency ω _1, increase its power emit the amplified signal in the direction of the same satellite repeater in the same time t ₃ by the clock of the first item by using the same code sequence form the same noise-like microwave Igna, register it in the first paragraph, taking onboard equipment satellite repeater signal at frequency ω ₁ and transmitting it to the first and second points at the frequency ω ₂ while preserving phase relationships, the received signal at the frequency ω ₂ is converted by frequency in the two receiver channels with using the voltage of the second local oscillator in the first receiving channel, and the voltage of the second local oscillator shifted in phase + 90 ° in the second receiving channel, isolating the voltage of the second intermediate frequency in the first and second received channels, shifting in phases at -90 ° voltage of the second intermediate frequency of the second reception channel, it is summed with the voltage of the second intermediate frequency of the first reception channel, re-emit the obtained total voltage with the received signal, allocate the harmonic voltage at frequency ω _r2 of the second local oscillator is detected and it is used to permit further processing of the received signal differs from the closest analogue in that in order to eliminate the non-identity of the receiving channels, a complex (amplitude-phase) identification system is introduced, in which uses a harmonic calibrated signal, the frequency ω _{K of} which differs from the second intermediate frequency ω _CR2 by a small amount Δω ₁ = ω _{K -ω CR2} , pass a harmonic vibrational signal through the first and second receiving channels, select harmonic calibrated signals at their outputs, and detect them and fed to the two inputs of the subtraction unit, the output of which is formed by a low-frequency voltage of positive or negative polarity, they affect the control inputs of the amplifiers of the second intermediate The frequencies of the first and second receiving channels, changing their transmission coefficients so that the output voltage of the subtraction unit tends to zero, realizing the amplitude identity of the receiving channels, at the same time, the harmonic calibration signals extracted in the first and second receiving channels are multiplied among themselves, and the low-frequency voltage is positive or negative polarity, they affect the control inputs of the adjustable phase shifters installed in the first and second receiving channels, changing the phase e shifts of calibration signals so that the output low-frequency voltage tends to zero, realizing the phase identity of the receiving channels.

The problem is solved in that the clock synchronization device, containing, in accordance with the closest analogue, an artificial Earth satellite and two separated ground stations, each of which contains a frequency and time standard in series, a pseudo-noise signal generator, a switch, a first mixer, the second input of which through the first local oscillator connected to the second output of the frequency and time standard, an amplifier of the first intermediate frequency, a first power amplifier, a duplexer, the input-output of which is connected to a transmitting antenna, a second power amplifier and a second mixer, the second input of which is connected through the second local oscillator to the second output of the frequency and time standard, connected in series to the second output of the pseudo noise signal generator, the first clipper, the second input of which is connected to the third output of the frequency and time standard, the first a buffer memory and a meter of delays and their derivatives, the output of which is the output of the device, the first amplifier of the second intermediate part connected in series stots, adder, the second input of which is connected through the phase shifter by -90 ° to the output of the second amplifier of the second intermediate frequency, the first multiplier, the second input of which is connected to the output of the second power amplifier, the first narrow-band filter, the first amplitude detector, the key, the second input of which is connected to the output of the adder, a second clipper, the second input of which is connected to the third output of the frequency and time standard, and a second buffer memory device, the output of which is connected to the second input of the delay meter and their production one, serially connected to the output of the second local oscillator + 90 ° phase shifter and the third mixer, the second input of which is connected to the output of the second power amplifier, differs from the closest analogue in that it is equipped with a calibrator, two adjustable phase shifters, two amplifiers, two low-pass filters, a subtraction unit, a second and third narrow-band filters, second and third amplitude detectors, and a second narrow-band amplifier is connected in series to the output of the first amplifier of the second intermediate frequency a filter, a second amplitude detector, a subtraction unit, the second input of which is connected through a series-connected third narrow-band filter and a third amplitude detector to the output of the second amplifier of the second intermediate frequency, the first low-pass filter and the first inverse amplifier, the two outputs of which are connected to the control inputs of the first and the second amplifiers of the second intermediate frequency, respectively, to the output of the second narrow-band filter, a second multiplier is connected in series, the second input of which is dinene with the output of the third narrow-band filter, the second low-pass filter and the second inverse amplifier, the two outputs of which are connected to the control inputs of the first and second adjustable phase shifters, respectively, each of which is connected between the output of the mixer and the input of the amplifier of the second intermediate frequency of the corresponding receiving channel, the second inputs of the first and a second adjustable phase shifter connected to the output of the calibrator.

The geometric arrangement of ground points A, B, satellite A repeater S is shown in figure 1, where the following notation is introduced: O is the center of mass of the Earth; d is the base of the interferometer; r is the radius vector of the satellite repeater. The timing diagram of the duplex clock comparison method is shown in FIG. 2, where the following notation is introduced: S, A, B-time scales of the satellite relay and ground stations A and B, respectively. The structural diagram of the equipment of one of the ground points (A) that implements the proposed method for clock synchronization, is presented in figure 3. A frequency diagram illustrating the formation of additional receiving channels is shown in FIG.

Each ground station contains a frequency and time standard 1 in series, a pseudo noise signal generator 3, a switch 4, a first mixer 5, a second input of which is connected to a second output of a frequency and time standard 1 through a first local oscillator 2.1, an amplifier 6 of a first intermediate frequency, a first amplifier 7 power, duplexer 8, the input-output of which is connected to the transceiver antenna 9, the second power amplifier 12, the second mixer 13, the second input of which is connected to the second output of the standard 1 frequencies through the second local oscillator 2.2 s and time, the first adjustable phase shifter 28, the second input of which is connected to the output of the calibrator 27, the first amplifier 14 of the second intermediate frequency, the adder 22, the second input of which is connected through the phase shifter 21 by -90 ° to the output of the second amplifier 20 of the second intermediate frequency, the first multiplier 23, the second input of which is connected to the output of the second power amplifier 12, the first narrow-band filter 24, the first amplitude detector 25, key 26, the second input of which is connected to the output of the adder 22, the second clipper 15, the second input of which is connected ene with 1 third output standard time and frequency, the second buffer memory 16 and a meter 17 delays and their derivatives, whose output is the output device. The first clipper 10, the second input of which is connected to the third output of the frequency and time standard, and the first buffer memory 11, the output of which is connected to the second input of the delay meter 17 and their derivatives, are connected in series to the second output of the pseudo-noise signal generator 3. To the output of the second local oscillator 2.2, a phase shifter 18 is connected + 90 °, a third mixer 19, the second input of which is connected to the output of the second power amplifier 12, a second adjustable phase shifter 29, the second input of which is connected to the output of the calibrator 27, the second amplifier 20 of the second intermediate frequency, a third narrow-band filter 31, a third amplitude detector 33, a subtraction unit 34, the second input of which is connected through a second narrow-band filter 30 and a second amplitude detector 32 in series with the output of the first amplifier I 14 of the second intermediate frequency, a first filter 35 and lowpass first inverted amplifier 36, whose two outputs are connected to control inputs of the first 14 and second 20 second intermediate frequency amplifiers, respectively. The second multiplier 37, the second input of which is connected to the output of the third narrow-band filter 31, the second low-pass filter 38 and the second inverse amplifier 39, the two outputs of which are connected to the control inputs of the first 28 and second 29 adjustable phase shifters, respectively, are connected to the output of the second narrow-band filter 30 in series.

Clock synchronization by the proposed method is as follows.

At time t ₁ ^A, according to the clock of the first point A, a noise-like microwave signal is generated using a code sequence (signal α ₁ ) (figure 2):

u _c (t) = U _c cos [ω _c t + φ _k (t) + φ _c ], 0≤t≤T _c ,

where U _c , ω _c , φ _c , T _c - amplitude, carrier frequency, initial phase and signal duration;

φ _k (t) = {0, π} - manipulated component phase mapping law phase shift keying in accordance with the code sequence M (t), wherein φ _k (t) = const at kτ _E <t <(k + 1) τ _Oe and can change abruptly at t = kτ _Oe , i.e. at the boundaries between elementary premises (K = 1, 2, ... N-1);

τ _E , N - the duration and number of chips that make up a signal of duration T _s (T _s = Nτ _E ),

in generator 3 using standard 1 frequency and time.

The specified signal is supplied to the input of the clipper 10, and then is registered in the buffer memory 11. Registration is synchronized by standard 1 frequency and time.

U _c (t) generated signal is supplied to a first input of the first mixer 5 to the second input of which a voltage is applied first LO 2.1

U _Г1 (t) = U _Г1 cos (ω _Г1 + φ _Г1 ).

At the output of the mixer 5, voltages of combination frequencies are generated. Amplifier 6 distinguishes the voltage of the first intermediate (total) frequency

u _CR1 (t) = U _CR1 cos [ω _CR1 t + φ _k (t) + φ _CR1 ], 0≤t≤T _c ,

;Where $$U_{PR\mathrm{one}}=\frac{\mathrm{one}}{2}{K}_{\mathrm{one}}{U}_c{U}_{G\mathrm{one}}$$

;

K ₁ - gear ratio of the mixer;

_pr1 ω = ω _c + ω _D1 - first intermediate (cumulative) frequency;

φ _CR1 = φ _c + φ _G1 ,

which, after amplification in the power amplifier 7, through the duplexer 8 and the transceiver antenna 9 is radiated in the direction of the satellite repeater at a frequency of ω ₁ = ω _pr1 .

по часам второго пункта В с помощью такой же кодовой последовательности M(t) формируют такой же шумоподобный СВЧ-сигнал (сигнал β₁). Регистрируют его на втором пункте В (сигнал β₁, который однако не отправляют на ретрансляцию). Принимают бортовой аппаратурой ИСЗ-ретранслятора на частоте ω₁ (сигнал α₁), переизлучают его на пункты А и В на частоте ω₂ с сохранением фазовых соотношений на интервале t_c.At that moment in time $$t_{\mathrm{one}}^A=t_{\mathrm{one}}^B$$

by the clock of the second point B using the same code sequence M (t) form the same noise-like microwave signal (signal β ₁ ). Register it at the second point (signal β ₁ , which, however, is not sent for relay). Accept on-board equipment of the satellite repeater at a frequency of ω ₁ (signal α ₁ ), re-emit it to points A and B at a frequency of ω ₂ while maintaining phase relationships in the interval t _c .

Retransmitted signal (α ₂₎ at the frequency ω ₂

u ₂ (t) = U ₂ cos [ω ₂ t + φ _k (t) + φ ₂ ], 0≤t≤T _c ,

is received by the transceiver antenna 9 and through the duplexer 8 and the power amplifier 12 is supplied to the first inputs of the second 13 and third 19 mixers and multiplier 23. The second inputs of the second local oscillator 2.2 are supplied to the second inputs of the mixers 13 and 19:

u _Г2 (t) = U _Г2 (ω _Г2 t + φ _Г2 ),

u _Г3 (t) = U _Г2 cos (ω _Г2 t + φ _Г2 + 90 °).

Moreover, the frequencies ω _G1 and ω _{G2 of the} first 2.1 and second 2.2 local oscillators are spaced at the second intermediate frequency

w _{G1 G2} -ω = ω _WP2

At the outputs of the mixers 13 and 19, voltages of combination frequencies are generated. Amplifiers 14 and 20 are allocated voltage of the second intermediate (differential) frequency:

u _CR2 (t) = U _CR2 cos [ω _CR2 (t) -φ _K1 (t) + φ _CR2 ],

u _CR3 (t) = U _CR2 cos [ω _CR2 (t) -φ _K1 (t) + φ _CR2 + 90 °], 0≤t≤T _c ,

;Where $${\mathrm{<figure-callout\; id="2"\; label="U\; P\; R"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">U\; </figure-callout>}}_{PR}=\frac{\mathrm{one}}{2}{K}_{\mathrm{one}}{U}_2{\mathrm{<figure-callout\; id="2"\; label="U\; G"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">U\; </figure-callout>}}_G$$

;

ω _CR2 = ω _G2 -ω ₂ - the second intermediate (difference) frequency;

_np2 φ = φ ₂ -φ _T2.

The voltage u _pr3 (t) from the input of the amplifier 20 of the second intermediate frequency is supplied to the input of the phase shifter 21 by -90 °, at the output of which a voltage is generated

u _CR4 (t) = U _CR2 cos [ω _CR2 t-φ _k (t) + φ _CR2 + 90 ° -90 °] = U _CR2 cos [ω _CR2 t-φ _k (t) + ω _CR2 ], 0≤ t≤

Voltages u _pr2 (t) and u _pr4 (t) from the output of amplifier 14 and phase shifter 21 by -90 ° are supplied to two inputs of the first adder 22, at the output of which the first total voltage is formed

u _Σ1 (t) = U _Σ1 cos [ω _CR2 t-φ _k1 (t) + φ _CR2 ], 0≤t≤T _c ,

where U _Σ1 = 2U _pr2 ,

which is fed to the second input of the multiplier 23. At the output of the latter, a harmonic voltage is generated

u ₁ (t) = U ₁ cos (ω _Г2 t + φ _Г2 ), 0≤t≤T _c ,

;Where $$U_{\mathrm{one}}=\frac{\mathrm{one}}{2}{K}_2{U}_2{\mathrm{<figure-callout\; id="2"\; label="U\; P\; R"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">U\; </figure-callout>}}_{PR}$$

;

K ₂ - transfer coefficient of the multiplier,

which is allocated by a narrow-band filter 24 (tuning frequency ω _n which is chosen equal to the frequency of the second local oscillator 2.2 ω _n = ω _G2 ), is detected by the amplitude detector 25 and is fed to the control input of the key 26, opening it. In the initial state, the key 26 is always closed.

The voltage u _Σ (t) from the output of the adder 22 through the public key 26 is supplied to the input of the clipper 15, where it is clipped and recorded in the buffer memory 16. The registration is synchronized by standard 1 frequency and time.

In the second step (when transmitting the signal from point B), the switch 4 must be open and the signal α ₃ from the generator 3 through the clipper 10 enters the same memory 11. The relay signal α _{4 is} recorded, like α ₂ , in the memory 16.

по часам второго пункта аналогично формируют и регистрируют шумоподобный СВЧ-сигнал (сигнал β₃). Сформированный сигнал преобразуют на частоту ω₁, усиливают его по мощности, излучают усиленный сигнал в направлении того же ИСЗ-ретранслятора.At an arbitrary point in time $$t_3^B=t_2^B+\Theta$$

by the hours of the second point, a noise-like microwave signal (signal β ₃ ) is similarly generated and recorded. The generated signal is converted to a frequency ω ₁ , amplified by power, emitted amplified signal in the direction of the same satellite repeater.

по часам первого пункта А с помощью той же кодовой последовательности формируют такой же шумоподобный СВЧ-сигнал (сигнал α₃). Регистрируют его на первом пункте А. Принимают бортовой аппаратурой ИСЗ-ретранслятора сигнал на частоте ω₁ (сигнал α₃), переизлучают его на пункты А и В на частоте ω₂ с сохранением фазовых соотношений, принимают ретранслированный сигнал на обоих пунктах, преобразуют его на видеочастоту, регистрируют в моменты времени t₄ ^A и t₄ ^B соответственно (сигнал α₄, β₄).At that moment in time $$t_3^B=t_3^A$$

by the clock of the first point A using the same code sequence form the same noise-like microwave signal (signal α ₃ ). Register it at the first point A. Accept the on-board equipment of the satellite relay signal at a frequency of ω ₁ (signal α ₃ ), re-emit it to points A and B at a frequency of ω ₂ while maintaining phase relationships, receive a relay signal at both points, convert it to video frequency, recorded at time t ₄ ^A and t ₄ ^B, respectively (signal α ₄ , β ₄ ).

The correlation processing of two pairs of registered signals in the meter 17 determines at each point the following time delays:

, $${\tau }_{\mathrm{one}}={\beta }_{\mathrm{one}}\otimes {\mathrm{<figure-callout\; id="2"\; label="\beta "\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">\beta </figure-callout>}}_2=t_2^B-t_{\mathrm{one}}^B=a_{\mathrm{one}}+b_{\mathrm{one}}+({\Delta }_{\mathrm{AT}}^{\mathrm{AND}}+{\Delta }_{\mathrm{AT}}^P+\Delta S)+\Delta t$$

,

, $${\mathrm{<figure-callout\; id="2"\; label="\tau "\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">\tau </figure-callout>}}_2={\mathrm{<figure-callout\; id="3"\; label="\alpha "\; filenames="00000034.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}_3\otimes {\alpha }_{\mathrm{four}}=t_{\mathrm{four}}^A-t_3^A=a_3+{\mathrm{<figure-callout\; id="2"\; label="b"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">b</figure-callout>}}_2+({\Delta }_{\mathrm{AT}}^{\mathrm{AND}}+{\Delta }_{\mathrm{BUT}}^P+\Delta S)-\Delta t$$

,

, $${\mathrm{<figure-callout\; id="3"\; label="\tau "\; filenames="00000034.png"\; state="\{\{state\}\}">\tau </figure-callout>}}_3={\alpha }_{\mathrm{one}}\otimes {\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}_2=t_2^A-t_{\mathrm{one}}^A=a_{\mathrm{one}}+a_2+({\Delta }_{\mathrm{BUT}}^{\mathrm{AND}}+{\Delta }_{\mathrm{BUT}}^P+\Delta S)$$

,

, $${\tau }_{\mathrm{four}}={\mathrm{<figure-callout\; id="3"\; label="\beta "\; filenames="00000034.png"\; state="\{\{state\}\}">\beta </figure-callout>}}_3\otimes {\beta }_{\mathrm{four}}=t_{\mathrm{four}}^B-t_3^B={\mathrm{<figure-callout\; id="2"\; label="b"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">b</figure-callout>}}_2+{\mathrm{<figure-callout\; id="3"\; label="b"\; filenames="00000034.png"\; state="\{\{state\}\}">b</figure-callout>}}_3+({\Delta }_{\mathrm{AT}}^{\mathrm{AND}}+{\Delta }_{\mathrm{AT}}^P+\Delta S)$$

,

and the corresponding interference frequencies F _i (i = 1, 2, 3, 4), which determine the derivatives of these delays:

$${\dot{\tau }}_i=\frac{d{\tau }_i}{dt}=\frac{F_i}{\overline{f}}$$

,Where $$\overline{f}=\frac{(f_{\mathrm{one}}+f_2)}{2}$$

,

a _j , b _j (j = 1, 2, 3) is the signal propagation time between the satellite and points A and B, respectively (Fig. 1);

, $${\Delta }_{В}^{И}$$

- задержки сигналов в излучающей аппаратуре обоих пунктов; $${\Delta }_{\mathrm{BUT}}^{\mathrm{AND}}$$

, $${\Delta }_{\mathrm{AT}}^{\mathrm{AND}}$$

- signal delays in the radiating equipment of both points;

, $${\Delta }_{В}^{П}$$

- задержки сигналов в приемно-регистрирующей аппаратуре; $${\Delta }_{\mathrm{BUT}}^P$$

, $${\Delta }_{\mathrm{AT}}^P$$

- signal delays in receiving and recording equipment;

ΔS - signal delay in the onboard satellite repeater;

- искомая разность показаний часов в один и тот же физический момент. $$\Delta t=t_{\mathrm{one}}^B-t_{\mathrm{one}}^A$$

- the desired difference in the clock at the same physical moment.

, $$b_j=\dot{b}$$

, получаем:Assuming a _j and b _j linear functions with derivatives $$a_j=\dot{a}$$

, $$b_j=\dot{b}$$

we get:

, $$\Delta t=\frac{\mathrm{one}}{2}({\tau }_{\mathrm{one}}-{\tau }_2)+\gamma +\delta +\epsilon +\nu$$

,

Where

, $$\gamma =\frac{\mathrm{one}}{2}(f+2b+\Theta )\dot{a}+\frac{\mathrm{one}}{2}(b+\Theta )\dot{b}$$

,

, $$\delta =\frac{\mathrm{one}}{2}({\Delta }_{\mathrm{BUT}}{}^P-{\Delta }_{\mathrm{AT}}{}^P)-\frac{\mathrm{one}}{2}({\Delta }_{\mathrm{BUT}}{}^{\mathrm{AND}}-{\Delta }_{\mathrm{AT}}{}^{\mathrm{AND}})$$

,

$$\epsilon =\frac{\mathrm{one}}{2}({\Delta }_{\mathrm{BUT}}"-{\Delta }_A\text{'}\text{'})-\frac{\mathrm{one}}{2}({\Delta }_B"-{\Delta }_B\text{'}\text{'})$$

, $$\nu =2\mathrm{<figure-callout\; id="2"\; label="\omega \; D\; c"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">\omega \; </figure-callout>}\frac{D}{c^{}}\frac{{}^2}{}$$

,

, $${\Delta }_{А,B}^{"}$$

- задержки сигнала в атмосфере на частотах ω₁ и ω₂ соответственно; $${\Delta }_{\mathrm{BUT},B}^{\text{'}\text{'}}$$

, $${\Delta }_{\mathrm{BUT},B}^{"}$$

- signal delays in the atmosphere at frequencies ω ₁ and ω _2, respectively;

ν - relativistic correction (Sagnac effect);

ω is the angular velocity of the Earth;

c is the speed of light;

D is the area of the quadrangle OA'S'B ', formed in the equatorial plane by the center of mass of the Earth, the projections of points A, B and the satellite S.

Corrections γ on the mobility of the satellite repeater during a single measurement is most easily reduced to zero by the corresponding choice of the free parameter Θ:

, $${\Theta }_o=-\left[\dot{a}\frac{(a+b)}{(\dot{a}+\dot{b})}+b\right]\approx -\frac{\mathrm{one}}{2}\left[{\dot{\tau }}_3\frac{({\tau }_{\mathrm{one}}+{\tau }_2)}{({\dot{\tau }}_{\mathrm{one}}+{\dot{\tau }}_2)}+{\dot{\tau }}_{\mathrm{four}}\right]$$

,

which should be calculated at the beginning of measurements by approximate ephemeris data, and then clarified by the results of current measurements.

As for the correction δ for hardware delays, it can be found by calibration using the “zero base” method.

The atmospheric correction ε is also taken into account.

At point B, the equipment works similarly, only the order of steps there is the opposite. To calculate the difference between the clock readings Δt, it is now sufficient to exchange the received digital data between the points, which can be done via ordinary telephone or telegraph communication channels.

The above operation of the device that implements the proposed method, corresponds to the reception of useful signals on the main channel at a frequency of ω ₂ (figure 4).

If the noise-like signal is received by the image channel at frequency ω _H

u _З (t) = U _З cos [ω _З t + φ _K2 (t) + φ _З ], 0≤t≤T _З ,

the amplifiers 14 and 20 of the second intermediate frequency are allocated the following voltages:

u _PR5 (t) = U _PR5 cos [ω _PR2 t + φ _К2 (t) + φ _pr5 ],

u _PR6 (t) = U _PR5 cos [ω _PR2 t + φ _К2 (t) + φ _PR5 -90 °], 0≤t≤T _З,

;Where $$U_{PR5}=\frac{\mathrm{one}}{2}{K}_{\mathrm{one}}{U}_3{\mathrm{<figure-callout\; id="2"\; label="U\; G"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">U\; </figure-callout>}}_G$$

;

ω _PR2 = ω _З -ω _G2 - the second intermediate (difference) frequency;

φ _ПP5 = φ _З -φ _Г2 .

The voltage u _PR6 (t) from the output of the amplifier 20 of the second intermediate frequency is supplied to the inputs of the phase shifters 21 by -90 ° and 27 by + 90 °, at the output of which the following voltages are generated:

_pr7 u (t) = U _np5 cos (ω t + φ _{np2 np5} -90-90 °) = - U _np5 cos (ω t + φ _{np2 np5),} 0≤t≤T _W.

The voltage u _CR5 (t) and u _CR7 (t) are supplied to the two inputs of the adder 22, compensated at its output.

Consequently, a false signal (interference) of the received image channel at frequency ω _W is suppressed.

For a similar reason, a false signal (interference) received on the second combination channel at a frequency of ω _{k2 is also} suppressed.

If the false signal (interference) is received on the first combinational channel at a frequency ω _k1

u _k1 = U _k1 cos (ω _k1 t + φ _k1 ), 0≤t≤T _k1 ,

then the amplifiers 14 and 20 of the second intermediate frequency are allocated the following

voltage:

u _CR8 (t) = U _CR2 cos (ω _CR2 t + φ _CR8 ),

u _PR9 (t) = U _CR8 cos (ω _CR2 t + φ _CR8 + 90 °), 0≤t≤T _k1 ,

;Where $$U_{PR9}=\frac{\mathrm{one}}{2}{K}_{\mathrm{one}}{U}_{k\mathrm{one}}{\mathrm{<figure-callout\; id="2"\; label="U\; G"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">U\; </figure-callout>}}_G$$

;

_np2 ω = 2ω _T2 -ω _k1 - second intermediate (difference) frequency;

φ _pr8 = φ _Г2 -φ _k1 .

The voltage u _pr9 (t) from the output of the amplifier 20 of the second intermediate frequency is supplied to the input of the phase shifter 21 by -90 °, at the output of which a voltage is generated

u _pr10 (t) = U _pr8 cos (ω _pr2 t + 90 ° -90 ⁹ ) = U _pr8 cos (ω _pr2 t + φ _pr8 ), 0≤t≤T _k1 .

_Pr8 voltage u (t) and _pr1 u (t) are input to the adder 22, the output of which is formed following voltage:

u _Σ (t) = U _Σ1 cos (ω _CR1 t + φ _CR8 ), 0≤t≤T _k1

where U _Σ1 = 2U _pr8 .

This voltage is supplied to the second input of the multiplier 23, the output of which produces the following harmonic voltage:

u ₂ (t) = U ₂ cos (2ω _Г2 t + φ _Г2 ), 0≤t≤T _k1 ,

.Where $${\mathrm{<figure-callout\; id="2"\; label="U"\; filenames="00000033.png,00000034.png"\; state="\{\{state\}\}">U</figure-callout>}}_2\frac{\mathrm{one}}{2}{K}_2{U}_{k2}{U}_{\Sigma \mathrm{one}}$$

.

This voltage does not fall into the passband of the narrow-band filter 24. Therefore, a false signal (interference) received on the first combinational channel at a frequency ω _k1 is suppressed.

To completely suppress false signals (interference) received through the mirror channel at the frequency ω _З , a complex (amplitude-phase) identification system is used, consisting of a calibrator 27, adjustable phase shifters 28 and 29, narrow-band filters 30 and 31, amplitude detectors 32 and 34 , a subtraction unit 35, two inverse amplifiers 36 and 39, a multiplier 37, low pass filters 35 and 38.

Full suppression of false signals (interference) is possible only with the identity of the receiving channels. However, the real amplifiers 14 and 20 of the second intermediate frequency, which are part of the receiving channels, have different characteristics. Differences increase due to other elements that are part of the receiving channels.

Complex (amplitude and phase) identification system utilizes harmonic calibration signal obtained from a separate generator (calibrator) 27, the frequency ω _to which it differs from the second intermediate frequency ω _np2 by an amount Δω (Δω = ω _to -ω _np2) (4 ) With a small value of Δω, the calibration signal carries information about the identity of the receiving channels. At the second intermediate frequency ω _CR2 due to the correlation of close values of the frequency characteristics.

The inputs of the first 14 and second 20 amplifiers of the second intermediate frequency through the adjustable phase shifters 28 and 29, respectively, from the output of the calibrator 27 receives a harmonic calibration signal

_{u k (t) = U k} cos (ω k t + φ k), 0≤t≤T k,

the frequency ω _to which differs from the second intermediate frequency (ω _CR2 by a small amount Δω (Fig. 4). At the output of the amplifiers 14 and 20 of the second intermediate frequency, calibration signals are extracted by narrow-band filters 30, 31 and after detection in the amplitude detectors 32, 33 are fed to 34 inputs the amplitude subtraction unit identification system. When receiving channels inequality transmission modules coefficients (K ₁ = K ₂₎ at the frequency ω _to the output of the subtractor 34 appears voltage (positive or negative) which, through folder 35 lowpass and inverse amplifier 36 acts on the control inputs of amplifiers 14 and 20, the second intermediate frequency by changing their transmission K ₁ and K ₂ in such ratios that the output voltage of the subtractor 34 converges to zero. In this case, the gains of amplifiers 14 and 20 of the second intermediate frequencies turn out to be almost identical at the frequency ω _{to the} calibration signal (K ₁ = K ₂ = K).

From the outputs of the narrow-band filters 30 and 31, the calibration signals are supplied to a phase identification system consisting of a multiplier 37, a low-pass filter 38, an inverse amplifier 39, and two adjustable phase shifters 28 and 29.

In the presence of phase non-identity of the receiving channels at the output of the phase detector, consisting of a multiplier 37 and a low-pass filter 38, a voltage is generated (positive or negative), which through the inverse amplifier 39 acts on the control inputs of the adjustable phase shifters 28 and 29, changing the phase shifts of the calibration signals so that the output voltage of the phase detector tends to zero. Thus, the phase identification of the receiving channels is achieved.

The presence of a strong correlation between the modules of the transmission coefficients and between their arguments at frequencies ω _pr2 and ω _k allows us to state the practical equality of the modules of the transmission coefficients and the equality of their arguments at the second intermediate frequency ω _pr2 .

The clock synchronization method allows you to:

- achieve extreme measurement accuracy (about 0.1 ns) using VLBI technology and relay technology, which is already widely used in practice;

- generate the necessary microwave signals for measurements at ground stations, which makes it possible to gradually increase the accuracy of measurements by optimizing the signal structure and improving the ground-based recording technique without interfering with the satellite onboard equipment;

- increase the efficiency of measurements, i.e. bring the time interval from the beginning of measurements to obtain results up to several tens of seconds (almost to the time of correlation signal processing);

- to avoid the installation on board of a satellite of highly stable time-keepers and time interval meters, to limit the on-board equipment to only a phase-stable microwave signal relay system.

Thus, the proposed method and device in comparison with prototypes provide increased noise immunity and accuracy of synchronization of remote time scales. This is achieved by completely suppressing false signals (interference) received through the mirror channel at a frequency of ω ₃ through the use of a complex (amplitude-phase) identification system of the receiving channels. The specified system uses a harmonic calibration signal received from a separate generator (calibrator), the frequency of which differs from the second intermediate frequency ω _AC2 by a certain value Δω (Δω = ω _{to -ω AC2} ). With a small value of Δω, the calibration signal carries information about the non-identity of the receiving channels at the second intermediate frequency ω _CR2 due to the correlation of close frequency characteristics.

Claims (2)

1. A clock synchronization method based on the simultaneous reception of noise-like microwave signals from an artificial Earth satellite by spaced ground points, their coherent conversion to a video frequency, digital recording of received signals and determining the time delay of the arrival of the same signal to synchronization points by correlation processing of recorded signals for which the magnitude comparison of time scales produce, while at the initial time t ₁ to the clock of the first item using the code pos edovatelnosti form a noise-like microwave signal is recorded it at the same point, the conditioned signal is converted to a frequency ω _1, increase its power emit the amplified signal in the direction of artificial Earth relay satellite in the same time t _1, the clock of the second paragraph using the same code sequence, the same noise-like microwave signal is generated, recorded at the second point, the signal at frequency ω _{1 is} received by the onboard equipment of the artificial Earth-relay satellite, and re-emitted to the first and second points at a frequency of ω ₂ while maintaining phase relationships, at an arbitrary time t ₃ according to the clock of the second point, a noise-like microwave signal is generated and recorded in a similar way, the generated signal is converted to a frequency of ω ₁ , amplified by power, emitted amplified signal in the direction of the same artificial satellite repeater earth, in the same moment of time t ₃ by the clock of the first item by using the same code sequence form the same noise-like microwave signal, it is recorded in the first paragraph, I agree onboard equipment artificial satellite repeater signal at frequency ω ₁ and re-emit it at first and second points at the frequency ω ₂ while preserving phase relationships, received at frequency ω ₂ is converted by frequency in the two receiving channels with a first receiving channel voltage of the second heterodyne and in the second receiving channel, the voltage of the second local oscillator phase-shifted by + 90 ° is isolated in the first and second receiving channels of the voltage of the second intermediate frequency, the second voltage is shifted in phase by -90 ° omezhutochnoy frequency of the second reception channel, summing it with the voltage of the second intermediate frequency of the first reception channel, multiply the resulting sum voltage with the received signal, allocate the harmonic voltage at frequency ω _r2 of the second local oscillator is detected it and is used to permit further processing of the received signal, characterized in that to eliminate the non-identity of the receiving channels, a complex (amplitude-phase) identification system is introduced, in which a harmonic calibration a signal whose frequency ω _to which differs from the second intermediate frequency ω _CR2 by an insignificant value Δω = ω _{to -ω CR2} , passes a harmonic calibration signal through the first and second receiving channels, selects harmonic calibration signals at their outputs, detects them and feeds them to two inputs a subtraction unit, at the output of which a low-frequency voltage of positive or negative polarity is formed, they affect the control inputs of the amplifiers of the second intermediate frequency of the first and second receiving channels, changing x transmission coefficients so that the output voltage of the subtraction unit tends to zero, realizing the amplitude identity of the receiving channels, at the same time the harmonic calibration signals extracted in the first and second receiving channels are multiplied among themselves, a low-frequency voltage of positive or negative polarity is allocated, they affect the control inputs of adjustable phase shifters installed in the first and second receiving channels, changing the phase shifts of the calibration signals so that the output low-frequency voltage tended to zero, realizing the phase identity of the receiving channels. 2. A clock synchronization device containing an artificial Earth satellite and two spaced ground stations, each of which contains a frequency and time standard in series, a pseudo-noise signal generator, a switch, a first mixer, the second input of which is connected through the first local oscillator to the second output of the frequency and time standard , an amplifier of the first intermediate frequency, a first power amplifier, a duplexer, the input-output of which is connected to a transceiver antenna, a second power amplifier and a second mixer, the second input of which through the second local oscillator is connected to the second output of the frequency and time standard, the first clipper is connected in series to the second output of the pseudo noise signal generator, the second input of which is connected to the third output of the frequency and time standard, the first buffer storage device and the delay meter and their derivatives, output which is the output of the device, the first amplifier of the second intermediate frequency is connected in series, an adder, the second input of which is connected to the -90 ° phase shifter the output of the second amplifier of the second intermediate frequency, the first multiplier, the second input of which is connected to the output of the second power amplifier, the first narrow-band filter, the first amplitude detector, the key, the second input of which is connected to the output of the adder, the second clipper, the second input of which is connected to the third output of the frequency standard and time, and a second buffer storage device, the output of which is connected to the second input of the delay meter and their derivatives, connected in series to the output of the second local phase oscillator +90 and a third mixer, the second input of which is connected to the output of the second power amplifier, characterized in that it is equipped with a calibrator, two adjustable phase shifters, two inverse amplifiers, two low-pass filters, a subtraction unit, a second and third narrow-band filters, the second and third amplitude detectors, and a second narrow-band filter, a second amplitude detector, a subtraction unit, the second input of which is connected to the output of the first amplifier of the second intermediate frequency a third series-pass filter and a third amplitude detector connected in series with the output of the second amplifier of the second intermediate frequency, the third low-pass filter and the first inverse amplifier, the two outputs of which are connected to the control inputs of the first and second amplifiers of the second intermediate frequency, respectively, to the output of the second narrow-band filter in series a second multiplier is connected, the second input of which is connected to the output of the third narrow-band filter, a second low-pass filter and a second second inverse amplifier, whose two outputs are connected to control inputs of the first and second variable phase shifters, each of which is connected between the mixer output and the input of the amplifier of the second intermediate frequency corresponding to the reception channel, the second inputs of the first and second variable phase shifters connected to the outlet of the calibrator.

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