RU2507529C1 - Radio navigation system for measurement of mobile object bearing - Google Patents

Abstract

FIELD: instrument making.

SUBSTANCE: radio beacon simultaneously from two points spatially distanced in the plane of measurements with available coordinates radiates orthogonally linearly polarised electromagnetic waves. Electromagnetic waves are received on a mobile object in a linear polarisation basis making the angle of 45° with the plane of measurements. According to the signals received at the outlet of the linear polarisation divider a summary and a difference signals are generated, and phase difference is measured between them, afterwards an angular coordinate of a mobile object is calculated.

EFFECT: proposed goniometrical system provides for better efficiency and accuracy of measurements with availability of strict restrictions for dimensions of a receiving antenna of a mobile object, where mass and dimensions of an antenna are of utmost importance.

2 dwg

Description

The invention relates to radio navigation and can be used in navigation systems to determine the angular coordinates of moving objects in both azimuthal and elevation planes.

A known radio navigation system [1, 2], in which the bearing of a moving object is determined on the basis of measuring the phase difference of the signals at the output of two spaced receiving antennas. This system contains a source of electromagnetic waves located at a point with known coordinates and located on a moving object, two receiving antennas spaced in space, each of which is connected in series with a corresponding receiver, a phase meter and a computing device, and the outputs of the receivers are connected to the corresponding two inputs phase meter, and its output is connected to the input of the computing device. The phasometer measures the phase difference of the voltages at the output of the receivers, and the computing device determines the bearing of the moving object from the measured phase difference.

The disadvantage of this radio navigation system is the low accuracy of the measurements of the bearing of a moving object in the presence of severe restrictions on the dimensions of the receiving antennas.

This disadvantage is due to the fact that the measurement accuracy of the bearing of a moving object is determined by the spatial separation of the receiving antennas. The greater the distance between the antennas, the higher the measurement accuracy and, conversely, with a decrease in the spatial separation of the receiving antennas, the measurement accuracy of the bearing of a moving object decreases.

Known radio navigation system for measuring the bearing of a moving object [3, 4]. This system contains a beacon located at a point with known coordinates and a receiving indicator located on a moving object. The beacon contains a transmitter with three transmitting antennas connected to it, through a synchronous switch, located at the vertices of an equilateral triangle. The phase difference of the signals coming from any of the pairs of antennas is determined by the angular position of the moving object. The receiving indicator contains a receiving antenna, the output of which is connected to the output of the receiver, and its output through a synchronous switch is connected to three receiving channels and a phase meter, two inputs of which are connected to the outputs of any of the pairs of receiving channels by a switch. The operation of the system is based on alternating, in time, radiation of electromagnetic waves from two points with known coordinates located in the measurement plane at a distance d from each other. Moreover, the lengths and amplitudes of the emitted electromagnetic waves are equal and the initial phases coincide. On a moving object, electromagnetic waves from any of the pairs of transmitting antennas are received sequentially in time and their phase difference is measured, after which the bearing of the moving object is calculated.

The disadvantage of this radio navigation system is the low speed of the measurements of the bearing of a moving object, due to the time separation of the transmitted and, accordingly, received signals.

A known radio navigation system for measuring the bearing of a moving object (AS No. 1355955, M class ⁴ , G01S 3/02, priority dated 12/12/1985 [5]), in which the bearing of a moving object is determined based on measuring the phase difference Δφ between orthogonally linearly polarized electromagnetic waves simultaneously emitted with equal amplitudes, phases and wavelengths from two points with known coordinates located in the measurement plane at a distance d from each other. In this case, the bearing α of the moving object is determined relative to the equal signal direction, which coincides with the normal to the middle of the line connecting the emission points of the orthogonally linearly polarized electromagnetic waves according to the formula:

$$\alpha =\arcsin \left(\frac{\mathrm{<figure-callout\; id="2"\; label="\lambda "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\lambda </figure-callout>}}{2\pi d}\cdot \Delta \varphi \right)\text{(one)}$$

where λ is the wavelength.

The navigation system comprises a transmitter with two transmitting antennas connected to it with orthogonal linear eigenpolarizations. On a moving object, the navigation system contains a receiving all-polarized antenna, a circular waveguide section with a built-in quarter-wave phase plate, a linear polarizing separator, an amplitude discriminator and a computer. Moreover, the quarter-wave phase plate is oriented at an angle of 45 ° to one of the walls of a rectangular waveguide of a linear polarizing separator.

The navigation system operates as follows.

The transmitter through the two transmitting antennas connected to it with orthogonal linear intrinsic polarizations emit linearly orthogonally polarized electromagnetic waves with equal amplitudes, phases and wavelengths.

At a moving object, the total electromagnetic wave is completely received by the all-polarized receiving antenna and arrives at the serially connected section of the circular waveguide with an integrated quarter-wave phase plate and a linear polarizing separator. The combination of a circular waveguide section with an integrated quarter-wave phase plate and a linear polarizing separator allows the reception of a total electromagnetic wave in a circular polarizing basis on a moving object and, thus, splitting the total electromagnetic wave arriving at the input into two waves orthogonally polarized in a circle. From the outputs of the arms of the linear polarizing separator, the signals are fed to the input of the amplitude discriminator, where a voltage is formed equal to the ratio of the amplitudes E ₁ / E _{2 of the} signals in the form [5]

$$S\left(\alpha \right)=\frac{E_{\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="E"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">E</figure-callout>}}_2}=\frac{\sqrt{\mathrm{one}-\cos \Delta \varphi }}{\sqrt{\mathrm{one}+\cos \Delta \varphi }}=|\; |\; |t\mathrm{<figure-callout\; id="2"\; label="g\; \Delta \; \phi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">g\; </figure-callout>}\frac{\Delta \varphi }{}\frac{}{2}|\; |\; |.\left(\text{2}\right)$$

After that, the output signal (2) of the amplitude discriminator is fed to the computer, where, taking into account (1) and (2), the bearing of the moving object is calculated according to the formula [5]

$$\alpha =\arcsin \left[\frac{\mathrm{<figure-callout\; id="2"\; label="\lambda "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\lambda </figure-callout>}}{2\pi d}\cdot \left(2arctg\frac{E_{\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="E"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">E</figure-callout>}}_2}\pm 2n\pi \right)\right],\left(\text{3}\right)$$

where n = 0, 1, 2, ....

This radio navigation system has a number of disadvantages. Firstly, it has low measurement accuracy of the bearing of a moving object in an equal-signal direction and directions close to it, due to the low steepness of the direction-finding characteristic (2) in these directions for a fixed ratio d / λ. Secondly, it is not possible to determine the side of the deviation of the moving object from the equal-signal direction. The latter is due to the fact that the ratio of the amplitudes of the signals E ₁ / E ₂ at the output of the amplitude discriminator is always positive, therefore the direction-finding characteristic (2) has a symmetrical shape with respect to the equal-signal direction.

The closest set of features to the claimed radio navigation system is a device for measuring the bearing of a moving object (USSR patent No. 1251003 M. class. ⁴ G01S, 3/02, priority from 01/29/85) [6]. This device contains a transmitter with two transmitting antennas connected to it with orthogonal proper polarizations and located at points with known coordinates in the measurement plane at a distance d from each other and located on a moving object, a receiving all-polarized antenna, a linear polarizing separator, an amplitude-phase discriminator and a computer while the output of the receiving all-polarized antenna is connected to the input of the linear polarizing separator, and two of its outputs are connected to two the inputs of the amplitude-phase discriminator, and its output is connected to the input of the calculator, and the linear polarization separator is oriented so that the unit vectors of its own coordinate system coincide with the axes of the arms of the rectangular waveguides and are oriented at an angle θ = 45 ° to the measurement plane.

The operation of the device is that the transmitter through the two transmitting antennas connected to it simultaneously from two points emit orthogonally elliptically polarized electromagnetic waves with equal amplitudes, phases and wavelengths.

On a moving object, the total electromagnetic wave, whose Jones vector in the Cartesian polarization basis has the form [6]

$${\overrightarrow{\dot{E}}}_{\Sigma }=\left[\begin{array}{c}\cos \epsilon +j\sin \epsilon e^{j\Delta \varphi }\\ j\sin \epsilon +\cos \epsilon e^{j\Delta \varphi }\end{array}\right],\left(\text{four}\right)$$

where ε is the angle of ellipticity of the emitted orthogonally polarized electromagnetic waves,

- фазовый сдвиг между ортогонально эллиптически поляризованными электромагнитными волнами в точке приема в направлении α, (λ - длина волны), $$\Delta \varphi =\frac{2\pi d}{\lambda }\sin \alpha$$

- phase shift between orthogonally elliptically polarized electromagnetic waves at the receiving point in the direction α, (λ is the wavelength),

fully received by the all-polarized receiving antenna, after which the signal is fed to the input of a linear polarizing separator, the unit vectors of its own coordinate system coincide with the axes of the arms of the rectangular waveguides and are oriented at an angle θ = 45 ° to the measurement plane. A linear polarizing separator divides the total electromagnetic wave arriving at its input into two linear electromagnetic waves orthogonal in polarization. In this case, the signals at the outputs of the arms of the linear polarizing separator, omitting the time dependence of the signals, are determined using transformations [6]

, $${\dot{E}}_{\mathrm{one}}=\left[\begin{array}{cc}\mathrm{one}& 0\\ 0& 0\end{array}\right]\left[\begin{array}{cc}\cos \theta & -\sin \theta \\ \sin \theta & \cos \theta \end{array}\right]\left[\begin{array}{c}\cos \epsilon +j\sin \epsilon e^{j\Delta \varphi }\\ j\sin \epsilon +\cos \epsilon e^{j\Delta \varphi }\end{array}\right]$$

,

, $${\dot{E}}_2=\left[\begin{array}{cc}0& 0\\ 0& \mathrm{one}\end{array}\right]\left[\begin{array}{cc}\cos \theta & -\sin \theta \\ \sin \theta & \cos \theta \end{array}\right]\left[\begin{array}{c}\cos \epsilon +j\sin \epsilon e^{j\Delta \varphi }\\ j\sin \epsilon +\cos \epsilon e^{j\Delta \varphi }\end{array}\right]$$

,

- оператор перехода из декартового поляризационного базиса, в котором записаны векторы Джонса излучаемых волн, в систему координат поляризационного разделителя;Where $$\left[\begin{array}{cc}\cos \theta & -\sin \theta \\ \sin \theta & \cos \theta \end{array}\right]$$

- the transition operator from the Cartesian polarization basis, in which the Jones vectors of the emitted waves are recorded, into the coordinate system of the polarization separator;

- оператор поляризатора первого плеча линейного поляризационного разделителя; $$\left[\begin{array}{cc}\mathrm{one}& 0\\ 0& 0\end{array}\right]$$

- polarizer operator of the first shoulder of the linear polarizing separator;

- оператор поляризатора второго плеча линейного поляризационного разделителя. $$\left[\begin{array}{cc}0& 0\\ 0& \mathrm{one}\end{array}\right]$$

- polarizer operator of the second shoulder of the linear polarizing separator.

и $${\dot{E}}_2$$

на выходе линейного поляризационного разделителя вида:After the transformations, we obtain analytical expressions for the signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the output of a linear polarizing separator of the form:

$${\dot{E}}_{\mathrm{one}}=\cos \theta \cos \epsilon +j\cos \theta \sin \epsilon e^{j\Delta \varphi }-j\sin \theta \sin \epsilon -\sin \theta \cos \epsilon e^{j\Delta \varphi },\left(\text{5}\right)$$

$${\dot{E}}_2=\sin \theta \cos \epsilon +j\sin \theta \sin \epsilon e^{j\Delta \varphi }+j\cos \theta \sin \epsilon +\cos \theta \cos \epsilon e^{j\Delta \varphi },\left(\text{6}\right)$$

поступает на вход разностного канала, а сигнал $${\dot{E}}_2$$

поступает на вход суммарного канала амплитудно-фазового дискриминатора. Амплитуды A₁ и А₂ сигналов $${\dot{E}}_1$$

и $${\dot{E}}_2$$

на входе амплитудно-фазового дискриминатора имеют вид [6]From the outputs of the arms of the linear polarizing separator, the signals described by the analytical expressions (5) and (6) are fed to the inputs of the amplitude-phase discriminator. Moreover, the signal $${\dot{E}}_{\mathrm{one}}$$

arrives at the input of the difference channel, and the signal $${\dot{E}}_2$$

arrives at the input of the total channel of the amplitude-phase discriminator. Amplitudes A ₁ and A ₂ signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the input of the amplitude-phase discriminator have the form [6]

$$A_{\mathrm{one}}=\sqrt{\mathrm{one}-\mathrm{<figure-callout\; id="2"\; label="sin"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">sin</figure-callout>}2\mathrm{<figure-callout\; id="2"\; label="\epsilon \; cos"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\epsilon \; </figure-callout>}\cos 2\theta \sin \Delta \varphi -\mathrm{<figure-callout\; id="2"\; label="sin"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">sin</figure-callout>}2\theta \cos \Delta \varphi },\left(\text{7}\right)$$

$$A_2=\sqrt{\mathrm{one}+\sin 2\mathrm{<figure-callout\; id="2"\; label="\epsilon \; cos"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\epsilon \; </figure-callout>}\cos 2\theta \sin \Delta \varphi +\mathrm{<figure-callout\; id="2"\; label="sin"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">sin</figure-callout>}2\theta \cos \Delta \varphi }.\left(\text{8}\right)$$

и $${\dot{E}}_2$$

на входе амплитудно-фазового дискриминатора зависят не только от измеряемого параметра разности фаз Δφ, но и от угла эллиптичности ε излучаемых электромагнитных волн и от угла ориентации θ собственной системы координат линейного поляризационного разделителя относительно плоскости измерений. При θ=45° амплитуды A₁ и A₂, а также фазы ψ₁ и ψ₂ сигналов $${\dot{E}}_1$$

и $${\dot{E}}_2$$

на входе амплитудно-фазового дискриминатора имеют вид [6]From the analysis of (7) and (8) it can be seen that the amplitudes A ₁ and A _{2 of the} signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the input of the amplitude-phase discriminator, they depend not only on the measured parameter of the phase difference Δφ, but also on the ellipticity angle ε of the emitted electromagnetic waves and on the orientation angle θ of the intrinsic coordinate system of the linear polarizing separator relative to the measurement plane. At θ = 45 °, the amplitudes A ₁ and A ₂ , as well as the phases ψ ₁ and ψ _{2 of the} signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the input of the amplitude-phase discriminator have the form [6]

$$A_{\mathrm{one}}=\sqrt{\mathrm{one}-\cos \Delta \varphi },\left(\text{9}\right)$$

$${\psi }_{\mathrm{one}}=\frac{\pi }{2}-\frac{2\epsilon -\mathrm{<figure-callout\; id="2"\; label="\Delta \; \phi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\Delta \; </figure-callout>}\varphi }{2},\left(\text{eleven}\right)$$

$${\mathrm{<figure-callout\; id="2"\; label="A"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">A</figure-callout>}}_2=\sqrt{\mathrm{one}+\cos \Delta \varphi },\left(\text{10}\right)$$

$${\mathrm{<figure-callout\; id="2"\; label="\psi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_2=\frac{2\epsilon +\mathrm{<figure-callout\; id="2"\; label="\Delta \; \phi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\Delta \; </figure-callout>}\varphi }{2},\left(\text{12}\right)$$

and their phase difference has the form

$$\Delta \psi ={\psi }_{\mathrm{one}}-{\mathrm{<figure-callout\; id="2"\; label="\psi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_2=\frac{\pi }{2}-2\epsilon .\left(13\right)$$

и $${\dot{E}}_2$$

на входе амплитудно-фазового дискриминатора постоянна и, наоборот, определяется только углом эллиптичности ε излучаемых ортогонально поляризованных электромагнитных волн и не зависит от измеряемого параметра разности фаз Δφ.Thus, at θ = 45 °, the amplitudes A ₁ and A ₂ depend only on the measured parameter of the phase difference Δφ between the orthogonally elliptically polarized electromagnetic waves received on the moving object from the first and second transmitting antennas and are independent of the ellipticity angle ε of these waves. At the same time, the phase difference Δψ between the signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the input of the amplitude-phase discriminator is constant and, conversely, is determined only by the ellipticity angle ε of the emitted orthogonally polarized electromagnetic waves and is independent of the measured phase difference parameter Δφ.

и $${\dot{E}}_2$$

определялись выражениями (9) и (10). Из анализа (13) видно, что для передающих антенн, излучающих линейно ортогонально поляризованные электромагнитные волны с углом эллиптичности ε=0°, это условие выполняется. В случае, если передающие антенны излучают в общем случае ортогонально эллиптические поляризованные электромагнитные волны с углом эллиптичности ε, то, как следует из (13), их разность фаз отличается от 90° на величину, равную удвоенному углу эллиптичности ε излучаемых электромагнитных волн. В этом случае необходимо излучать электромагнитные волны с равной амплитудой и длиной волны, но с начальной разностью фаз, равной удвоенному углу эллиптичности ε [6].To ensure the normal operation of the amplitude-phase discriminator, it is necessary, as noted in [6], that the phase difference between the signals arriving at its input is equal to 90 °, and the amplitudes A ₁ and A _{2 of the} signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

were determined by expressions (9) and (10). From analysis (13), it is seen that for transmitting antennas emitting linearly orthogonally polarized electromagnetic waves with an ellipticity angle ε = 0 °, this condition is satisfied. If transmitting antennas emit generally orthogonally elliptical polarized electromagnetic waves with an ellipticity angle ε, then, as follows from (13), their phase difference differs from 90 ° by an amount equal to twice the ellipticity angle ε of the emitted electromagnetic waves. In this case, it is necessary to radiate electromagnetic waves with equal amplitude and wavelength, but with an initial phase difference equal to twice the ellipticity angle ε [6].

и $${\dot{E}}_2$$

относительно сигнала $${\dot{E}}_2$$

. В результате на выходе амплитудно-фазового дискриминатора формируется выходное напряжение, знак которого учитывает знак разности фаз Δφ сравниваемых сигналов $${\dot{E}}_1$$

и $${\dot{E}}_2$$

, пропорциональное отношению вида [6]In the amplitude-phase discriminator, frequency conversion and amplification takes place, taking into account the work of automatic gain control, normalizing the signals at an intermediate frequency $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

regarding signal $${\dot{E}}_2$$

. As a result, an output voltage is formed at the output of the amplitude-phase discriminator, the sign of which takes into account the sign of the phase difference Δφ of the compared signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

proportional to the relation of the form [6]

$$S\left(\alpha \right)=k\cdot \frac{A_{\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="A"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">A</figure-callout>}}_2}=k\cdot t\mathrm{<figure-callout\; id="2"\; label="g\; \Delta \; \phi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">g\; </figure-callout>}\frac{\Delta \varphi }{}\frac{}{2},\left(\text{fourteen}\right)$$

where k is the coefficient of proportionality, depending on the identity of the amplitude and phase-frequency characteristics of the receiving channels.

Setting k = 1 from (14) it follows that

$$\Delta \varphi =\pm 2arctg\frac{A_{\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="A"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">A</figure-callout>}}_2}\pm \mathrm{<figure-callout\; id="2"\; label="n"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">n</figure-callout>}2\pi ,\left(\text{fifteen}\right)$$

where n = 0, 1, 2, ....

From the output of the amplitude-phase discriminator, the signal goes to the computer, where, taking into account (1) and (15), the operation of calculating the bearing and the moving object by the formula

$$\alpha =\arcsin \left[\frac{\lambda }{\pi d}\left(\pm arctg\frac{A_{\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="A"\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">A</figure-callout>}}_2}\pm n\pi \right)\right].\left(\text{16}\right)$$

The dependence of the measured ratio A ₁ / A ₂ (14) on the angular coordinate α of the moving object, in fact, is the direction-finding characteristic of the goniometer. Using relation (14) taking into account (1), setting k = 1, we can show that the steepness of the direction-finding characteristic at the point α = 0 is determined by the relation

$$\mu \left(\alpha \right)={|\; |\; |\frac{dS\left(\alpha \right)}{d_{\alpha }}|\; |\; |}_{\alpha =0}={|\; |\; |\frac{d}{d_{\alpha }}tg\left(\frac{\pi d}{\lambda }\sin \alpha \right)|\; |\; |}_{\alpha =0}=\frac{\pi d}{\lambda }.\left(\text{17}\right)$$

.Thus, the steepness of the direction finding characteristic and, consequently, the direction finding accuracy increase with increasing ratio $$d/\lambda$$

.

The disadvantage of this radio navigation system is the low measurement accuracy of the bearing α of a moving object located in the same-signal direction and directions close to the equal-signal one. This disadvantage is due to the low steepness of the direction-finding characteristic in these directions, with a fixed ratio d / λ.

Figure 1 presents the structural electrical diagram of a radio navigation system for measuring bearing of a moving object.

The radionavigation system contains a transmitter 1, transmitting antennas 2 and 3, located at points with known coordinates and spaced apart in the measurement plane at a distance d from each other, on board a moving object, the radionavigation system contains a receiving all-polarized antenna 4, a linear polarizing separator 5, coaxial waveguide transitions 6 and 7, the sum-difference block 8, the phase angular discriminator 9 and the calculator 10.

Figure 2 presents the structural electric diagram of the phase angular discriminator 9, which includes the first frequency mixer 11, the second frequency mixer 12, phase shifter π / 2 13, local oscillator 14, the first intermediate frequency amplifier (IFA) with an amplitude limitation of 15, the second Amplifier with limited amplitude 16, phase detector 17.

The radio navigation system operates as follows.

The transmitter 1, through two transmitting antennas 2 and 3 connected to it, emits electromagnetic waves, respectively, with horizontal and vertical orientations of the plane of polarization with equal amplitudes, phases and wavelengths.

On a moving object, the total electromagnetic wave, the Jones vector of which in the direction α in the Cartesian polarization basis, taking into account (4), subject to ε = 0 °, has the form:

$${\dot{\overrightarrow{E}}}_{\Sigma }=\left[\begin{array}{c}\mathrm{one}\\ e^{j\Delta \varphi }\end{array}\right],\left(\text{eighteen}\right)$$

- фазовый сдвиг между ортогонально линейно поляризованными электромагнитными волнами в точке приема в направлении α, (λ - длина волны),Where $$\Delta \varphi =\frac{2\pi d}{\lambda }\sin \alpha$$

- phase shift between orthogonally linearly polarized electromagnetic waves at the receiving point in the direction α, (λ is the wavelength),

fully accepted by the all-polarized antenna 4 and fed to the input of the linear polarizing separator 5, the unit vectors of its own coordinate system coincide with the axes of the shoulders of rectangular waveguides orthogonally located with respect to each other and are oriented, in general, at an angle θ with the measurement plane.

Then the orthogonally linearly polarized signals at the outputs of the arms of the linear polarizing separator 5, omitting the time dependence of the signals, are determined using transformations of the form:

$${\dot{\overrightarrow{E}}}_{\mathrm{one}}=\left[\begin{array}{cc}\mathrm{one}& 0\\ 0& 0\end{array}\right]\left[\begin{array}{cc}\cos \theta & -\sin \theta \\ \sin \theta & \cos \theta \end{array}\right]\left[\begin{array}{c}\mathrm{one}\\ e^{j\Delta \varphi }\end{array}\right],\left(\text{19}\right)$$

$${\dot{\overrightarrow{E}}}_2=\left[\begin{array}{cc}0& 0\\ 0& \mathrm{one}\end{array}\right]\left[\begin{array}{cc}\cos \theta & -\sin \theta \\ \sin \theta & \cos \theta \end{array}\right]\left[\begin{array}{c}\mathrm{one}\\ e^{j\Delta \varphi }\end{array}\right],\left(\text{twenty}\right)$$

- оператор перехода из декартового поляризационного базиса, в котором записаны векторы Джонса излучаемых волн, в систему координат поляризационного разделителя;Where $$\left[\begin{array}{cc}\cos \theta & -\sin \theta \\ \sin \theta & \cos \theta \end{array}\right]$$

- the transition operator from the Cartesian polarization basis, in which the Jones vectors of the emitted waves are recorded, into the coordinate system of the polarization separator;

- оператор поляризатора первого плеча линейного поляризационного разделителя; $$\left[\begin{array}{cc}\mathrm{one}& 0\\ 0& 0\end{array}\right]$$

- polarizer operator of the first shoulder of the linear polarizing separator;

- оператор поляризатора второго плеча линейного поляризационного разделителя. $$\left[\begin{array}{cc}0& 0\\ 0& \mathrm{one}\end{array}\right]$$

- polarizer operator of the second shoulder of the linear polarizing separator.

и $${\dot{\overrightarrow{E}}}_2$$

на выходе линейного поляризационного разделителя 5 вида:After the transformations, we obtain analytical expressions for the signals $${\dot{\overrightarrow{E}}}_{\mathrm{one}}$$

and $${\dot{\overrightarrow{E}}}_2$$

at the output of the linear polarizing separator 5 of the form:

$${\dot{\overrightarrow{E}}}_{\mathrm{one}}=\cos \theta -\sin \theta e^{j\Delta \varphi },\left(\text{21}\right)$$

$${\dot{\overrightarrow{E}}}_2=\sin \theta +\cos \theta e^{j\Delta \varphi }.\left(\text{22}\right)$$

и $${\dot{\overrightarrow{E}}}_2$$

, через соответствующие им коаксиально-волноводные переходы 6 и 7 поступают на входы суммарно-разностного блока 8 и имеют видFrom the outputs of the shoulders of the linear polarizing separator 5, oriented at an angle θ = 45 ° to the measurement plane, the signals $${\dot{\overrightarrow{E}}}_{\mathrm{one}}$$

and $${\dot{\overrightarrow{E}}}_2$$

, through the corresponding coaxial-waveguide transitions 6 and 7 arrive at the inputs of the total-difference block 8 and have the form

$${\dot{E}}_{\mathrm{one}}=\frac{\sqrt{2}}{2}\left(\mathrm{one}-e^{j\Delta \varphi }\right),\left(\text{23}\right)$$

$${\dot{E}}_2=\frac{\sqrt{2}}{2}\left(\mathrm{one}+e^{j\Delta \varphi }\right).\left(\text{24}\right)$$

и $${\dot{E}}_2$$

на входе суммарно-разностного блока 8 имеют вид:Accordingly, the amplitudes A ₁ and A ₂ , as well as the phases ψ ₁ and ψ _{2 of the} signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the input of the total-difference block 8 are:

$$A_{\mathrm{one}}=\sqrt{\mathrm{one}-\cos \Delta \varphi },\left(\text{25}\right)$$

$${\psi }_{\mathrm{one}}=-arctg\frac{\sin \Delta \varphi }{\mathrm{one}-\cos \Delta \varphi },\left(\text{26}\right)$$

$$A_2=\sqrt{\mathrm{one}+\cos \Delta \varphi },\left(\text{27}\right)$$

$${\mathrm{<figure-callout\; id="2"\; label="\psi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_2=arctg\frac{\sin \Delta \varphi }{\mathrm{one}+\cos \Delta \varphi },\left(\text{28}\right)$$

and their phase difference, after transformations, has the form:

$$\Delta \psi ={\psi }_{\mathrm{one}}-{\mathrm{<figure-callout\; id="2"\; label="\psi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_2=\frac{\mathrm{<figure-callout\; id="2"\; label="\pi "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\pi </figure-callout>}}{2}.\left(\text{29th}\right)$$

и $${\dot{E}}_2$$

на входе суммарно-разностного блока 8 зависят только от измеряемого параметра разности фаз Δφ между принимаемыми на борту подвижного объекта ортогонально линейно поляризованными электромагнитными волнами от передающих антенн 2 и 3.From the analysis of (25) and (27) it follows that at θ = 45 ° the signal amplitudes $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the input of the sum-difference block 8, they depend only on the measured parameter of the phase difference Δφ between the orthogonally linearly polarized electromagnetic waves from the transmitting antennas 2 and 3 received on board the moving object.

и $${\dot{E}}_2$$

на входе суммарно-разностного блока 8 постоянна и, наоборот, не зависит от измеряемого параметра разности фаз Δφ.At the same time, analysis of (29) shows that the phase difference Δφ between the signals $${\dot{E}}_{\mathrm{one}}$$

and $${\dot{E}}_2$$

at the input of the total-difference block 8 is constant and, conversely, does not depend on the measured parameter of the phase difference Δφ.

, суммарный $${\dot{E}}_{\Sigma }$$

и разностный $${\dot{E}}_{\Delta }$$

сигналы вида:The outputs of the total-difference block 8 are formed normalized, accurate to a constant coefficient $$\mathrm{one}/\sqrt{2}$$

total $${\dot{E}}_{\Sigma }$$

and differential $${\dot{E}}_{\Delta }$$

signals of the form:

и $${\dot{E}}_{\Delta }=\frac{1}{\sqrt{2}}\cdot \left({\dot{E}}_1-{\dot{E}}_2\right).\left(\text{31}\right)$$

$${\dot{E}}_{\Sigma }=\frac{\mathrm{one}}{\sqrt{2}}\cdot \left({\dot{E}}_{\mathrm{one}}+{\dot{E}}_2\right),\left(\text{thirty}\right)$$

and $${\dot{E}}_{\Delta }=\frac{\mathrm{one}}{\sqrt{2}}\cdot \left({\dot{E}}_{\mathrm{one}}-{\dot{E}}_2\right).\left(\text{31}\right)$$

и разностного $${\dot{E}}_{\Delta }$$

сигналов на выходе суммарно-разностного блока 8 вида:Substituting (23) and (24) in (30) and (31) and introducing, for clarity of representations, the time dependence of the signals, we obtain analytical expressions for the total $${\dot{E}}_{\Sigma }$$

and difference $${\dot{E}}_{\Delta }$$

signals at the output of the total-difference block 8 of the form:

и $${\dot{E}}_{\Delta }=-1\cdot e^{j\left(\omega t+\Delta \varphi \right)}.\left(\text{33}\right)$$

$${\dot{E}}_{\Sigma }=\mathrm{one}\cdot e^{j\omega t},\left(\text{32}\right)$$

and $${\dot{E}}_{\Delta }=-\mathrm{one}\cdot e^{j\left(\omega t+\Delta \varphi \right)}.\left(\text{33}\right)$$

и разностного $${\dot{E}}_{\Delta }$$

сигналов и их разность фаз ψ_Δ-ψ_Σ имеют вид:Accordingly, the amplitudes A _Σ and A _{Δ of the} total $${\dot{E}}_{\Sigma }$$

and difference $${\dot{E}}_{\Delta }$$

signals and their phase difference ψ _Δ -ψ _Σ have the form:

$$A_{\Sigma }=A_{\Delta }=\mathrm{one,}\left(\text{34}\right)$$

and their phase difference

$${\psi }_{\Delta }-{\psi }_{\Sigma }=\pm arctgtg\Delta \varphi \pm n\pi ,\left(\text{35}\right)$$

where n = 0, 1, 2, ...,

or, after transformations, taking into account n = 0, we obtain

$${\psi }_{\Delta }-{\psi }_{\Sigma }=\pm \Delta \varphi .\left(\text{36}\right)$$

и разностного $${\dot{E}}_{\Delta }$$

выходных сигналов суммарно-разностного блока 8 постоянны и равны между собой (34) и не зависят от углового положения подвижного объекта. В то же время разность фаз ψ_Δ-ψ_Σ между разностным $${\dot{E}}_{\Delta }$$

и суммарным $${\dot{E}}_{\Sigma }$$

сигналами на выходе суммарно-разностного блока 8, наоборот определяется угловым положением подвижного объекта и совпадает с измеряемой разностью фаз Δφ между принимаемыми ортогонально линейно поляризованными электромагнитными волнами и связаны между собой соотношением (36).From the analysis of (34) and (36) it follows that when the linear polarizing separator 5 is oriented at an angle θ = 45 ° to the measurement plane, the amplitudes A _Σ and A _{Δ of the} total $${\dot{E}}_{\Sigma }$$

and difference $${\dot{E}}_{\Delta }$$

the output signals of the total-difference block 8 are constant and equal to each other (34) and do not depend on the angular position of the moving object. At the same time, the phase difference ψ _Δ -ψ _Σ between the difference $${\dot{E}}_{\Delta }$$

and total $${\dot{E}}_{\Sigma }$$

signals at the output of the sum-difference block 8, on the contrary, is determined by the angular position of the moving object and coincides with the measured phase difference Δφ between the received orthogonally linearly polarized electromagnetic waves and are interconnected by relation (36).

и разностный $${\dot{E}}_{\Delta }$$

сигналы, описываемые аналитическими выражениями (32) и (33) с параметрами (34) и (36) поступают на входы фазового углового дискриминатора 9 (см. фиг.2), т.е. поступают соответственно на первые входы смесителей частоты 11 и 12, а на их вторые входы поступает сигнал с выхода гетеродина 14. После чего, сигнал с выхода первого смесителя частоты 11 через фазовращатель на 90° 13 поступает на вход первого УПЧ с ограничением по амплитуде 15, а сигнал с выхода второго смесителя частоты 12 поступает на вход второго УПЧ с ограничением по амплитуде 16. В УПЧ 15 и 16, имеющих идентичные амплитудно-фазочастотные характеристики, осуществляется усиление сигналов промежуточной частоты, а также производится их нормировка за счет амплитудного ограничения усиливаемых сигналов промежуточной частоты с порогом ограничения U₀. Затем выходной сигнал УПЧ 15 поступает на первый вход фазового детектора 17, а выходной сигнал УПЧ 16 поступает на второй вход фазового детектора 17. На выходе фазового детектора 17 формируется сигнал пропорциональный синусу разности фаз ψ_Δ-ψ_Σ входных сигналов и имеет вид:From the outputs of the total differential block 8 total $${\dot{E}}_{\Sigma }$$

and differential $${\dot{E}}_{\Delta }$$

the signals described by analytical expressions (32) and (33) with parameters (34) and (36) are fed to the inputs of the phase angular discriminator 9 (see Fig. 2), i.e. respectively, are fed to the first inputs of the frequency mixers 11 and 12, and a signal from the output of the local oscillator 14 is received at their second inputs. After that, the signal from the output of the first frequency mixer 11 through the phase shifter is 90 ° 13 fed to the input of the first amplifier with a limit of amplitude 15, and the signal from the output of the second frequency mixer 12 is fed to the input of the second amplifier with a limitation of amplitude 16. In the amplifiers 15 and 16, which have identical amplitude-phase-frequency characteristics, the signals of the intermediate frequency are amplified, and their normalization They score clipping the amplified intermediate frequency signal to a threshold limit U _0. Then, the output signal of the amplifier 15 is supplied to the first input of the phase detector 17, and the output signal of the amplifier 16 is fed to the second input of the phase detector 17. At the output of the phase detector 17, a signal is generated proportional to the sine of the phase difference ψ _Δ -ψ _{Σ of the} input signals and has the form:

$$S\left(\alpha \right)={\mathrm{<figure-callout\; id="0"\; label="U"\; filenames="00000076.png"\; state="\{\{state\}\}">U</figure-callout>}}_0\cdot \sin \left({\psi }_{\Delta }-{\psi }_{\Sigma }\right),\left(\text{37}\right)$$

or considering (36)

$$S\left(\alpha \right)={\mathrm{<figure-callout\; id="0"\; label="U"\; filenames="00000076.png"\; state="\{\{state\}\}">U</figure-callout>}}_0\cdot \sin \Delta \varphi ,\left(\text{38}\right)$$

where U ₀ = const.

и подставляя из (18) значение Δφ в (38) получим выражение для пеленгационной характеристики вида:Normalizing (38) $$S\left(\alpha \right)/{\mathrm{<figure-callout\; id="0"\; label="U"\; filenames="00000076.png"\; state="\{\{state\}\}">U</figure-callout>}}_0$$

and substituting the value Δφ into (38) from (18), we obtain the expression for the direction-finding characteristic of the form:

$$\frac{S\left(\alpha \right)}{{\mathrm{<figure-callout\; id="0"\; label="U"\; filenames="00000076.png"\; state="\{\{state\}\}">U</figure-callout>}}_0}=\sin \left(\frac{2\pi d}{\lambda }\cdot \sin \alpha \right).\left(\text{39}\right)$$

For small values of α, assuming U ₀ = 1, dependence (39) has an approximately linear character:

$$\frac{S\left(\alpha \right)}{{\mathrm{<figure-callout\; id="0"\; label="U"\; filenames="00000076.png"\; state="\{\{state\}\}">U</figure-callout>}}_0}\approx \frac{2\pi d}{\lambda }\alpha .\left(\text{40}\right)$$

Thus, by the signal from the output of the phase detector 17, it is possible to determine the value and sign of the angle of deviation α from the perpendicular restored to the middle of the base d formed by the transmitting antennas 2 and 3.

в точке α=0°, получим выражение для крутизны пеленгационной характеристики вида:Finding the derivative of the normalized output signal of the phase detector 17 $$S\left(\alpha \right)/{\mathrm{<figure-callout\; id="0"\; label="U"\; filenames="00000076.png"\; state="\{\{state\}\}">U</figure-callout>}}_0$$

at the point α = 0 °, we obtain the expression for the steepness of the direction-finding characteristic of the form:

$$\mu \left(\alpha \right)={|\; |\; |\frac{d\left(S\left(\alpha \right)\right)/U_0}{d\alpha }|\; |\; |}_{\alpha =0}={|\; |\; |\frac{d}{d\alpha }\sin \left(\frac{2\pi d}{\lambda }\sin \alpha \right)|\; |\; |}_{\alpha =0}=\frac{2\pi d}{\lambda }.\left(\text{41}\right)$$

.It follows from (41) that the steepness of the direction-finding characteristic, and, consequently, the direction-finding accuracy increase with increasing ratio $$d/\lambda$$

.

и суммарным $${\dot{E}}_{\Sigma }$$

выходными сигналами суммарно-разностного блока 8, с учетом (37) и (18), рассчитывается пеленг α подвижного объекта в заданной системе координат и при заданном отношении $$d/\lambda$$

, по формуле:From the output of the phase detector 17, the signal enters the calculator 10, where according to the measured phase difference ψ _Δ -ψ _Σ between the difference $${\dot{E}}_{\Delta }$$

and total $${\dot{E}}_{\Sigma }$$

the output signals of the total-difference unit 8, taking into account (37) and (18), the bearing α of the moving object is calculated in a given coordinate system and for a given ratio $$d/\lambda$$

, according to the formula:

$$\alpha =\arcsin \left\{\frac{\mathrm{<figure-callout\; id="2"\; label="\lambda "\; filenames="00000076.png,00000077.png"\; state="\{\{state\}\}">\lambda </figure-callout>}}{2\pi d}\left[\pm \left({\psi }_{\Delta }-{\psi }_{\Sigma }\right)\pm n\pi \right]\right\},\left(\text{42}\right)$$

where n = 0, 1, 2, ....

Let us conduct a comparative analysis of the accuracy of measurements of the bearing α of a moving object between the prototype device and the claimed radio navigation system.

From a comparative analysis of (17) and (41), it follows that when a moving object is on an equal-signal direction, or in directions close to an equal-signal direction, the steepness of the direction-finding characteristic in the inventive radionavigation system, for the same d / λ ratio, is two times higher than the prototype device. And since it is known [1, 7] that, for the same signal-to-noise ratio at the input of the receiving channels, the direction finding error δ _α is related to the steepness of the direction-finding characteristic µ by the ratio [7]:

$${\delta }_{\alpha }=\frac{\mathrm{one}}{\mu },\left(\text{43}\right)$$

then an increase in its slope by half reduces the direction finding error δ _α, respectively, by a factor of two, and thereby provides a higher accuracy of measurements of bearing α.

Improving the accuracy of measurements of the bearing α of a moving object in these directions is achieved through the use of total-difference signal processing at the output of a linear polarizing separator oriented at an angle of 45 ° to the measurement plane.

Conducting a comparative analysis of expressions (25) and (29) with the corresponding expressions (34) and (36) we see that the total-difference signal processing allows you to convert the angular information about the moving object, which is contained in the prototype device in the amplitude relationships (25) and (27) the signals at the output of the linear polarizing separator, into the phase relationships of the signals (36) at the output of the total-difference block in the inventive radio navigation system. In this case, the direction-finding characteristic (14) is transformed into the direction-finding characteristic (37). Moreover, these transformations are carried out at a high frequency to the inputs of the receiving channels using passive elements in the form of coaxial lines due to their simplicity and stability of characteristics. In addition, the use of sum-difference processing of the output signals of the linear polarization separator at a high frequency allows, as in phase sum-difference monopulse systems [1, 7], to impose less stringent requirements on the identity of the amplitude-phase characteristics of the receiving channels, compared to direct measurement of the phase difference Δφ. It should also be noted that the total-difference processing is carried out at the output of the linear polarizing separator, oriented, as in the prototype, at an angle of 45 ° to the measurement plane. This circumstance makes it possible to provide complex independent measurements of the bearing α due to the simultaneous use of both the amplitude ratios (25) and (27) of the signals at the output of the linear polarizing separator, and the use of phase relations (36) of the signals at the output of the sum-difference block, which, of course, will increase the accuracy and reliability of the received navigation information.

In the 3 cm wavelength range, the inventive radio navigation system can be implemented as follows.

As the transmitter 1 can be used, for example, a standard generator of high-frequency oscillations of the type ГЧ-83, the output of which is connected to a power divider made in the form of a double waveguide T-shaped splitter [8]. Moreover, the first output of the splitter is connected to the transmitting antenna 2, and its second output is connected through a segment of a rectangular waveguide twisted by 90 ° to the transmitting antenna 3.

As transmitting antennas 2 and 3, a horn antenna can be used [9].

The receiving all-polarized antenna 4 can be made in the form of a round horn [8].

The linear polarizing separator 5 is made in the form of a circular waveguide with a transition to two orthogonally located rectangular waveguides [10].

The total-difference block 8 is made on coaxial lines [1].

The phase angular discriminator 9 can be performed according to the well-known [7] scheme of the phase-phase monopulse system.

The computer 10 can be performed on the basis of the on-board computer of a moving object.

The inventive radio navigation system allows to increase the accuracy of bearing measurements of a moving object by two times in cases when the moving object is on the equal signal direction and in directions close to the equal signal, due to the greater steepness of the direction finding characteristic, while maintaining the dimensions of the receiving antenna unchanged. The latter allows you to increase the effectiveness of the practical application of radio navigation aids in moving objects, where the mass and dimensions of the receiving antenna are of paramount importance.

Sources of information used in the preparation of the description of the invention:

1. Shirman Y.D. Theoretical foundations of radar. - M.: "Transport", 1973.

2. Yarlykov M.S. Statistical theory of radio navigation. - M.: “Radio and Communications”, 1985. - 343 p.

3. Astafiev G.P., Shebshaevich B.C., Yurkov Yu.A. Radio engineering navigation devices for aircraft. - M .: "Soviet Radio", 1962. - 963 p.

4. Kazarinov Yu.M. and other radio engineering systems. - M .: "Soviet Radio", 1968. - 495 p.

5. Badulin NN, Gulko V.L. Navigation system for bearing detection. - A.S. No. 1355955, M. cl. ⁴ , G01S 3/02, priority dated 12/9/1985.

6. Badulin NN, Gulko V.L. A method of measuring the bearing of a moving object and a device for its implementation. - USSR patent No. 1251003, M. cl. ⁴ , G01S 3/02, priority from 01/29/1985.

7. Leonov A.I., Fomichev K.I. Monopulse radar. - M.: “Radio and Communications”, 1984. - 312 p.

8. Zhuk M.S., Molochkov Yu.B. Design of lens scanning wide-range antennas and feeder devices. - M .: "Energy", 1973. - 401 p.

9. Drabkin A.L. and other Antenna-feeder devices. - M .: "Soviet Radio", 1974.

10. Kanareikin D.B., Pavlov N.F., Potekhin V.A. Polarization of radar signals. - M .: "Soviet Radio", 1966. - 440 p.

Claims (1)

A radionavigation system for measuring the bearing of a moving object, comprising a transmitter, the output of which is connected to the inputs of two spaced transmit antennas with orthogonal proper polarizations located at points with known coordinates in the measurement plane at a distance d from each other, and a receiving all-polarized antenna located on the moving object, the output of which is connected to the input of a linear polarizing separator, made in the form of a transition from a circular waveguide to two orthogonally located x with respect to each other a rectangular waveguide, the unit vectors of the coordinate system of which coincide with the axes of the shoulders of the rectangular waveguides and are oriented at an angle of 45 ° to the measurement plane, and a calculator, characterized in that the first and second coaxial-waveguide transitions, a sum-difference block, are introduced performed on coaxial lines and a phase angular discriminator, while the inputs of the first and second coaxial waveguide transitions from the side of rectangular waveguides are connected to the outputs of the linear polarization separator, and their outputs are connected to the inputs of the sum-difference block, and its two outputs are connected to the corresponding two inputs of the phase angular discriminator, and its output is connected to the input of the calculator, and the transmitting antennas have horizontal and vertical linear eigenpolarizations, respectively.

Images