RU2115099C1 - Source of electric signal proportional to absolute temperature - Google Patents

Abstract

FIELD: microelectronic temperature sensitive elements, sources of reference voltage. SUBSTANCE: for exclusion of necessity of use of ohmic resistors with low temperature coefficient of resistance and outside resistor for conversion of current to voltage proposed source of electric signal proportional to absolute temperature has heat-sensitive couple composed of two transistors T, current density in first one being less than in second one, ohmic resistor R placed in current circuit TI, controlled current sources connected to output electrodes of transistors. Inputs of single- stage differential amplifier are connected to transistors. Transistor in diode connection placed as driving one by circuit of current mirror with transistor is load to differential amplifier on side of inverting input. Source is inserted with additional current source which currents are proportional to currents in transistors and ohmic resistor R having same temperature coefficient of resistance as R has. EFFECT: improved functional reliability of source. 3 cl, 6 dwg

Description

 The invention relates to electronic equipment and can be used to create microelectronic temperature sensors and voltage reference sources.

 Known devices for generating an electrical signal proportional to absolute temperature, based on the exponential dependence of the increment of the output current of the transistors on the ratio of the increment of the input voltage to the absolute temperature. This property is possessed by bipolar transistors and MOS transistors in a state of weak inversion. For the convenience of describing devices using this common property for bipolar and MOS transistors, hereinafter, everywhere, the input voltage of bipolar transistors means base-emitter voltage, and the input voltage of MOS transistors means gate-source voltage. By input and output electrodes for bipolar transistors are meant a base and a collector, respectively, and for MOS transistors, a gate and a drain, respectively.

The difference in input voltages of a pair of such devices operating at a constant ratio of output current densities other than unity is proportional to the absolute temperature. The current density of bipolar transistors refers to the ratio of the collector current to the emitter area. And under the current density of MOS transistors is the ratio of the product of the drain current by the channel length to its width. It is known that the MOS transistor is in a state of weak inversion (in the subthreshold region of the current-voltage characteristics) if the drain current density is much less than μCU $\genfrac{}{}{0ex}{}{\text{2}}{\text{t}}$ , where μ is the mobility of charge carriers in the channel, C is the specific capacity of the gate oxide, U _g = kT / g is the thermal potential.

The circuit of one of such devices (Fig. 1) with a thermosensitive pair consisting of MOS transistors in a state of weak inversion is described in "IEEE J. of Sol.-State Circ.", 1985, v.20, N. 3, p. 657- 665. The current mirror on the pMOS transistors VT3, VT4 maintains the required ratio of current densities in the pMOS transistors VT1, VT2 of the thermosensitive pair, and the voltage drop (U _o ) on the resistor R1 is equal to the difference in the input voltages of the transistors of the thermosensitive pair and is proportional to the absolute temperature:
U _o = U _t L _n (j ₂ / j ₁ ),
or
U _o = U _t L _n [(W ₁ / L ₁ ) (L ₃ / W ₃ ) (L ₂ / W ₂ ) (W ₄ / L ₄ )],
where U _t = kT / g is the thermal potential; j ₁ and j ₂ - current density in transistors VT1 and VT2;
W ₁ , W ₂ , W ₃ , W ₄ and L ₁ , L ₂ , L ₃ , L ₄ are the width and channel length of the transistors VT1, VT2, VT3, VT4, respectively.

The current value in the transistors of a thermosensitive pair, determined by the size ratio of all four transistors, resistance R ₁ and temperature, is chosen so as to provide transistors VT1, VT2 with a weak inversion state. With reasonable transistor sizes, this current does not exceed 1 μA.

The disadvantage of the described circuit is the small value of the output voltage, in practice not exceeding several tens of millivolts. Direct amplification of such voltage using a simple CMOS op-amp (op-amp) is not possible due to the large zero-offset CMOS op-amp, which usually also amounts to tens of millivolts. Another disadvantage is the dependence of the output signal on the supply voltage and the corresponding error arising from the difference in the drain voltages of transistors VT1 and VT2 and the final value of their output resistance. Indeed, the difference between their runoff voltages is approximately equal to EV _TP -V _Tr , where E is the supply voltage, V _TP and V _Tr are the threshold voltages of n and pMOS transistors.

 A better circuit described in "IEEE J. of Sol.-State Circ.", 1979, v. 14, N 3, p. 573-577, shown in FIG. 2. To increase the voltage proportional to the absolute temperature, a resistor R2 is added here, the voltage drop at which is proportional to the absolute temperature, since the current through R2 is proportional to the current through R1, and the voltage to R1 is proportional to the absolute temperature.

The disadvantages of this circuit are the increase in the minimum allowable supply voltage (due to a voltage drop by R2), the dependence of the output voltage on the supply voltage due to the difference in the drain voltages of the thermosensitive couple transistors, as in the circuit in FIG. 1. Finally, the leakage current of the p-pocket of transistors VT1 and VT2 flowing through R2 becomes comparable to the useful current and creates a significant error in the output voltage even at temperatures above 50 ^o C.

Another embodiment of a circuit for generating a voltage proportional to absolute temperature is shown in FIG. 3. This two-pin Texas Instruments STP-35 temperature sensor is described in Wigleb G.'s book “Sensors”. - M .: Mir, 1989, p. 29 - 33. The thermosensitive pair in this circuit consists of bipolar transistors VT1 and VT2, which, together with the loads R5, R4 and current source I ₁ , form a differential amplifier with a zero offset directly proportional to the absolute temperature. Due to the feedback action, including the operational amplifier D1 and the transistor VT3, the circuit seeks to install on the divider R2, R1, R3 such voltage U _{st so} that the voltage on R1 is equal to the zero offset of the differential amplifier. As a result, the STP-35 has a current-voltage characteristic like that of a zener diode with a stabilization voltage
U _st = (R2 + R1 + R3) / R1 U _t L _n (N),
Where
U _t kT / g - thermal potential;
N is the ratio of current densities in transistors VT1 and VT2, determined by their geometry and the resistance of resistors R4, R5.

The parameters of the circuit elements are selected so that the sensitivity U _article to temperature is 10 mV / K, then at a temperature from -40 to +120 ^o CU _article varies from 2.3 to 4 V.

The disadvantage of this scheme is the large value of the minimum allowable supply voltage, which is associated with the need to power the device from an external current source, which in the simplest case can be a ballast resistor. In FIG. 3, the STP-35 circuit itself is indicated by a dotted frame, and the external ballast is indicated by Rb. The minimum allowable supply voltage, including the voltage drop across the ballast resistor and U _st , cannot be less than 6 V to ensure operability and limit the sensor’s self-heating over the entire temperature range. The disadvantage of this circuit is also the large current consumption (0.5 - 5 mA). Its significant part is the current of the resistive divider R2, R1, R3, which should be much larger than the base currents of the transistors of the thermosensitive pair. It is difficult to implement such a scheme using CMOS technology, since all its components must, in particular, operate at a temperature of -40 ^o C, when U _st = 2.3 V, and the threshold voltage of the transistors is maximum. In addition, in the CMOS version of the circuit, self-excitation cannot be eliminated.

 Closest to the claimed object is the scheme described in US patent 4123698, publ. 10.31.78, which is shown in FIG. 4. Here the thermosensitive pair consists of bipolar transistors VT1 and VT2. The constancy of the ratio of current densities in them is ensured by controlled current sources connected to collectors VT1 and VT2 on transistors VT7 and VT8. The collector voltage difference VT1 and VT2 is measured by a single-stage differential amplifier having a differential pair of identical transistors VT4 and VT5, a current source on the diode-switched transistor VT3 of the same type as the transistors VT1 and VT2 of the thermosensitive pair, and a diode-switched transistor VT6, which is the load for VT5 . Hereinafter, a diode-switched transistor means a transistor with a closed input and output electrodes (a base with an emitter for a bipolar and a gate with a drain at the MOSFET). The transistor VT6 is connected to the transistor VT7 and VT8 according to the current mirror circuit as a master, and the base VT 3 is connected to the bases VT1 and VT2. In the emitter circuit VT1 there is an ohmic resistor R1, the voltage drop on which is proportional to the difference between the input voltages of the transistors of the thermosensitive pair VT1 and VT2.

To ensure equal collector currents VT2 and VT8 with a balanced differential amplifier, it is necessary that the ratio of current density in VT8 to current density in VT2 be equal to half the ratio of current density in VT6 to current density in VT3, which is determined by the appropriate choice of the geometric dimensions of these transistors. And the equality of the collector currents VT1 and VT7 simultaneously with the equality of the collector currents VT2 and VT8 is possible only provided that the ratio of the current density in VT1 to the current density in VT7 is less than the ratio of the current in VT2 to the current density in VT8, which is also achieved by choosing the geometry of these transistors . For the circuit under consideration, the latter condition is equivalent to a lower current density in VT1 compared to VT2. A voltage is automatically set on resistor R1
U = U _t • L _n (j ₂ / j ₁ ),
where U _t = kT / g is the thermal potential, j ₁ and j ₂ are the current densities in VT1 and VT2, respectively. The resistor R1 according to US patent N 4123698 has a very low temperature coefficient of resistance (TCR), so the current in R1 and in the transistor VT1 is proportional to the absolute temperature. Due to the proportionality of the currents in transistors VT7 and VT8, the current in transistor VT2 is also proportional to the absolute temperature. Finally, the current of the transistor VT3 is also proportional to the absolute temperature, since VT3 has the same input voltage as VT2. Thus, the current consumed by the circuit is proportional to the absolute temperature. This current is the output electrical signal proportional to the absolute temperature. Due to the action of a differential amplifier equalizing the collector voltages of the thermosensitive couple transistors, the circuit has good metrological characteristics, and its output current is very slightly dependent on the supply voltage, which can vary over a wide range. To eliminate the self-excitation of a specific implementation of this circuit, the Analog Devices AD590 thermal sensor uses a correction circuit including a capacitor C1 connected between the collectors of transistors VT5 and VT2. The sensitivity of such a device (it is indicated by a dotted frame in Fig. 4) is usually adjusted to a value of 1 μA / K, and to convert its current output signal to voltage, an external calibrated resistor R is connected in series with it.

 The disadvantage of this device is the need to use a special material (chromium silicide) with a low TCR for the resistor R1. Other disadvantages associated with the current output signal. Firstly, this is a large value of the minimum allowable supply voltage. Although the AD590 circuitry itself works, starting with a voltage at its terminals of about 3-4 V, the total supply voltage includes a voltage drop across an external calibrated resistor, which reaches 4 V at a standard sensitivity of 10 mV / K. Therefore, the minimum allowable sensor supply voltage cannot to be less than 7 - 8 V. Secondly, it is the need to use an external calibrated resistor to convert the current output signal to voltage, in addition to eliminate the error of the measuring circuit due to fluctuations in its temperature d lzhen have a low TCR, which increases the cost of the meter. Thirdly, the described circuit cannot be directly implemented by CMOS technology, since the output current level dictated by the requirement of a state of weak inversion for transistors of a thermosensitive pair and VT3 would be too small to ensure reliable conversion to voltage on an external calibrated resistor. That is, the current output signal is also an obstacle to reducing the current consumed by the circuit and the power dissipation.

 The aim of the invention is to eliminate the need to use ohmic resistors from special materials with low TCS in the source of the electric signal, which is proportional to the absolute temperature, and to eliminate the need to use an external calibrated resistor to convert the current signal to voltage. Another objective of the invention is to reduce the minimum allowable supply voltage, current consumption and power dissipation of the source.

 This goal is achieved by the fact that in the source of the electric signal proportional to the absolute temperature, containing a thermosensitive pair consisting of transistors of one type, in the first of which the current density is lower than in the second; an ohmic resistor, the voltage on which is proportional to the difference between the input voltages of the transistors of the thermosensitive pair, included in the current circuit of the first transistor of this pair; controlled current sources made on transistors having a different type with respect to the transistors of the thermosensitive pair, connected to the output electrodes of the transistors of the thermosensitive pair, one for each transistor of this pair; a single-stage differential amplifier with non-inverting and inverting inputs connected to the output electrodes of the first and second transistors of the thermosensitive pair, having transistors of the differential pair and a diode-connected transistor of the current source of this differential pair of the same type as the transistors of the thermosensitive pair, the input electrode of the transistor of the current source of the differential the pair is connected to the input electrodes of the thermosensitive pair transistors, and the load transi the torus of the differential pair of a single-stage differential amplifier from the side of its inverting input is a diode-switched transistor having the type of transistors of controlled current sources and connected with them according to the current mirror circuit as a leading one, an additional current source proportional to the currents in the controlled current sources, and an additional ohmic resistor, powered by current from this additional current source, and the ohmic resistor and the additional ohmic resistor have the same temperature th coefficient of resistance.

 To reduce the minimum allowable supply voltage of the described source of electric signal, an additional current source is made on transistors having the type of transistors of controlled current sources, and is connected with them according to the current mirror circuit driven.

 To reduce the current consumption and dissipated power of the described electric signal source, the transistors of the thermosensitive couple and the current source for the differential pair are nMOS transistors in a state of weak inversion in a common grounded p-pocket, the differential pair is made on nMOS transistors, and the controlled current sources, an additional current source and diode-connected transistor, which is the load of the transistor of the differential pair of a single-stage differential amplifier from its inverter entrance guide, provided on pMOP transistors.

 To exclude self-excitation of the described source of the electric signal made on MOS transistors, a correction capacitor is connected between the output electrode of the second thermosensitive pair transistor and the ground or power bus.

 In FIG. 1 is a known source of an electrical signal proportional to absolute temperature on MOS transistors; in FIG. 2 is a known source of an electrical signal proportional to absolute temperature on MOS transistors with an output signal strength; in FIG. 3 is a block diagram of a known Texas Instruments bipolar transistor absolute temperature sensor STP-35; in FIG. 4 is a diagram of an electric signal source proportional to absolute temperature, according to US Pat. No. 4,123,698; in FIG. 5 is an example of a circuit of the inventive source of an electrical signal proportional to absolute temperature; in FIG. 6 is another example of a circuit of the inventive source of an electrical signal proportional to absolute temperature.

 In an example of the inventive source of an electrical signal proportional to the absolute temperature shown in FIG. 5, the active elements are bipolar transistors. Compared to the circuit of FIG. 4, an external calibrated resistor is excluded here and a chain is added from series-connected additional current source VT9 and additional ohmic resistor R2, connected in parallel with the rest of the circuit. This additional chain has practically no effect on the rest of the circuit, which operates in the same way as the circuit already described in FIG. 4. The output voltage taken in the resistor R2 is proportional to the absolute temperature due to the proportionality of the current in the additional ohmic resistor R2 to the current in the ohmic resistor R1, the equality of the TCS of these resistors and the proportionality of the voltage on the ohmic resistor R1 to the absolute temperature. In this case, of course, it is assumed that the temperature of both resistors is the same, as is the case with transistors of a thermosensitive pair. This condition is automatically satisfied if the signal source is made in the form of an integrated circuit with closely spaced transistors of a thermosensitive pair and closely spaced ohmic resistors.

 The additional current source VT9 in the example of FIG. 5 is made similarly to the controlled current sources VT7, VT8 and is included with them according to the slave current mirror circuit, which automatically ensures proportionality of the currents in them. The resistance of the additional resistor R2 and the current in it can be greater than the resistance and current R1, which is determined by the choice of the geometry of the resistors R1, R2 and transistors VT7, VT9. This allows you to get the necessary gain of the output signal. In this case, it is sufficient to fulfill the requirement of the ohmicity of the resistors R1 and R2 and the equality of their TCS, which is much easier than providing a small value of their TCS, as in the circuit in FIG. 4 Resistors R1 and R2 can be made, for example, from polysilicon.

 The output voltage of the circuit of FIG. 5, taken from the resistor R2, is not a term of the total supply voltage, as was the case in the circuit of FIG. 4. Therefore, the minimum allowable value of the supply voltage at a standard sensitivity value of 10 mV / K for this circuit is less than that of the circuit in FIG. 4, about 4 V.

 Since the output of the circuit of FIG. 5 is a voltage, not a current, then an external calibrated resistor is not required to convert the output signal to voltage. For the same reason, even in a bipolar transistor circuit, its current consumption and power dissipation can be slightly reduced.

 An even greater reduction in current consumption and power dissipation is achieved in the manufacture of the inventive source using CMOS technology. An example of such a scheme to illustrate claim 3 of the claimed formula is shown in FIG. 6. The thermosensitive pair here consists of nMOS transistors VT1 and VT2, which are in a state of weak inversion. The ohmic resistor R1 is connected between the source VT1 and the ground bus. Managed current sources on pMOS transistors VT7 and VT8 are connected to the drains of the thermistor pairs VT1 and VT2. The differential amplifier includes a differential pair of identical nMOS transistors VT4 and VT5 with a common source, powered by the current from the diode-switched nMOS transistor VT3, the gate of which is connected to the gates VT1 and VT2 and which is located together with them in a common grounded p-pocket. Placing VT1, VT2 and VT3 in a common grounded pocket eliminates the influence of the leakage current of the p-n junction of the pocket-substrate on the output voltage. The load for VT5 is the diode-switched pMOS transistor VT6, the gate of which is connected to the gates of the controlled current sources VT7, VT8. This provides the transistor VT6 with a leading role in the current mirror circuit formed by transistors VT6, VT7 and VT8. An additional current source on the pMOS transistor VT9 is included in the same current mirror circuit as a slave similar to VT7 and VT8, and its drain is connected to an additional ohmic resistor R2, from which the output voltage proportional to the absolute temperature is removed. The diode-switched pMOS transistor VT10, used as a VT4 load, is needed only for greater symmetry of the arms of a single-stage differential amplifier. To eliminate the self-excitation of this circuit, the correction capacitor C1 is connected between the drain VT2 and the ground bus, since the inclusion of C1 between the input and output VT5, as in FIG. 5, for CMOS circuitry is inefficient.

The transistors in the circuit of FIG. 6 have the following parameters:
threshold voltage of n-pMOS transistors at room temperature - 0.8 - 1.3 V,
gate oxide thickness - 40 - 50 nm
channel dimensions (width / length) in microns:
VT1 - 1200/8
VT2, VT3 - 200/8
VT4, VT5 - 40/10
VT6, VT10 - 10/60
VT7, VT8 - 20/60
VT9 - 100/60
Resistor Resistance:
R1 - 300 kOhm
R2 - 3900 kOhm
The capacitance of the correction capacitor C1 is 10 pF.

The operation shown in FIG. 6 diagram as follows. The differential amplifier, thanks to the feedback complex, seeks to equalize the voltages at the inputs of the differential pair, and also to establish and maintain at its inputs a common-mode voltage at which the voltage across the resistor R1 is exactly equal to U _t L _n (K), where K is the ratio current densities in transistors of a thermosensitive pair, and U _t = kT / q is the thermal potential.

 Indeed, with increasing voltage at the inverting input compared to the non-inverting current, the current in VT6 and VT8, VT7 sources controlled by it increases, while the voltage at the VT1 drain rises faster than at the VT2 drain, since the differential output resistance VT1 at the operating point is greater, than VT2 because of the resistor R1 in the source circuit, and the voltage equality at the inputs of the differential amplifier is restored.

On the other hand, a common-mode voltage is established and maintained at the inputs of the differential amplifier at which the currents in the transistors of the thermosensitive pair are equal, which is possible only if the difference in their input voltages, which is the voltage drop across R1, is U _t L _n (K ), where K is the ratio of current densities in transistors of a thermosensitive pair. If the common-mode voltage at the input of the differential pair increases, the current through VT3 and the voltage at its gate and at the gates VT2, VT1 of the thermosensitive couple transistors increase. VT1 (together with R1) has less slope at the operating point than VT2 because of the resistor R1 in the source circuit, so the voltage on its drain will increase faster than on the drain of VT2. The resulting differential voltage will redistribute the current in the differential pair, so that the current in its inverting arm VT5, VT6 decreases. The current in the VT8, VT7 current sources controlled by the transistor VT6 will also decrease, and the input common-mode voltage will decrease until its equilibrium value is restored.

The equality of drain currents VT2 and VT8 with a balanced differential amplifier in this example is ensured by the fact that the transistors VT2 and VT3 are identical, and VT8 is two times wider than VT6. The 6 times lower current density in VT1 compared to VT2 is ensured by the fact that the transistors VT8 and VT7 are identical, and the transistor VT1 is 6 times wider than VT2. If a weak inversion state is provided for transistors VT1, VT2, and VT3, then the voltage across the resistor R1 will be U _t L _n (6), where U _t = kT / q is the thermal potential. The weak inversion mode for transistors VT1, VT2 and VT3 is provided by the ratio of their sizes and the value of R1. Since the resistance of R2 is 13 times greater than R1, and the transistor VT9 is 5 times wider than VT7, the voltage on R2 will be 130 times higher than on R1. In this case, the sensitivity of the output voltage to the temperature is almost exactly 10 mV / K, the minimum allowable supply voltage is about 4.5 V, and the current consumed by the circuit at room temperature is 1.2 μA, which is 250 times less than that of the circuit in FIG. . 4. Due to the action of a differential amplifier, equalizing the potentials of the drains of the thermosensitive couple transistors, the sensitivity of the output voltage of the circuit in FIG. 6 to the supply voltage does not exceed 2 mV / V.

 An additional current source proportional to the currents in the controlled current sources can be performed on transistors of the same type as the transistors of the thermosensitive pair. For example, it can be one such transistor, included together with the second transistor of the thermosensitive pair according to the current mirror circuit as a slave. In this case, an additional resistor is connected between its drain and the positive terminal of the power source, which is convenient if an output signal of negative polarity is required. To increase the output resistance of an additional current source, it can be a cascaded pair of transistors.

 Thus, the claimed technical solution not only combines the most valuable properties of the technical solution known from US Pat. No. 4,123,698, such as the independence of the output signal from the supply voltage and the precision of converting temperature to an electrical signal with a small number of elements, but compared with it, it can be eliminated the need to use resistors in low TCS, makes it unnecessary to use an external calibrated resistor, can significantly reduce the minimum allowable voltage ix, current consumption and power dissipation of the device.

 The CMOS version of the circuit that allows the implementation of the claimed solution should also have a higher temporal stability of the characteristics, since the ohmic resistors in this case are under the influence of current with a significantly lower density than in the bipolar version, and significantly higher values of the resistors in the CMOS variant cause less influence of the resistance of their contacts on the characteristics of the circuit.

Claims (4)

 1. A source of an electric signal proportional to absolute temperature, containing a thermosensitive pair consisting of one type of transistors, in the first of which the current density is lower than in the second, an ohmic resistor, the voltage on which is proportional to the difference in input voltages of the thermosensitive pair transistors included in the current circuit the first transistor of this pair, controlled current sources made on transistors having a different type with respect to the transistors of the thermosensitive pair, connected the output electrodes of transistors of a thermosensitive pair, one for each transistor of that pair, a single-stage differential amplifier with non-inverting and inverting inputs connected to the output electrodes of the first and second transistors of the thermosensitive pair, having transistors of the differential pair and a diode-connected transistor of the current source of this differential pair of the same transistors of a thermosensitive pair of type, with the input electrode of the transistor of the current source differential of the pair is connected to the input electrodes of the thermosensitive pair of transistors, and the load of the transistor of the differential pair of a single-stage differential amplifier from the side of its inverting input is a diode-switched transistor having the type of transistors of controlled current sources and connected with them according to the current mirror circuit leading, characterized in that it is introduced an additional current source proportional to the currents in the controlled current sources, and an additional ohmic resistor powered by the current from this additional ADDITIONAL current source, wherein an ohmic resistor and a further ohmic resistor have the same resistance temperature coefficient.  2. The electric signal source according to claim 1, characterized in that the additional current source is made on transistors having the type of transistors of controlled current sources, and is connected with them according to the current mirror circuit driven.  3. The electric signal source according to claim 1, characterized in that the transistors of the thermosensitive pair and current source for the differential pair are nMOS transistors in a state of weak inversion in a common grounded p-pocket, the differential pair is made on nMOS transistors, and the controlled current sources are additional The current source and the diode-connected transistor, which is the load of the transistor of the differential pair of a single-stage differential amplifier from the side of its inverting input, are made on a pMOS transistor rah.  4. The source of the electrical signal according to claim 3, characterized in that between the output electrode of the second transistor of the thermosensitive pair and the ground or power bus, a correction capacitor is included.

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