JPH0322863A - current resonant converter - Google Patents

Abstract

PURPOSE:To prevent the generation of beat or malfunction and improve transient responding characteristics by providing a feedforward loop to absorb the fluctuation of an input voltage. CONSTITUTION:A current resonance type converter producing a constant output voltage despite variations in input and load is provided with a LC resonance circuit, consisting of two variable inductors 6, 7 and a capacitor 9, and an output voltage control circuit 16 for increasing or decreasing the control current of the variable inductor 7 in accordance with a difference between the output voltage and a reference voltage. Further, another input voltage control circuit 15 is provided for increasing or decreasing the control current of the other variable inductor 6 in accordance with an input voltage. According to this method, the inductance values of two variable inductors 6, 7 are varied with the control current to vary the resonance frequency whereby the output voltage may be controlled to a given value.

Description

[Detailed Description of the Invention] Industrial Application Fields The present invention relates to improving the control characteristics of a current resonant converter. Conventional technology> Conventional current resonant converters utilize LC resonance to
MOSF [T] is a type of switching method that changes the current of a switching element such as a smoothly changing sinusoidal waveform, and the current waveform during switching is a sinusoidal waveform (resonant waveform), and the resonant current is zero. Since it can switch when the current condition occurs, it has the characteristic of low noise and switching loss during switching. Therefore, this current resonant converter is considered to be an extremely effective method for reducing the noise and increasing the frequency of switching power supplies (reducing the size of devices).The output voltage control in this current resonant converter is as follows: Ratio of switching frequency fs to resonant frequency fn (fs/f
This is done by changing n), and the relationship is expressed by the following formula. Vo /Vi = fS /fn - ■Where, Vi: Input voltage (DC) VO: Output voltage (DC). <Problem to be solved by the invention> However, in the current resonant converter shown in the above-mentioned prior art, since a fixed inductor or a fixed capacitor is used as a resonant element, the resonant frequency fn shown in the above formula (■) is not constant. Yes, in order to control the output voltage VO,
It was necessary to change the switching frequency fs. Therefore, since the switching frequency fluctuates due to load fluctuations and human power fluctuations, the output filter must be designed at the lowest frequency, making it impossible to make the device compact. Furthermore, when converters are operated in parallel, beats and malfunctions are likely to occur. Furthermore, since the switching frequency varies over the range, noise is distributed over a wide range, making it difficult to reduce noise. The present invention has been made based on the problems of the prior art described above, and it is possible to control the output voltage while keeping the switching frequency fixed in a current resonant converter. The purpose is to provide a current resonant converter that can improve voltage accuracy and transient response characteristics. Means for Solving the Problems> Another aspect of the present invention to solve the above problems is to provide a current resonant converter that controls the output voltage to a constant value in response to input fluctuations and load fluctuations. An LC resonant circuit from a capacitor, an output voltage control concave path that increases or decreases the control current of one of the variable inductors in accordance with the difference between the output voltage and a reference voltage, and the control current of the other variable inductor to the input voltage. An input voltage control circuit is provided to increase or decrease the input voltage accordingly, and the inductance value of the two variable inductors is changed by the control current, thereby changing the resonance frequency, thereby controlling the output voltage to a constant value. It is characterized by <Operation> According to the present invention, the variable inductor for resonance is divided into two, one corresponds to the input voltage, and the other corresponds to the output voltage, and the inductance value is changed to change the resonant frequency, thereby achieving switching. While the frequency remains constant,
Not only can the output voltage be controlled to a constant value, but also the inductance can be changed in a similar manner to the input voltage, making it possible to improve the controllability of the feedback loop. Embodiments> The present invention will be explained below based on the drawings. FIG. 1 is a block diagram showing an embodiment of a current resonant converter according to the present invention, and shows all waveforms of a half-bridge circuit type.

In Figure 1, V1 is the input voltage, 1 is the input smoothing capacitor, 2 and 3 are dividing capacitors that divide the input voltage V1 into two, and 4 and 5 are switching transistors ( 6 is a resonant variable inductor with one end of the inductor winding connected to the connection point of the split capacitors 2 and 3;
7 is a resonant variable inductor in which one end of the inductor winding is connected to the other end of the inductor winding of the resonant variable inductor 6; 8 is the connection point between the transistors 4 and 5 and the inductor winding of the variable inductor 7; A transformer is connected to both ends of the primary winding at the other end, 9 is a resonance capacitor connected to both ends of the primary winding of the transformer 8, and 10 and 11 are transformer 8.
rectifying diodes whose anodes are connected to both ends of the secondary windings of the rectifying diodes 10 and 11; 12 is a choke whose one end is connected to the cathodes of the rectifying diodes 10 and 11;
14 is fI of each choke coil 12! ! An output smoothing capacitor and a load resistor are connected between the end and the midpoint of the transformer 8, and 12 and 13 are horizontal output filters. Vo is the output voltage applied across the load resistor 14. Further, 15 is an input voltage control circuit that applies a control current corresponding to a change in the input voltage V1 to the control winding of the variable inductor 6, and 16 is a circuit that controls the variable inductor 7 by applying a control current corresponding to the difference between the output voltage VO and the reference voltage. This is a voltage control circuit that applies voltage to the winding. Also, V61 and ■62 are transistors 4,
5, os is the drain-source voltage of the transistor 4, os is the voltage across the resonance capacitor 9, and I1 is the t current flowing through the resonance variable inductors 6 and 7. Also, FIG. 2 shows the variable inductors 6 and 7 used in FIG.
These are the configuration diagram (Figure A), equivalent circuit diagram (Figure B), and characteristic diagram (Figure 8). In Figure 2 (A), a control current T,1 (Io2) is passed through the b1-b2 windings wound around the center leg, and the 1-a2 windings are arranged so that the generated magnetic flux cancels each other out in the middle leg. It is then wrapped around two legs that sandwich the middle leg. The equivalent circuit is shown in figure (b), and the voltage applied to the b1-b2 windings is:
In the a1 and a2 windings, they cancel each other out and become zero. Its characteristics are (c) As shown in the figure, the core is saturated by the control current I,1 (io2) flowing through the b1-b2 windings, and b
The inductance L of the 1-b2 winding changes. In Figure 1, the horizontal commands in Figure 2 are used for input and output, respectively. Note that the variable inductor is not limited to the one shown in Figure 2, but any type that can make the inductance variable depending on the control current may be used. 3 and 4 show the input voltage control circuit 15 and output voltage control circuit 16 used in FIG.
This is a horizontal diagram showing a specific example. In the input voltage control circuit 15 shown in FIG. 3, the input voltage Vi is divided by voltage dividing resistors R1 and R2, and the divided output voltage E. Perform functional operations on . The output E of Part 1 is current amplified by a circuit consisting of an operational amplifier OP, a transistor Q021, and a resistor R,i, and is made into a control current I,1 that drives the control winding of the variable inductor 6. On the other hand, in the output voltage control circuit 16 shown in FIG. 4, the difference between the output voltage VO and the reference voltage V is calculated and amplified by a subtraction circuit consisting of resistors R to R4 and an amplifier OP1. The output is current amplified by a circuit consisting of an operational amplifier OP2, a transistor Q1, and a resistor Rfo, and is used as a control current Ic2 that drives the control winding of the variable inductor 7.

FIG. 5 is an operation waveform diagram for explaining the operation of FIG. 1. When the transistor 4 is turned on, that is, the gate drive voltage VG1 of the transistor 4 becomes high level, a resonance current is generated in the resonance variable inductors 6 and 7 and the resonance capacitor 9.
, flows in the direction of the arrow shown in Figure 1. Since this resonant t-flow 11 is in a resonant state, it becomes a sine wave. Returning to A and Zero, the parasitic diode of transistor 4 (omitted in the diagram)
Therefore, it continues to flow in the opposite direction. The negative part of resonance current ■1 is a state in which energy is generated back to the input. In this state, the train-source voltage of transistor 4 ■, s
is very small due to the parasitic diode 1ζ, and since no current flows through transistor 4, switching loss will be extremely small if transistor 4 is turned off in this state. Transistors 4 and 5 operate symmetrically, and current and voltage in opposite directions are generated in the transformer winding when transistor 4 is on and when transistor 5 is on. This is passed through U with the secondary open, and the output voltage V
o. Here, FIG. 6 is a diagram showing the input force characteristics, and the horizontal axis shows the ratio (fs
/fn), and the vertical axis is (2N-Vo/Vi)
is taken and expressed, where N is 1 of transformer ′r゛
Next, the ratio of the number of turns n1 to the number of ports on the secondary side n2 (n1/n2)
It is. Also, r = R 0 /Zn is a normalized resistance, but it is assumed that the load resistance is 14, and z0 is a characteristic impedance, which is 7 L = L, +Lr2 r when Zo = 9. As shown in Figure 6, the load is ``(n (r = 1 -
+ r = 10), the input/output characteristics hardly change, indicating that it is resistant to load fluctuations. However, the output voltage Vo
is proportional to the input voltage Vi, and the effects of input fluctuations appear directly on the output. The relationship in Figure 5 is 2N-Vo /Vi =k- fs /fn -...
■However, k: can be expressed as a constant of proportionality, but this ■formula can be transformed as shown in the following formula. 2N − Vo /Vi =k−fs・2πF bow 1f...■ However, fn=1/2πF11 - ``''. From this -L equation, if C7 is made proportional to 1/V i, the output voltage 0 will not depend on the input voltage V1 and will be constant. In other words, it is possible to realize a device that is not affected by human movement and load fluctuations. In conventional devices, input fluctuations are also stabilized by feedback tb,
A large dynamic range of feedback is required. Therefore, loop gain is sacrificed, resulting in problems such as decreased output voltage accuracy or poor transient response characteristics. Therefore, in the present invention, as shown in FIG. 1, the variable inductor is divided into two, and one variable inductor 6 is changed in accordance with the input voltage Vi.

Here, in the input voltage control [u1 path 15 shown in FIG. 3, the relationship between the input voltage ■1 and the control current Icl is 1,1=
In the characteristic diagram of the variable inductor shown in FIG. 2 (C), the inductance L and the control current ■. The relationship of 1 is L=g(1.1)
...If you shift from ■, then from the formulas ■ and ■, L=g If (Vi) )=k/(Vi)
By selecting g and f so that , feedback rube i
The burden is greatly reduced. In addition, since the full waveform current resonant converter is hardly affected by load fluctuations, if the influence of input fluctuations is eliminated by feedforward, the control range of the feedback loop can be made considerably smaller. (Control range of variable inductor 6) Furthermore, since controlling the variable inductor becomes an inductance load, reducing the control current to reduce loss in the drive circuit increases the number of turns in the control winding. Although the inductance becomes larger and the transient response characteristics become worse (the magnitude of current change becomes smaller), the control range becomes smaller, so
The number of turns in the control winding can also be reduced, making it possible to speed up transient response characteristics. Effects of the Invention> As described above in detail with the embodiments, according to the present invention, (1) the converter is capable of controlling the output voltage to a constant value while keeping the switching frequency constant; It is possible to eliminate the occurrence of beats and malfunctions that occur when operating in parallel. (2) Noise countermeasures are easy because the output filter can be designed to match the switching frequency. (3) By increasing the switching frequency, the output filter can be made smaller.

In addition, a feedforward loop is provided to absorb input voltage fluctuations, and (4) the variable inductance of the feedback loop can be reduced, improving transient response characteristics. (5) Input changes are generally gradual and transient response characteristics do not pose much of a problem, so
With the variable inductance of the input song, the number of turns can be increased, and the limited power can be reduced. (6) Since the control range of the feedback loop is small, there is no need to set the open loop gain of the feedback loop to a large value, and the loop stability and output voltage consistency can be improved. It is possible to realize a current resonant converter.

[Brief explanation of drawings]

Fig. 1 is a horizontal diagram showing one implementation of a current resonant converter according to the present invention, and Fig. 2 is an 'iiT diagram used in Fig. 1.
The configuration diagram, equivalent circuit diagram, and characteristic diagram of the variable inductor; Figure 3 is the top diagram of the input voltage control circuit used in Figure 1;
The figure is a horizontal view of the output voltage control circuit used in Figure 1, Figure 5 is an operating waveform diagram for explaining the operation of Figure 1, and Figure 6 is an input/output characteristic diagram of Figure 1. 6, 7... Variable inductor for resonance, 9... Capacitor for resonance, 15... Input voltage control circuit, 16...
Output voltage control circuit, ■1...Input voltage, Vo...y
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Claims (1)

[Claims] In a current resonant converter that controls the output voltage to a constant value in response to input fluctuations and load fluctuations, an LC resonant circuit consisting of two variable inductors and a capacitor is used, and the control current of one of the variable inductors is controlled between the output voltage and the reference voltage. An output voltage control circuit that increases or decreases the control current of the other variable inductor according to the input voltage is provided, and the inductance of the two variable inductors is controlled by the control current. A current resonant converter characterized by being configured to control the output voltage to a constant value by changing the value and changing the resonance frequency.