JP2006033068A - Antenna and portable wireless device equipped with the antenna - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a dipole antenna having a radiating element with a U-turned tip and capable of defining the correlation between the dimension of the shape and the multiple resonance frequency.
An antenna includes a feeding point and a dipole radiating element. The dipole radiating element 2 is composed of two radiating elements, a first radiating element 3 and a second radiating element 4. The first radiating element 3 and the second radiating element 4 have a shape in which the open end is U-turned to the middle of the connection points 6 and 8, and the one-way dimension of the overlapping portions 5 and 7 that overlap in parallel by the U-turn is B . The parallel radiation element interval C is sufficiently smaller than B. Let A be the dimension of the non-overlapping portions at both ends of the feeding point 1. The total length when the first radiating element 3 and the second radiating element 4 are extended without making a U-turn is A + 4B + 2C. The total length − (B / 2) − (B / 2) and λ / 2 of the first resonance frequency on the low frequency side substantially coincide. The dimension B and λ / 4 of the second resonance frequency on the high frequency side substantially coincide with each other.
[Selection] Figure 1

Description

  The present invention relates to an antenna and a portable radio apparatus equipped with the antenna, and more particularly, to a dipole antenna capable of achieving multiple resonances and miniaturization and a portable radio apparatus equipped with the antenna.

  In recent years, in a portable wireless device, in order to support dual band and multimedia, it is essential to increase frequency and bandwidth.

(Background Technology 1)
There is a multi-resonant antenna having an appearance that can be seen as one antenna (see, for example, Patent Document 1). In the multi-resonant antenna of Patent Document 1, a monopole type linear conductor is U-turned to form a main conductor portion and an auxiliary conductor portion, and the distance between the main conductor portion and the auxiliary conductor portion is short. The multi-resonance characteristics are determined by the relationship between the amplitude and phase of the current flowing on the auxiliary conductor.

(Background Technology 2)
There is a composite antenna that transmits and receives signals in a multi-frequency band, which is two or more arbitrary frequency bands, using a single antenna element (see, for example, Patent Document 2). In the composite antenna of Patent Document 2, the antenna element is formed in a U-turn shape, and adjacent elements are capacitively coupled or inductively coupled to each other, so that the current pattern can be changed and the resonance frequency can be shifted. Specifically, the electrical length changes due to the coupling current due to coupling, the position of the portion where the current distribution becomes 0 changes, the basic resonance frequency f0 and 3 times 3f0 thereof are shifted, and two resonances of an arbitrary frequency Is realized.
Japanese Patent Application Laid-Open No. 11-127022 (pages 3 to 4, FIG. 2, FIG. 4 (a)) JP 2001-223519 A (pages 3 to 4, FIG. 1)

The multi-resonance antenna disclosed in the related art 1 is a monopole type and requires a ground plane (earth conductor plate). FIG. 2B of Patent Document 1 already has two resonances at 900 MHz and 2.2 GHz, regardless of the measurement result of the VSWR (voltage standing wave ratio) of the monopole antenna and the ground plane model. 2 (c) and 2 (d) show antennas using the background art. As shown in FIG. 2 (b), the resonance frequency depends on the size of the ground plane, and the main conductor portion and auxiliary conductor portion of the linear conductor. The size of each cannot be defined independently, and there is a problem that the design is difficult.
Further, in the composite antenna disclosed in the background art 2 of the related art, the feeding terminal and the tip of the U-turned antenna element are fixed at the same position, and the resonance frequency is adjusted by adjusting the coupling between adjacent elements. ing. However, there is no description about the specific correlation, and it is difficult to design. In addition, since the double frequency is a short circuit of a quarter wavelength, there is a problem that impedance becomes infinite and matching cannot be achieved. Therefore, it can be realized only around the triple frequency, which is an odd harmonic that is easy to match.

  The present invention has been made to solve the above-mentioned problems, and is a dipole antenna type that does not use a ground plane, and has a shape in which the tip of an antenna element is U-turned. An object is to provide an antenna capable of defining a correlation. Moreover, even if it is a dipole antenna, it aims at providing the antenna which does not require mounting space, and the portable radio apparatus carrying the said antenna.

  In order to achieve the above object, a dipole antenna of the present invention is a dipole antenna having a feeding point and a dipole radiating element, and the dipole radiating element includes a first radiating element and a second radiating element, One radiating element has one end as a connection point for receiving power supply, and the other end has a shape that makes a U-turn halfway in the direction of the connection point, and the one-way dimension of the U-turn overlap part is desired. The second resonance frequency is approximately a quarter wavelength, the adjacent interval between the U-turn overlap portions is sufficiently narrower than the one-way dimension, and the second radiating element has a connection point for receiving power at one end. The other end is U-turned halfway in the direction of the connection point, and the one-way dimension of the U-turn superposition part is approximately one-quarter wavelength of the second resonance frequency. The distance between adjacent parts is one-way When the first radiating element and the second radiating element are extended without making a U-turn, the total length of the radiating elements is lower than the second resonance frequency. A length obtained by adding approximately half of the one-way dimension of the first radiating element and approximately half of the one-way dimension of the second radiating element to a half wavelength of a resonance frequency, and the connection points of the dipole radiating elements are the feeding points. It is characterized by connecting to.

  According to the present invention, the tip of a dipole antenna type antenna element has a U-turn shape, and the correlation between the size of this shape and the multiple resonance frequency can be defined. Moreover, even if it is a dipole antenna, the shape which does not require mounting space can be obtained.

  Embodiments of the present invention will be described below with reference to the drawings.

FIGS. 1-11 is a figure explaining the dipole antenna which concerns on Example 1 of this invention.
FIG. 1 is a diagram illustrating a configuration of a dipole antenna that is a basis of the first embodiment. The antenna 100 includes a feeding point 1 and a dipole radiating element 2. Further, the dipole radiating element 2 is composed of two radiating elements, a first radiating element 3 and a second radiating element 4, and is disposed at both ends of the feeding point 1. The electrical dimensions of both ends viewed from the feeding point 1 are symmetric.
The first radiating element 3 has a shape in which one end is connected to the feeding point 1 with the connection point 6 and the other open end is U-turned halfway in the direction of the connection point 6 and overlaps in parallel by the U-turn. Let B be the one-way dimension of 5. The interval between the parallel radiating elements is C, and C is sufficiently smaller than B.
The second radiating element 4 has a shape in which one end is connected to the feeding point 1 with a connection point 8 and the other open end is U-turned halfway in the direction of the connection point 8 and overlaps in parallel by the U-turn. Let B be the one-way dimension of 7. The interval between the parallel radiating elements is C, and C is sufficiently smaller than B.
Let A be the dimension of the non-overlapping portions at both ends of the feeding point 1. When the first radiating element 3 and the second radiating element 4 are extended without making a U-turn, the total length of the dipole radiating element 2 is A + 4B + 2C.

FIG. 2 is a diagram for explaining a configuration of a dipole antenna having a three-dimensional shape that is easy to incorporate in the apparatus, and dimensions A, B, and C are the same as the dimensions of each part of the antenna 100 of FIG.
The antenna 110 of FIG. 2A is formed by bending the overlapping portions 5 and 7 in the minus Z direction with respect to the portion of the dimension A (X direction) that is a portion where the dipole radiating element 2 is not overlapped, so It is in shape. The dimension C is a three-dimensional structure in which the Y direction is set and the radiating elements configure X, Y, and Z directions. Thereby, it can be built in a narrow device such as a cellular phone. The material of the radiating element can be made of a sheet metal, a resin molded product formed with a three-dimensional circuit, or the like.

  In the antenna 111 of FIG. 2B, the non-overlapping portion of the dipole radiating element 2 is formed in a zigzag meander shape, and the effective electrical length is A. As a result, the mechanical dimension of the portion of the effective length A can be reduced, and it becomes easier to incorporate in the apparatus. The meander shape may be a spiral.

FIG. 3 is a diagram for explaining a configuration of a dipole antenna having a shape arranged on a common plane that is easy to be built in the apparatus, and dimensions A, B, and C are the same as the dimensions of each part of the antenna 100 of FIG. It is.
The antenna 120 shown in FIG. 3A is formed in a U shape by bending the overlapping portions 5 and 7 in a direction perpendicular to the portion of the dimension A, which is a portion where the dipole radiating element 2 is not overlapped. The open ends of the overlapping portions 5 and 7 are arranged outside. Since all the radiating elements are on a common plane, they can be made as conductors of the substrate. Thereby, it can also be incorporated in a thin device such as a cellular phone.
The antenna 121 in FIG. 3B is substantially the same as the antenna 120, and the different ends are arranged with the open ends of the overlapping portions 5 and 7 inside. Similarly, it can be incorporated in a thin device such as a cellular phone.

In the antenna 122 of FIG. 3C, the first radiating element 3 and the second radiating element 4 are arranged at right angles in an L shape with the feeding point 1 as a base point. Thereby, it becomes easy to incorporate in a corner portion of a personal computer or the like.
The antenna 123 in FIG. 3D has a shape in which a portion bent at a right angle in an L shape is slightly shifted. Similarly, it becomes easy to be incorporated in a corner portion of a personal computer or the like.
In addition, you may make the part of the dimension A which is a part which the dipole radiation element 2 of the antennas 120-123 does not overlap with meander shape.

  4 and 5 are simulation data on the VSWR (voltage standing wave ratio) of each antenna of Example 1, and the same data is obtained for each antenna. As a representative, for the antenna 110 (FIG. 2A) having a U-shaped three-dimensional shape, resonance frequency simulation data is shown with the antenna dimensions as parameters. This is a case where the dimension C is fixed to 2.5 mm.

FIG. 4 (a) shows sample no. 1 is the case where the dimensions of the antenna are A = 45 mm, B = 40 mm, and C = 2.5 mm, and the resonance frequency is a first resonance frequency of 920 MHz on the low frequency side and a second resonance frequency of 2000 MHz on the high frequency side. .
The half wavelength λ / 2 of the first resonance frequency 920 MHz is
λ / 2 = (3 × 10 8 [m / S] / 920 MHz) / 2 = 163 mm.
The quarter wavelength λ / 4 of the second resonance frequency of 2000 MHz is
λ / 4 = (3 × 10 8 [m / S] / 2000 MHz) /4=37.5 mm.

FIG. 4B shows sample No. 2, the dimensions of the antenna are A = 45 mm, B = 30 mm, and C = 2.5 mm. The resonance frequency is the first resonance frequency 1120 MHz (λ / 2 = 134 mm) on the low band side and the first on the high band side. Two resonance frequencies of 2540 MHz (λ / 4 = 29.5 mm) are generated.
FIG. 4 (c) shows sample no. 3, the antenna dimensions are A = 45 mm, B = 20 mm, and C = 2.5 mm. The resonance frequency is the first resonance frequency of 1420 MHz (λ / 2 = 106 mm) on the low band side and the first on the high band side. Two resonance frequencies of 3460 MHz (λ / 4 = 21.7 mm) are generated.
FIG. 5 (a) shows sample no. 4, the antenna dimensions are A = 35 mm, B = 40 mm, and C = 2.5 mm. The resonance frequency is the first resonance frequency 960 MHz (λ / 2 = 156 mm) on the low frequency side and the resonance frequency on the high frequency side. Two resonance frequencies of 2060 MHz (λ / 4 = 36.4 mm) are generated.
FIG. 5 (b) shows sample no. 5 is the case where the dimensions of the antenna are A = 25 mm, B = 40 mm, and C = 2.5 mm. As the resonance frequency, the first resonance frequency is 1020 MHz (λ / 2 = 147 mm) on the low band side, and the first is on the high band side. Two resonance frequencies of 2160 MHz (λ / 4 = 34.7 mm) are generated.

FIGS. 6 and 7 are simulation data for the VSWR of each antenna according to the first embodiment, and are obtained when the dimension C is varied.
6A shows a sample No. 6 is the case where the dimensions of the antenna are A = 45 mm, B = 40 mm, and C = 1 mm, and the resonance frequency is a first resonance frequency of 980 MHz (λ / 2 = 153 mm) on the low frequency side and a second resonance on the high frequency side. A frequency of 2000 MHz (λ / 4 = 37.5 mm) is generated.
FIG. 6B shows sample No. 7 is the case where the dimensions of the antenna are A = 45 mm, B = 40 mm, and C = 2 mm. The resonance frequency is the first resonance frequency 940 MHz (λ / 2 = 160 mm) on the low frequency side and the second resonance on the high frequency side. A frequency of 2000 MHz (λ / 4 = 37.5 mm) is generated.
FIG. 6C shows sample No. 8 is the case where the dimensions of the antenna are A = 45 mm, B = 40 mm, and C = 3 mm, and the resonance frequency is the first resonance frequency 900 MHz (λ / 2 = 167 mm) on the low frequency side and the second resonance on the high frequency side. A frequency of 2000 MHz (λ / 4 = 37.5 mm) is generated.
In FIG. 9, the antenna dimensions are A = 45 mm, B = 40 mm, and C = 4 mm. The resonance frequency is the first resonance frequency 900 MHz (λ / 2 = 170 mm) on the low frequency side and the second resonance on the high frequency side. A frequency of 2000 MHz (λ / 4 = 37.5 mm) is generated.
FIG. 7B shows sample No. 10 is the case where the dimensions of the antenna are A = 45 mm, B = 40 mm, and C = 5 mm. The resonance frequency is the first resonance frequency 840 MHz (λ / 2 = 179 mm) on the low frequency side and the second resonance on the high frequency side. A frequency of 2000 MHz (λ / 4 = 37.5 mm) is generated.
In FIG. 11 is the case where the dimensions of the antenna are A = 45 mm, B = 40 mm, and C = 6 mm. The resonance frequency is the first resonance frequency 820 MHz (λ / 2 = 183 mm) on the low frequency side and the second resonance on the high frequency side. A frequency of 2000 MHz (λ / 4 = 37.5 mm) is generated.
FIG. 8 summarizes the simulation data (C = fixed) of FIGS. 4 and 5 in a table. In the antenna dimension column, dimensions A, B, and C, and the total length (= A + 4B + 2C) and the total length − (B / 2) − (B / 2) when the dipole radiating element 2 is extended are shown. The simulation data column shows the resonance frequency and wavelength, which are the results of the simulation. As can be seen, the total length is slightly longer than λ / 2 of the first resonance frequency on the low frequency side, and is approximately 1.23 to 1.29 times λ / 2. In addition, the total length − (B / 2) − (B / 2) and λ / 2 of the first resonance frequency on the low frequency side substantially coincide (approximately 1.02 to 1.04 times). That is, the total length is approximately (B / 2) + (B / 2) longer than λ / 2 of the first resonance frequency on the low frequency side. In addition, the dimension B and the λ / 4 of the second resonance frequency on the high frequency side are approximately the same (approximately 0.92 to 1.15).

  FIG. 9 summarizes the simulation data of FIGS. 6 and 7 (C = 1 to 6 mm variable) in a table. The total length is slightly longer than λ / 2 of the first resonance frequency on the low frequency side, and is approximately 1.19 to 1.35 times λ / 2. Further, the total length − (B / 2) − (B / 2) and λ / 2 of the first resonance frequency on the low frequency side substantially coincide (approximately 0.97 to 1.09 times). That is, the total length is approximately (B / 2) + (B / 2) longer than λ / 2 of the first resonance frequency on the low frequency side. Further, the dimension B and the λ / 4 of the second resonance frequency on the high frequency side are substantially the same (approximately 1.07).

  Here, there is regularity in the change of the dimension C and the change of λ / 2 of the first resonance frequency. As the dimension C increases from 1 mm to 6 mm, the total length / (λ / 2) sequentially increases from 1.35 to 1. .19, and (total length− (B / 2) − (B / 2)) / (λ / 2) gradually decreases from 1.09 to 0.97. This is considered that the inductive influence becomes smaller as the distance between the radiating elements in the U-turn overlapped portion becomes wider, and approaches the direction in which the total length and λ / 2 of the first resonance frequency coincide. Therefore, the first resonance frequency can be adjusted by changing the dimension C.

  FIG. 10 is a current distribution diagram of the second resonance frequency on the high frequency side, which is considered based on the simulation data tables described in FIGS. 8 and 9. The current vector is indicated by an arrow line. In the current distribution of λ / 4 of the second resonance frequency on the high frequency side, a short circuit of λ / 4 is formed by the overlapping portion, the current becomes maximum at the U-turn point that is the short circuit point, and the virtual feeding points 1a and 1b are It is thought that it appeared.

FIG. 11 is a current distribution diagram of the first resonance frequency on the low frequency side, and is represented by the ratio of the antenna dimension and λ / 2 in the simulation data of FIG.
In sample 6 (C = 1 mm), “total length− (B / 2) − (B / 2)” is slightly larger than λ / 2, and in sample 8 (C = 3 mm), it almost matches, and sample 11 (C = 6 mm) Then, “total length− (B / 2) − (B / 2)” is slightly smaller than λ / 2. However, in both cases, when the distance C between the radiating elements is sufficiently smaller than the one-way distance B of the U-turn overlapped portion, “total length− (B / 2) − (B / 2)” and λ / 2 substantially coincide with each other. I can say that.
As the radiating element interval C increases, the influence of the inductive property of the U-turn overlap portion decreases, and the length approaches the direction in which λ / 2 of the first resonance frequency matches.
According to the first embodiment of the present invention, when the second resonance frequency on the high frequency side and the first resonance frequency on the low frequency side required by the device on which the antenna is mounted are determined, the antenna dimensions can be determined accordingly. . As an order, the antenna dimension B is determined in accordance with λ / 4 of the required second resonance frequency on the high frequency side. Next, the “full length− (B / 2) − (B / 2)” (= A + 3B + 2C) of the antenna dimensions is determined according to λ / 2 of the required first resonance frequency on the low frequency side, and the dimension A is determined. be able to.

FIGS. 12-14 is a figure explaining the structure of the dipole antenna based on Example 2 of this invention.
FIG. 12 is a diagram illustrating the configuration of the dipole antenna according to the second embodiment. The differences from the first embodiment will be mainly described. The dimensions of the overlapped portion of the U-turn are different from each other, the overlapped portion 5 of the first radiating element 3 is dimension B, and the overlapped portion 7 of the second radiating element 4 is dimension D. In addition, although the superposition | polymerization part 5 and the superposition | polymerization part 7 are made into right angle with respect to the part of the dimension A which has not superposed | polymerized, it is not limited to this shape, but each antenna shown in Example 1 The same shape may be sufficient as long as the dimensions of the overlapping portion 5 and the overlapping portion 7 are different.

  FIG. 13 is simulation data on the VSWR (voltage standing wave ratio) of the antenna of the second embodiment. The antenna dimensions are A = 45 mm, B = 40 mm, D = 30 mm, and C = 2.5 mm, and three resonances occur. A resonance with a first resonance frequency of 1000 MHz (λ / 2 = 150 mm) on the low frequency side, a second resonance frequency of 1720 MHz (λ / 4 = 44 mm), and a third resonance frequency of 2280 MHz (λ / 4 = 33 mm) on the high frequency side occurs. To do.

  FIG. 14 summarizes the simulation data of FIG. 13 in a table. In the antenna dimension column, dimensions B and D, and the total length (= A + 2B + 2D + 2C) and the total length − (B / 2) − (D / 2) when the dipole radiating element 2 is extended are shown. The simulation data column shows the resonance frequency and wavelength, which are the results of the simulation. As can be seen from this, the total length − (B / 2) − (D / 2) is substantially equal to λ / 2 of the first resonance frequency on the low frequency side. The dimension B and the second resonance frequency λ / 4 substantially coincide with each other. In addition, the dimension D is substantially equal to λ / 4 of the third resonance frequency on the high frequency side. This is presumably because a current distribution is generated for the same reason as in the first embodiment.

  According to the second embodiment of the present invention, when the three resonance frequencies required by the device on which the antenna is mounted are determined, the antenna dimensions can be determined accordingly.

FIGS. 15-17 is a figure explaining the structure of the dipole antenna based on Example 3 of this invention.
FIG. 15 is a diagram illustrating the configuration of the dipole antenna according to the third embodiment. The differences from the first embodiment will be mainly described. The antenna 140 shown in FIG. 15A is provided with a U-turn overlap portion 5 only in the first radiating element 3, the overlap portion 5 has a dimension B, and the non-overlap portion has a dimension A. In addition, although the superposition | polymerization part 5 is made into the right angle with respect to the part of the dimension A which has not superposed | polymerized, it is not restricted to this, The shape similar to each antenna shown in Example 1 may be sufficient, Only one is sufficient. Further, as shown in FIG. 15B, the second radiating element 4 may be refracted.

  FIG. 16 is simulation data on the VSWR (voltage standing wave ratio) of the antenna of the third embodiment. The antenna dimensions are A = 45 mm, B = 40 mm, and C = 2.5 mm, and two resonances occur. A first resonance frequency of 1320 MHz (λ / 2 = 114 mm) on the low frequency side and a second resonance frequency of 2160 MHz (λ / 4 = 35 mm) on the high frequency side are generated.

  FIG. 17 summarizes the simulation data of FIG. 16 in a table. In the column of the antenna dimension, a dimension B, a total length (= A + 2B + C), and a total length − (B / 2) (= A + 2B + C− (B / 2)) are shown. The simulation data column shows the resonance frequency and wavelength, which are the results of the simulation. As can be seen from this, the total length − (B / 2) is substantially equal to λ / 2 of the first resonance frequency on the low frequency side. In addition, the dimension B and the λ / 4 of the second resonance frequency on the high frequency side are substantially the same. This is presumably because a current distribution is generated for the same reason as in the first embodiment.

  According to the third embodiment of the present invention, when two resonance frequencies necessary for a device equipped with an antenna are determined, the antenna dimensions can be determined in accordance with the two resonance frequencies.

18-20 is a figure explaining the structure of the dipole antenna based on Example 4 of this invention.
FIG. 18 is a diagram illustrating the configuration of the dipole antenna according to the fourth embodiment. The differences from the first embodiment will be mainly described. The antenna 150 is newly provided with a parasitic element 9. This position is arranged in the vicinity of the feeding point 1 and in parallel with the portion of the dimension A that does not overlap. The dipole radiating element 2 may have the same shape as each antenna shown in the first embodiment.

FIG. 19 is simulation data on the VSWR (voltage standing wave ratio) of the antenna of the third embodiment. FIG. 19A shows a case where there is no parasitic element, and the case where the antenna dimensions are A = 45 mm, B = 20 mm, and C = 2.5 mm, and two resonances occur. A first resonance frequency of 1420 MHz (λ / 2 = 106 mm) on the low frequency side and a second resonance frequency of 3460 MHz (λ / 4 = 21.7 mm) on the high frequency side are generated.
FIG. 19B shows a case with a parasitic element, where the distance between the parasitic element and the feeding point 1 is 1.5 mm. It can be seen that the band in the vicinity of the second resonance frequency of 3460 MHz on the high frequency side is wider than in the case without the parasitic element in FIG. This is because resonance is generated at a frequency at which the parasitic element P = 37 mm is λ / 2, and multiple resonance is performed on the high frequency side. Thereby, it is possible to realize a wide band as well as multiple resonances.

  FIG. 20 is a table summarizing simulation data in the case of no parasitic element shown in FIG. Sample No. 1 in FIG. 3 and the description is omitted.

FIGS. 21-24 is a figure explaining the structure of the dipole antenna which concerns on Example 5 of this invention.
FIG. 21 is a diagram illustrating the configuration of the dipole antenna according to the fifth embodiment. FIG. 21A shows a structure in which radiating elements are arranged on a common plane, FIG. 21B shows a three-dimensional structure, and the functions are the same, and will be described below with reference to FIG. .

The antenna 160 in FIG. 21A includes a feeding point 1, a first dipole radiating element 12, a second dipole radiating element 22, and a loop element 21. The dimension of the loop element 21 in the Z-axis direction = F + G, and the dimension in the X-axis direction is H.
The structure of the first dipole radiating element 12 is the same shape as the dipole radiating element shown in the first to third embodiments. The first radiating element 13 and the second radiating element 14 are composed of two radiating elements. The first radiating element 13 has an overlapping portion 15 (one-way dimension E) and a connection point 16, and the second radiating element 14 is overlapped. It has a portion 17 (one-way dimension E) and a connection point 18. The dimension of the non-overlapping portions at both ends of the feeding point 1 is D.
The structure of the second dipole radiating element 22 has the same shape as the dipole radiating element shown in the first to third embodiments. The first radiating element 23 and the second radiating element 24 are composed of two radiating elements. The first radiating element 23 has a superposition part 25 (one-way dimension B) and a connection point 26, and the second radiating element 24 is superposed. A portion 27 (one-way dimension B) and a connection point 28 are included. The dimension of the non-overlapping portions at both ends of the feeding point 1 is A.

  The connection point 16 and the connection point 18 of the first dipole radiating element 12 are connected to the feeding point 1. The loop element 21 is connected to the feeding point 1. The connection point 26 and the connection point 28 of the second dipole radiating element 22 are connected between the dimension F and the dimension G in the Z-axis direction of the loop element 21 and are connected to the feeding point 1 via the loop element 21. The The parallel distance between the first dipole radiating element 12 and the second dipole radiating element 22 is the dimension F.

FIG. 22 is a diagram for explaining the function of the loop element 21. A first dipole radiating element 12 and a loop element 21 are shown. The current path L1 indicates a current path that is dominant in series resonance, and the current path L2 indicates a current path that is dominant in parallel resonance, and these two resonances are dominant in the frequency at which each length is a half wavelength.
The current path L1 is a mode that resonates using a path that passes through the feeding point 1 in the same manner as an ordinary dipole antenna, and maximizes the current value at the feeding point. Therefore, the current path L1 is a path that generates series resonance. On the other hand, the current path L2 is a mode that resonates without passing through the feed point 3, and minimizes the current value at the feed point. Therefore, the current path L2 is a path that generates parallel resonance. The shape of the loop element 21 is referred to as a folded structure.
In the folded structure of the loop element 21, there is a difference between the lengths of the current paths L1 and L2, thereby determining the frequency at which series resonance and parallel resonance occur. When the current paths L1 and L2 are close to each other, the resonance frequencies of the series resonance and the parallel resonance approach, and the current of the series resonance when there is no folded structure is also generated in the parallel resonance current. The impedance can be increased by decreasing. That is, when the lengths of the current paths L1 and L2 approach each other, it is considered that not only the series resonance but also the influence of the parallel resonance is increased and the impedance is increased.
Therefore, by arranging another second dipole radiating element 22 via the loop element 21, the first dipole radiating element 12 and the second dipole radiating element 22 can be adjusted independently.

FIG. 23 shows simulation data for the VSWR (voltage standing wave ratio) of the antenna shown in FIG. 21, and multiple resonances are independently caused by the dimensions of the first dipole radiating element 12 and the second dipole radiating element 22. The occurrence will be described below.
The dimensions of the first dipole radiating element 12 are D = 25 mm, E = 10 mm, and C = 2.5 mm. The dimensions of the second dipole radiating element 22 are A = 45 mm, B = 40 mm, and C = 2.5 mm. The dimensions of the loop-loop element 21 part are F = 5 mm, G = 10 mm, and H = 5 mm. As a resonance frequency, a first resonance frequency 910 MHz, a second resonance frequency 1960 MHz, a third resonance frequency 2470 MHz, and a fourth resonance frequency 6520 MHz are generated in order from the low frequency side.

FIG. 24 summarizes the simulation data of FIG. 23 in a table. FIG. 24A is a table related to the first dipole radiating element 12, and FIG. 24B is a table related to the second dipole radiating element 22.
As shown in FIG. 24A, the total length − (E / 2) − (E / 2) of the first dipole radiating element 12 and λ / 2 of the third resonance frequency 2470 MHz substantially coincide. The dimension E of the overlapping portions 15 and 17 and λ / 4 of the fourth resonance frequency 6520 MHz substantially coincide.
As shown in FIG. 24B, the total length − (B / 2) − (B / 2) of the second dipole radiating element 22 and λ / 2 of the first resonance frequency 910 MHz substantially coincide. The dimension B of the overlapping portions 25 and 27 and λ / 4 of the second resonance frequency of 1960 MHz substantially coincide.
Note that the resonance of 5080 MHz shown in FIG. 23 is 3λ / 4 = 44 mm, which is a third harmonic of the dimension B of the second dipole radiating element 22. Multiple resonances may be made using this odd multiple harmonic.

  As shown in FIGS. 24A and 24B, the first dipole radiating element 12 and the second dipole radiating element 22 indicate that the resonance frequency can be adjusted independently.

  The shape of the dipole radiating element of the fifth embodiment is not limited to FIGS. 21A and 21B, and any shape may be used as long as the antennas of the first to fourth embodiments are coupled by the loop element 21.

25 and 26 are external views of an apparatus equipped with a dipole antenna according to Embodiment 6 of the present invention.
FIG. 25 shows a state where the antenna 110 (FIG. 2) is mounted inside the mobile phone 200. The antenna 110 has a U-shape and can be mounted even on a device having a narrow width in the X direction, such as the mobile phone 200. If the antennas 120 and 121 (FIG. 3) are mounted, the mobile phone 200 can be mounted even if the thickness of the mobile phone 200 in the Y direction is thin. The antenna to be mounted may be another U-shaped antenna. The antenna to be mounted may be any antenna described in the first to fifth embodiments as long as the mobile phone 200 has a mounting space.

  FIG. 26 shows a state in which the antenna 122 (FIG. 3C) is mounted inside the personal computer 300. The antenna 122 is L-shaped, and can be mounted on the corner portion of the lower casing 302 of the personal computer 300 while avoiding other mounting components (FIG. 26A). Note that other corners of the upper housing 301 may be avoided (FIG. 26B). The antenna to be mounted may be another L-shaped antenna. The antenna to be mounted may be any antenna described in the first to fifth embodiments as long as the personal computer 300 has a mounting space. Further, a plurality of antennas may be installed to perform selection diversity.

  In the sixth embodiment, the cellular phone and the personal computer are shown, but other portable wireless devices may be used.

1 is a configuration diagram of a basic dipole antenna according to Embodiment 1 of the present invention. FIG. 1 is a configuration diagram of a common three-dimensional dipole antenna according to Embodiment 1 of the present invention. FIG. The block diagram of the dipole antenna of the shape arrange | positioned on the common plane which concerns on Example 1 of this invention. The simulation figure (C fixation) (1/2) which concerns on Example 1 of this invention. The simulation figure (C fixation) which concerns on Example 1 of this invention (2/2). The simulation figure (C variable) which concerns on Example 1 of this invention (1/2). The simulation figure (C variable) which concerns on Example 1 of this invention (2/2). The simulation table | surface (C fixation) which concerns on Example 1 of this invention. The simulation table | surface (C variable) which concerns on Example 1 of this invention. The current distribution figure of the 2nd resonance frequency of the high region side concerning Example 1 of the present invention. FIG. 3 is a current distribution diagram of a first resonance frequency on the low frequency side according to the first embodiment of the present invention. The block diagram of the dipole antenna which concerns on Example 2 of this invention. The simulation figure which concerns on Example 2 of this invention. The simulation table | surface which concerns on Example 2 of this invention. The block diagram of the dipole antenna which concerns on Example 3 of this invention. The simulation figure which concerns on Example 3 of this invention. The simulation table | surface which concerns on Example 3 of this invention. The block diagram of the dipole antenna which concerns on Example 4 of this invention. The simulation figure which concerns on Example 4 of this invention. The simulation table | surface which concerns on Example 4 of this invention. The block diagram of the dipole antenna which concerns on Example 5 of this invention. The figure explaining the loop function which concerns on Example 5 of this invention. The simulation figure which concerns on Example 5 of this invention. The simulation table | surface which concerns on Example 5 of this invention. The external view of the mobile telephone carrying the dipole antenna which concerns on Example 6 of this invention. The external view of the personal computer carrying the dipole antenna which concerns on Example 6 of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Feeding point 2 Dipole radiation element 3 1st radiation element 4 2nd radiation element 5, 7 Superposition | polymerization part 6, 8 Connection point 9 Parasitic element 12 1st dipole radiation element 13 1st radiation element 14 2nd radiation elements 15 and 17 Superposition part 16, 18 Connection point 21 Loop element 22 Second dipole radiation element 23 Third radiation element 24 Fourth radiation element 25, 27 Superposition part 26, 28 Connection point 100, 110, 111, 120, 121, 122, 123, 130, 140, 150, 160, 161 Antenna 200 Mobile phone 300, 301 Personal computer

Claims (12)

A dipole antenna having a feeding point and a dipole radiating element,
The dipole radiating element includes a first radiating element and a second radiating element;
The first radiating element includes:
One end is used as a connection point for receiving power supply, and the other end is shaped to make a U-turn partway in the direction of the connection point. The one-way dimension of the U-turn overlap part is the desired second resonance frequency. And the adjacent interval of the U-turn overlap part is sufficiently narrower than the one-way dimension,
The second radiating element is
One end is used as a connection point for receiving power, and the other end is shaped to overlap by making a U-turn partway along the connection point. The one-way dimension of the U-turn overlap part is the second resonance frequency. The wavelength is set to about a quarter wavelength, and the adjacent interval between the U-turn overlapping portions is sufficiently narrower than the one-way dimension.
The total length of the radiating elements when the first radiating element and the second radiating element are extended without making a U-turn is a half wavelength of a desired first resonance frequency that is lower than the second resonance frequency in the previous period. A length obtained by adding approximately half of the one-way dimension of the first radiating element and approximately half of the one-way dimension of the second radiating element;
A dipole antenna characterized in that each connection point of the dipole radiating element is connected to the feeding point. A dipole antenna having a feeding point and a dipole radiating element,
The dipole radiating element includes a first radiating element and a second radiating element;
The first radiating element includes:
One end is used as a connection point for receiving power, and the other end is U-turned halfway along the connection point. The one-way dimension of the U-turn overlap part is the desired third resonance frequency. And the adjacent interval of the U-turn overlap part is sufficiently narrower than the one-way dimension,
The second radiating element is
One end is used as a connection point for receiving power supply, and the other end is shaped to make a U-turn partway in the direction of the connection point. The one-way dimension of the U-turn overlap part is the desired second resonance frequency. And the adjacent interval of the U-turn overlap part is sufficiently narrower than the one-way dimension,
When the first radiating element and the second radiating element are extended without making a U-turn, the total length of the radiating elements is lower than the third resonance frequency and the second resonance frequency. A length obtained by adding approximately half of the one-way dimension of the first radiating element and approximately half of the one-way dimension of the second radiating element to a half wavelength of a resonance frequency;
A dipole antenna characterized in that each connection point of the dipole radiating element is connected to the feeding point. A dipole antenna having a feeding point and a dipole radiating element,
The dipole radiating element includes a first radiating element and a second radiating element,
The first radiating element includes:
One end is used as a connection point for receiving power supply, and the other end is shaped to make a U-turn partway in the direction of the connection point. The one-way dimension of the U-turn overlap part is the desired second resonance frequency. And the adjacent interval of the U-turn overlap part is sufficiently narrower than the one-way dimension,
The second radiating element is
One end is a connection point for receiving power, and the other end is extended or refracted without making a U-turn,
When the first radiating element is stretched or refracted without making a U-turn, the total length of the radiating elements when combined is a half wavelength of a desired first resonance frequency that is lower than the second frequency in the previous period. The length plus approximately half of the one-way dimension of the element,
A dipole antenna characterized in that each connection point of the dipole radiating element is connected to the feeding point.   The dipole antenna according to any one of claims 1 to 3, further comprising a parasitic element disposed in the vicinity of the feeding point and in close proximity to the dipole radiating element. A dipole antenna having a feeding point, a first dipole radiating element, a second dipole radiating element, and a loop-shaped element,
The first dipole radiating element according to any one of claims 1 to 3, wherein each connection point of the dipole radiating element is connected to the feeding point,
The second dipole radiating element is the dipole radiating element according to any one of claims 1 to 3,
The loop-shaped element is connected to the feeding point, and includes two connection points for connecting to the second dipole radiating element,
A dipole antenna characterized in that each connection point of the second dipole radiating element is connected to two connection points of the loop-shaped element.   6. The dipole antenna according to claim 5, further comprising a parasitic element disposed in the vicinity of the feeding point and in close proximity to the second dipole radiating element.   The dipole antenna according to any one of claims 1 to 6, wherein the first radiating element and the second radiating element of the dipole radiating element are arranged on a common plane.   The dipole antenna according to claim 1, wherein the first radiating element and the second radiating element of the dipole radiating element have a three-dimensional shape.   The dipole antenna according to any one of claims 1 to 8, wherein the dipole radiating element has a U-shape.   The dipole antenna according to any one of claims 1 to 8, wherein the dipole radiating element has an L shape.   A portable radio apparatus comprising the dipole antenna according to claim 9.   A portable radio apparatus comprising the dipole antenna according to claim 10.

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