HK1169737B - Wireless sensor reader - Google Patents

Description

Wireless sensor reader

Cross Reference to Related Applications

This non-provisional application is a continuation-in-part application of U.S. patent application 12/419,326 filed on 7/4/2009, U.S. patent application 12/419,326 is a continuation-in-part application of U.S. patent application 12/075,858 filed on 14/3/2008, and U.S. patent application 12/075,858 claims priority to U.S. provisional application 60/918,164 filed on 15/3/2007, wherein the entire contents of each application are incorporated herein by reference.

Technical Field

The present invention relates generally to reading passive wireless sensors, and more particularly to reader circuits and methods for stimulating and sensing data from passive wireless readers.

Background

Passive wireless sensor systems employing resonant circuit technology are known. These systems utilize passive wireless sensors to communicate remotely with excitation and reader circuitry. The wireless sensor is often implanted at a specific location, such as within the human body, to detect and report the sensed parameter. The sensed parameter causes a resonant circuit frequency of the wireless sensor to change. The reader device samples the resonant frequency of the wireless sensor to determine the sensed parameter.

Early researchers, Haynes (h.e. Haynes and a.l. witches, "Medical electronics, the pill that is 'talk'", RCA engine, vol 5, pp.52-54.1960, disclosed swallowable tablets containing wireless pressure sensors, in which a large reader device was wrapped around the body of a subject and the frequency was measured using a discriminator circuit. Nagumo (j.nagumo, a.uchiyama, s.kimoto, t.watanuki, m.hori, k.suma, a.ouchi, m.kumano and h.watanabe, "Echo capsule for use in a Medical application", IRE Transactions on Bio-Medical Electronics, vol.bme-9, pp.195-199, 1962) discloses a similar system, wherein the sensor comprises an energy storage capacitor to power the sensor during resonance.

U.S. patent 4,127,110 to Bullara discloses a sensor for measuring cerebral fluid pressure. U.S. patent 4,206,762 to Cosman discloses a similar sensor for measuring intracranial pressure. Specifically, the Cosman patent describes using a grid dip system for wirelessly measuring the resonant frequency of the sensor.

Several methods of reading passive wireless sensors have also been described in prior patents. For example, the Cosman patent discloses an external oscillator circuit tuned using an implanted sensor and a grid dip measurement system for making measurements of the resonant frequency of the sensor. U.S. patent 6,015,386 to Kensey et al discloses a reader that excites a passive sensor by transmitting a frequency sweep and uses a phase detector on the transmitted signal to identify the point during the sweep where the transmitted frequency coincides with the resonant frequency of the sensor. U.S. patent 6,206,835 to spilman et al discloses medical implant applications for the reader technology disclosed in U.S. patent 5,581,248 to spilman et al. The reader technology utilizes the sensed parameters of the sensor to detect frequency dependent variable impedance loading effects on the reader. Us patent 7,432,723 to Ellis et al discloses a reader having supply loops that are tuned and spaced apart at different frequencies transmitted by the supply loops in a manner that ensures that the bandwidth of the sensor allows resonant excitation of the sensor. Ellis uses the ring down response from a suitable supply loop to determine the sensor resonant frequency. Allen et al, U.S. patent 6,111,520, discloses a method of sending a "chirp" of white noise to a sensor and detecting a ring down response.

Some readers utilize phase-locked loop ("PLL") circuitry to lock onto the resonant frequency of the sensor. U.S. patent 7,245,117 to Joy et al discloses an active PLL circuit and a signal processing circuit for adjusting the transmit PLL frequency until the receive signal phase and the transmit PLL signal phase match. When this matching occurs, the transmit PLL frequency is equal to the sensor resonant frequency.

The PLL circuit may contain a sample-and-hold (S/H) function for sampling the input frequency and holding the PLL at a given frequency. PLLs with S/H can be used in a variety of applications. For example, U.S. patent 4,531,526 to Genest discloses a reader that uses a PLL circuit with an S/H circuit to adjust the reader' S transmit frequency to match the resonant frequency received from the sensor. This operation is performed to maximize the sensor response for the next transmission, and the decay rate of the sensor resonance amplitude is measured to extract the sensing parameter value. U.S. patent 4,644,420 to Buchan describes a PLL with S/H for sampling a tape data stream and maintaining an appropriate sampling frequency to evaluate digital data pulses on the tape. Additional improvements to this concept are provided by Buchan et al, U.S. patent 5,006,819. Denny, U.S. patent 5,920,233, describes a high speed sampling technique that uses an S/H circuit with a PLL to reduce charge pump noise from the phase frequency detector, thereby enhancing the low jitter performance of the frequency synthesizing circuit. U.S. patent 4,511,858 to Charavit et al discloses a PLL with an S/H circuit for predetermining the position of the control voltage of a voltage controlled oscillator when the PLL locking frequency is being changed. This is done to increase the response speed of the PLL when changing the desired synthesis frequency. Fischer U.S. patent 6,570,457 and Fischer et al U.S. patent 6,680,654 disclose PLLs with S/H circuits for improving PLL frequency stepping and offset correction features. U.S. patent 3,872,455 to Fuller et al discloses a PLL with a number S/H for freezing the frequency display and preloading the frequency counter when PLL phase lock is detected.

Readers have also been found for implementing direct signal sampling and frequency analysis techniques. One example is Eggers et al, U.S. patent 7,048,756, in which a resonant sensor having a Curie temperature to show a change in response at a temperature threshold is used to measure the temperature in the body.

In addition, readers are known that use digital signal analysis to improve performance and response. Us patent 7,466,120 to Miller et al describes the use of a Digital Signal Processor (DSP) to evaluate the response of a passive blood pressure sensor that has been excited with frequency pulses, and then to evaluate the response signal from the three-frequency excitation for relative phase delay.

Current passive sensor reader designs such as those described above suffer from a number of drawbacks. The early Haynes and Nagumo "pulse-echo ringing systems" required large, high-power reader devices. In addition, Collins (c. Collins, "Miniature professional pressure transducer for imaging in the Eye", ieee transactions on Bio-Medical Engineering, vol BME-14, No.2, April 1967) discloses that these systems are subject to inaccuracies and poor resolution due to difficulties in measuring the frequency of transient ringing signals, resulting in the replacement of these systems by various frequency sweeping methods.

Swept-frequency sensor readers similar to the techniques described in the Cosman, Kensey, Ellis, and spilman patents, as well as the pulse method described by Allen, require government agencies that regulate radio transmissions to allow for relatively wide bandwidths. This limits other uses of the spectrum and makes interference a potential problem. Similar problems exist with readers such as Genest, Ellis, and Joy for tracking the resonant frequency of passive resonant sensors using variable frequency transmitters. The additional circuitry required for the frequency sweep method and/or the digital tracking method is extremely large, thereby increasing reader size, cost, and failure rate. Furthermore, the amount of power required to transmit, signal process, sample and track the resonant frequency of the sensor using a digitally controlled frequency tracking or sweeping system is significant, and limits the ability to use battery power in the reader and limits the life of the battery within the battery powered reader. Accordingly, there is a need in the art for improved passive sensors and reader systems.

Drawings

The detailed description is made in conjunction with the following drawings:

FIG. 1 is a block diagram of a passive wireless sensor system;

FIG. 2 is a flow chart illustrating a process of taking readings from a sensor;

FIG. 3 is a graph qualitatively illustrating frequency characteristics of signal exchange between a sensor and a reader;

4A, 4B, and 4C are three sequential graphs that qualitatively illustrate the frequency characteristics of the exchange of signals between the sensor and the reader during reading acquisition;

FIG. 5 is a block diagram of the passive wireless sensor system of FIG. 1 expanded to include external data interface functionality and remote data processing functionality;

FIG. 6 is a block diagram of the passive wireless sensor system of FIG. 1 with the addition of an intermediate antenna;

FIG. 7 is a top level block diagram of the internal circuitry of the reader;

FIG. 8 is a block diagram of the timing and control portion of the reader circuit;

FIG. 9 is a block diagram of a transmitting portion of a reader circuit;

FIG. 10 is a block diagram of a receiving portion of a reader circuit;

FIG. 11 is a block diagram of a phase locked loop portion of a reader circuit;

FIG. 12 is a block diagram of a frequency counter portion of a reader circuit;

FIG. 13 is a block diagram of an alternative embodiment of a phase locked loop portion of the reader circuit shown in FIG. 11 with a digital sample timer and generation function for implementing sample and hold;

FIG. 14 is a block diagram showing an alternative embodiment of the reader internal circuitry of FIG. 7, in which the PLL and frequency counter are replaced with a digital sampling circuit and a spectrum analysis circuit;

FIG. 15 is a block diagram illustrating an alternative embodiment of the timing and control circuit of FIG. 8, in which the PLL timer and the spectrum counter timer are replaced with a digital sample timer and a spectrum analysis timer, respectively;

FIG. 16 is a block diagram of the internal architecture of the digital sampling circuit block of FIG. 14; and

fig. 17 is a block diagram of the internal architecture of the spectrum analysis circuit block of fig. 14.

Disclosure of Invention

A reader device is provided to interface with a wireless sensor, wherein the resonant frequency of the wireless sensor changes in proportion to a sensed parameter. The reader transmits a short pulse of energy at a fixed frequency to ring the wireless sensor at or near its resonant frequency immediately after the end of the transmission. The reader receives and amplifies the sensor ring signal and measures its frequency. In one embodiment, the reader performs this measurement by sending the signal to a phase locked loop ("PLL") that locks to the sensor ring frequency. Once the PLL has locked to the ring frequency, a voltage controlled oscillator ("VCO") of the PLL is placed in a hold mode to maintain the VCO frequency at the locked frequency. The VCO frequency is counted to determine the sensor resonant frequency. Optionally, the VCO control voltage itself is sampled and used to determine the sensor resonant frequency based on a known correlation. When sampling the VCO control voltage, there is no need to lock the VCO frequency if the voltage sampling is fast enough. Other frequency determination methods and systems involving digital spectral analysis are also disclosed.

Detailed Description

A passive wireless sensor system is provided, wherein the passive wireless sensor system includes a reader 10 in remote communication with a sensor 12. Reader 10 is capable of exciting sensor 12 (see fig. 1) by transmitting a signal, such as a radio frequency ("RF") pulse, at or near the resonant frequency of sensor 12. The sensor 12 may emit a ring signal for a short period of time in response to an excitation pulse from the reader 10.

The sensor 12 may be a passive device that does not itself contain a power source and is capable of emitting a ring signal 16 in response to the excitation signal 14 being at or near the resonant frequency of the sensor 12. The sensor 12 may be configured to sense a particular parameter. For example, the sensor 12 may include a fixed inductor 13 and a capacitor 15 that varies based on the sensed parameter. The changing capacitance or inductance changes the resonant frequency of the sensor 12. However, it should be understood that sensor 12 may be any wireless sensor known in the art capable of communicating remotely with reader 10. Further, although the sensor 12 is described as an RF resonant sensor, it should be understood that the sensor 12 may be an acoustic resonant sensor, an optical resonant sensor, or other similar sensors known in the art. The reader 10 may use a corresponding signal to activate the sensor 12. Further, the sensor 12 may be an active sensor or a passive sensor.

In an embodiment, the sensor 12 comprises at least one inductive element 13 and one capacitive element 15. To change the resonant frequency of the sensor 12 in proportion to the sensed parameter, the inductive element 13 or the capacitive element 15, or both, may be configured to change the inductance or capacitance in proportion to the sensed parameter. In the exemplary embodiment shown in fig. 1, the capacitive element 15 is variable and the inductive element 13 is fixed. Typical examples of such components are sensors that change capacitance in response to changes in pressure. These capacitive pressure sensors are well known in the art.

In one embodiment, at least one inductive element 13 in sensor 12 also functions as an antenna for sensor 12 to couple energy with another antenna 26 located on reader 10.

Reader 10 may energize sensor 12 by sending an excitation pulse 14 in the vicinity of sensor 12. For example, the reader may emit an RF excitation pulse 14 at or near the resonant frequency of the sensor 12. The sensor 12 may emit a ring signal 16 in response to the excitation pulse 14. The reader 10 may determine the frequency of the ringing signal 16 to determine the value of the sensed parameter.

FIG. 2 is a flow chart showing an example of steps that may be involved in the process of reader 10 taking a reading from sensor 12. The steps may include multiple interleaved steps and the steps may be interleaved in several layers. However, to clarify the sequence of operation of the reader during read acquisition, only the basic top-level steps are shown. In an initial condition 202, the sensor 12 has been configured such that its resonant frequency is proportional to the sensed parameter. Some examples of sensed parameters that may be measured using capacitive or inductive sensors include pressure, temperature, acceleration, angular velocity, PH level, glucose level, salinity, viscosity, dielectric constant, humidity, proximity, electrolyte level, and oxygen level. In addition, other known parameters may also be sensed.

The sensor 12 is disposed remotely from the reader 10. In one embodiment, the sensor 12 is implanted in a living human or animal body to make physiological measurements. Possible locations of interest include, but are not limited to, the following: blood vessels, skull, eye, bladder, stomach, lung, heart, muscle surface, bone surface, or any body cavity. The sensor 12 may be implanted for a short acute phase or a long chronic phase. The sensor 12 may be free standing or may be incorporated with other devices such as catheters, stents, shunts, filters, pacemakers, pacemaker leads, and vascular occlusion devices.

The sensor 12 is designed to have an operating frequency range 220 (not shown in fig. 2) that maps to a range of sensing parameter values. When it is desired to take a reading, reader 10 may send excitation pulse 14 in the vicinity of sensor 12, as in block 204 of FIG. 2. The pulses 14 may be short bursts of energy of a predetermined fixed frequency. The frequency of the pulses 14 may be selected to be at or near the middle of the operating frequency range 220 of the sensor 12, and the bandwidth of the pulses 14 may be narrow. The advantage of narrow bandwidth pulses is that it is less likely to electromagnetically interfere with other surrounding devices. Another advantage of narrow bandwidth pulses is that: by enabling a system designer to select a pulse frequency within a strict bandwidth dictated by governmental or industry regulations regarding the allocation of the electromagnetic spectrum, the system is allowed to more easily comply with these regulations. In one embodiment, the pulses 14 are narrow and centered at 13.56MHz, which is one of the so-called industrial, scientific, and medical (ISM) bands allocated by the International Telecommunications Union (ITU) for use by commercial RF devices. Yet another advantage of narrow bandwidth pulses is that: less power may be required than an equivalent continuous transmission scheme, thereby making reader 10 more suitable for battery operation and allowing the use of smaller components that typically require less heat dissipation than higher power counterparts. Finally, the advantage of sending a fixed frequency pulse 14 in step 204 of fig. 2 is that: the transmission circuit of the reader 10 is simple compared to a frequency sweep scheme or a continuous transmission scheme.

Since the sensor 12 is adjacent to the reader 10, step 206 of FIG. 2 is performed. The sensor 12 is powered with pulses 14 via inductive coupling between the antenna of the sensor 12 and the antenna of the reader 10. The pulse 14 causes a current to flow into the antenna of the sensor 12, thereby powering the "LC tank" circuit formed by the capacitor 15 and the inductor 13. The duration of the pulse 14 is typically short and in step 208, the reader 10 abruptly terminates the pulse 14. The energy stored in the LC tank circuit of the sensor 12 immediately begins to dissipate, causing the sensor 12 to oscillate at its resonant frequency. The antenna of the sensor 12 thereby emits a ringing signal 16 of this frequency. After terminating transmission, the reader 10 must enter a receive mode immediately, as in step 210, to detect the ring signal 16 and amplify the ring signal 16.

Depending on the measurement conditions, the ringing signal may be weak, noisy or short in duration, resulting in a penalty to the accuracy and resolution during frequency measurement. For this reason, in step 212, reader 10 may lock onto and hold the sampled ring signal at a constant frequency and strong amplitude for a sufficient time to obtain a high accuracy frequency measurement in step 214.

Fig. 3 qualitatively illustrates the ideal behavior of the reader 10 and sensor 12 in the frequency domain in an embodiment. The sensor 12 senses its physical parameter of interest within a predetermined range of values. The sensor 12 maps the physical parameter range onto a corresponding operating frequency range 220. Curve 224 is the transfer function of sensor 12 when the resonant frequency of sensor 12 is at the minimum of its operating frequency range 220. The peak of the sensor transfer function 224 is at the resonant frequency of the sensor 12. As the sensed parameter varies over the range of values, the sensor transfer function moves accordingly over the operating frequency range 220. Thus, depending on the value of the sensed physical parameter, the sensor transfer function may be centered at any location within the operating frequency range 220. Its resonance frequency (peak of the transfer function curve) corresponds to the value of the sensed parameter. When the sensed parameter is at the other end of its range, the sensor transfer function becomes the maximum frequency sensor transfer function 222.

The narrow band function 14 in fig. 3 represents the excitation pulse 14 shown in fig. 1. The narrow band function 14 is denoted f_xmtIs typically fixed at or near the center of the operating frequency range 220. The pulses 14 are typically narrow bandwidth, short in duration and fixed at a predetermined frequency f_xmt. These pulse characteristics give the reader 10 several advantages over other readers that must scan or change the frequency at which they transmit: simpler circuitry, simpler control software/firmware, reduced overall power consumption (enabling battery operation), lower power (and therefore smaller) components, less internal heat dissipation, reduced susceptibility to electromagnetic interference from external sources, reduced likelihood of electromagnetic interference with external devices, and increased ease of complying with government frequency allocation regulations.

Another important feature shown in fig. 3 is a horizontal line representing the minimum signal detection threshold 226 of the reader 10. After the excitation pulse 14 is turned off, the sensor 12 dissipates the energy it receives from the excitation pulse 14. In the absence of the mandatory excitation pulse 14, this energy causes an oscillation of the resonant frequency of the sensor 12, thereby emitting a ringing signal 16 (not shown in fig. 3). The signal strength (amplitude) of the ringing signal 16 is determined by the intersection of the excitation pulse 14 and the sensor transfer function: the amplitude of the ringing signal is limited by the product of these two functions at that point. In order for reader 10 to be able to detect and measure ring signal 16, the amplitude of the product at this intersection must be greater than or equal to signal detection threshold 226 of reader 10.

Fig. 4A, 4B, and 4C provide illustrative examples of typical signal exchanges between reader 10 and sensor 12 in the frequency domain. The processing shown in this figure is the same as that shown in flow chart form in fig. 2. In the initial condition shown in fig. 4A, the sensing parameter values are such that the transfer function 228 of the sensor 12 is centered at a frequency within the operating frequency range 220. Note that the sensed parameter (and thus the transfer function 228) changes on a much slower time scale than the electronic signals transferred between the sensor 12 and the reader 10, and thus the transfer function 228 is quasi-static with respect to these signals. Because the sensed parameter is quasi-static with respect to the electronic signal, the reader 10 is able to take multiple samples over a short time interval and average the samples to obtain a more accurate measurement.

In fig. 4B, reader 10 generates an excitation pulse 14. The pulse 14 is centered at a frequency f that is at or near the center of the operating frequency range 220_xmtA narrow bandwidth signal. Energy is transferred from reader 10 to sensor 12 when reader 10 generates an excitation pulse 14 in the physical vicinity of sensor 12. In one embodiment, the energy transfer utilizes f within the RF band_xmtOccurs through inductive coupling. Note the intersection point 230 between the reader excitation pulse 14 and the sensor transfer function 228. The product of these two amplitudes at this point will determine the amplitude of the ringing signal 16.

Next, in FIG. 4C, reader 10 stops transmitting excitation pulse 14. When the excitation energy ceases, the sensor 12 transitions from a forced driving characteristic at the transmit frequency at which phase error occurs due to resonance at the non-transmit frequency to a passive resonance characteristic at a frequency dependent on the resonant frequency of the sensor and its surroundings, which is approximately the peak of the curve 228. Due to the resonant energy within the inductor of the sensor 12, a time-varying magnetic field is generated around the sensor 12 at the resonant frequency, which the reader 10 can detect as a signal emitted at the resonant frequency.

Note that if the sensor 12 is subjected to the transfer function 228 toward the right in FIG. 4C (at f)_resIn increasing direction) of the sensed parameter, curve 228 is at point f_xmtThe amplitude of (a) is reduced, resulting in a reduction in the crossover level 230. Followed byIs on f_resFurther increases and reaches f_maxThe crossing amplitude 230 is equal to the minimum detection threshold 226 of the reader 10. If transfer function 228 moves further to the right, f_resExceeds f_maxAnd the crossing amplitude 230 falls below the detection threshold 226 of the reader 10. Now, the reader 10 can no longer detect the ringing signal 16, i.e. f_resOutside the operating frequency range 220 of the system. Note that the sensor 12 must be designed such that its transfer function 228 has a bandwidth wide enough to maintain the crossover amplitude 230 above the detection threshold 226 of the reader 10 throughout the operating frequency range 220. However, providing a sensor 12 with a wide transfer function 228 generally reduces the peak amplitude of the transfer function 228, and therefore a balance must be struck between amplitude and bandwidth. In general, as is clear from fig. 4A, 4B and 4C, the ability of reader 10 to detect and measure ring signal 16 will also depend on the power level of the ring signal after excitation pulse 14 has ceased, on system Q, and on the duration of ring signal 16.

The transfer function 228, the shape of the signals 14 and 16, and the operating range 220 shown in fig. 4A, 4B, and 4C are illustrated as examples. In some embodiments, transfer function 228 may have different characteristics and be relative to f at the peak_resMay be asymmetric. In addition, the operating range 220 is relative to f, which is the frequency of the excitation pulse 14_xmtMay be asymmetric. The asymmetry of the operating range 220 may occur as a result of characteristics of the sensor 12 or may be intentionally designed to counteract the asymmetry of the transfer function 228, the excitation signal 14, or the ringing signal 16.

In an alternative embodiment, reader 10 may transmit pulses that are not near the center of the operating range 220 of sensor 12. In this case, reader 10 transmits pulses at a frequency harmonically related to the frequency within the operating range 220 of sensor 12. That is, the higher or lower harmonics due to the transmitted pulse are used as the excitation pulses 14 shown in fig. 4A, 4B, and 4C.

In yet another embodiment, reader 10 may transmit more than two excitation pulses at different frequencies, either simultaneously or at different times. These multiple excitation pulses may excite different portions of the operating frequency range 220. Alternatively, a frequency generated by adding or subtracting a combination of these multiple pulses or their harmonics may be used as the excitation frequency 14 in fig. 4A, 4B, and 4C. The excitation pulse may also assume a gaussian shape or other non-sinusoidal shape.

Referring again to fig. 1, the reader 10 may also contain circuitry for converting ring frequency readings from the sensor 12 to digital form and storing the readings in on-board memory. The memory of the reader 10 may store other relevant data in addition to the measurements from the sensor 12. Examples include timestamp data, calibration coefficients, firmware required to implement system functions, firmware upgrades, part numbers, serial numbers, usage logs, historical data, configuration data, diagnostic data, information about the host location and applications of the sensor, and user-defined data.

The reader 10 may also contain a human-machine interface such as a display screen, LED, or audible indication corresponding to some aspect of the frequency data. In addition, the reader 10 may process the frequency data it receives for such functions as averaging, filtering, curve fitting, threshold monitoring, time stamping, trend analysis, and comparison to other data.

As shown in fig. 5, the reader 10 may also communicate with a data interface 17. Data interface 17 is located external to reader 10 and is configured to receive electronic signals from reader 10 and transmit signals to reader 10. In addition, the data interface 17 may provide power to the reader 10, for example, to charge a battery located within the reader 10. Examples of data interface 17 include a host computer, a docking station, a telephone network, a cellular telephone network, a GPS network, an optical network, a Bluetooth network, a storage area network, an Internet website, a remote database, a data entry device, audible sound, and a display.

The reader 10 and the data interface 17 may be directly connected to each other or indirectly connected via an intermediate device, or may communicate via a remote connection. The reader 10 and the data interface 17 may be provided in the same housing. The reader 10 and the data interface 17 may be connected via a cable or by a wireless link. The reader 10 may send information to the data interface 17. Examples include data related to the sensors 12, measurements taken from the sensors 12, time stamp data, part numbers, serial numbers, firmware modification information, usage logs, diagnostic data, historical data, status data, configuration data, information related to the host location and applications of the sensors, and user defined data. The data interface 17 may provide data and commands to the reader 10. For example, data interface 17 may provide reader 10 with information regarding the progress and interval for sampling sensor 12, calibration coefficients or look-up tables, firmware required to implement system functions, firmware upgrades, configuration settings, diagnostic commands, resets, reboots, user defined data, and user issued commands.

The data interface 17 may also communicate with a remote data system 18 to exchange status and control signals and provide sensor data. The remote data system 18 may include: a data collection module 19 for receiving data from the data interface 17; a data recording module 20 for storing the received data; and a data display 21 for displaying the sensor data. Like the data interface 17, the remote data system 18 can store and process data, issue commands, and distribute such data and commands, thereby allowing communication with multiple users on a data network. Like the connection between the reader 10 and the data interface 17, the connection between the data interface 17 and the remote data system 18 may be via a cable or may be wireless. The structure shown in fig. 5 in which the reader 10 is connected to the data interface 17 via a cable and the data interface 17 is wirelessly connected to the remote data system 18 is one example embodiment. Although the example of fig. 5 associates data recording and display functions with the remote data system 18, it will be apparent to those skilled in the art that these functions may also be performed using the external data interface 17 or the reader 10.

The system of reader 10, sensor 12 and data interface 17 described above is particularly advantageous in one embodiment in the field of biomedical telemetry. In the present embodiment, the sensor 12 is implanted within a living person to sense a physiological parameter such as blood pressure sensed from within an artery. Since the sensor 12 can be made very small by conventional techniques, the sensor 12 is well suited for this application, and since the sensor 12 is a passive sensor, the sensor 12 does not require an on-board power source that will eventually be depleted. Portions of reader 10 may be physically small enough to be hand-held, battery-powered, thermally cooled, and electromagnetically compatible with other electronic devices in its vicinity. These attributes derive from the simple low power circuitry generating narrow band fixed frequency excitation pulses 14 as described above. Thus, the reader 10 can be comfortably worn on a person's clothing in the vicinity of the implanted sensor 12 so that readings can be frequently read and processed/stored. The user may place reader 12 on data interface 17 in the form of a docking station periodically, for example daily. Data interface 17 may contain circuitry for charging the battery of reader 12, updating the settings and software of reader 12, and downloading its data. The data interface 17 may also communicate this data to the user, and other interested parties such as the user's physician, via the internet or telephone link. Because of the low power excitation scheme used by reader 12, such a system can frequently take accurate blood pressure readings with minimal inconvenience to the patient and efficiently communicate these blood pressure readings to the caregiver. It is clear that the present embodiment is also applicable to sensing any other internal physiological parameter that may cause a change in the resonant frequency of the passive LC sensor.

In a variation of the present embodiment, the sensor 12 is incorporated with another implantable medical device that performs a different function. For example, the sensor 12 may be a blood pressure sensor incorporated with a vasoocclusive device such as the Angio Seal product developed by st.jude Medical, Inc. In another variation of this embodiment, reader 10 may be incorporated with other devices. For example, the reader 10 may be mounted to a cellular phone, glasses, a hand-held music player, a video game machine, clothing, or a watch.

The sensor 12 including the capacitor 15 and the inductor 13 may be configured to assemble these circuit elements in a single package. Alternatively, in some applications it may be advantageous to place the capacitor 15 remote from the inductor 13 and connect the two elements, the capacitor 15 and the inductor 13, with conductive leads. By way of example, in embodiments where the sensor 12 is implanted in the human body, the pressure sensitive capacitor 15 may be placed at the site where the pressure of interest is found, and the inductor 13, which acts as an antenna, may be placed closer to the skin surface, thereby minimizing the wireless coupling distance between the sensor 12 and the reader 10. These conductive leads for connection may take any of a number of well-known forms including wires, printed flexible circuits, printed rigid circuits, feedthroughs, or rigid pins.

In implantable embodiments, it may also be advantageous to design the sensor 12 to accommodate minimally invasive implantation methods such as catheter-based delivery. Additionally, it may be desirable for a portion of the implantable sensor to be radiopaque or reflective of ultrasound to aid in implantation and post-implantation diagnosis.

The sensor 12 may be fabricated using a number of well-known techniques. The capacitive sensor 15 may be manufactured using micro-electromechanical systems (MEMS) technology, lithography technology or standard machining technology. The inductor 13 may include: a winding coil; FR4, Teflon, Rogers or other printed circuit boards; low temperature co-fired ceramic (LTCC), green tape, or other ceramic printed circuit boards; or any other inductor technology known to those skilled in the art. The inductor 13 may or may not be cored and may also utilize magnetic materials incorporated within one of the printed circuit board or ceramic technologies described above. The inductor and capacitor may be packaged together as a multi-chip module (MCM).

In another embodiment, as shown in fig. 6, the system of fig. 1 may further include an intermediate antenna 240. The intermediate antenna 240 includes two antennas, a reader-side antenna 242 and a sensor-side antenna 244, which are connected together in series. The intermediate antenna 240 may improve signal coupling between the reader 10 and the sensor 12 and may be used in situations where there are multiple barriers 246 and 248 between the reader 10 and the sensor 12 that are not easily penetrated by conductive leads. By way of example, for a sensor 12 implanted within a blood vessel, barrier layer 2(248) represents the vessel wall, and barrier layer 1(246) represents the skin surface. With the intermediate antenna 240 in place, the signal coupling between the reader 10 and the sensor 12 is more efficient because it occurs through conduction via the leads rather than through radiation via the medium in which the system is placed. In addition, antennas 242 and 244 may each be sized to match corresponding antennas of sensor 12 and reader 10, thereby further improving coupling efficiency. Finally, the sensor-side antenna 244 may be precisely aligned across the sensor inductor 13, thereby reducing errors due to inconsistencies between the reader 10 and the sensor 12 that may occur in the absence of the intermediate antenna 240. The intermediate antenna 240 may be made of a flexible circuit, a wire-wound coil, or other widely used components. Note also that the concept can be extended to applications where there are more than two barriers by adding more intermediate antennas 240 to each pair of barriers.

In another embodiment, the sensor 12 of FIG. 1 may also include a second LC tank with a separate inductor and capacitor, referred to as a reference resonator. The reference resonator can be fabricated using the same materials, processes and components as the sensing resonator, including inductor 13 and capacitor 15, but with two key differences. First, the values of the components of the reference resonator are fixed and do not change with the sensed parameter. Second, the fixed resonant frequency of the components of the reference resonator is designed to be outside the operating frequency range 220 of the sensing resonator. The purpose of the reference resonator is to provide a background reading that can be used to correct the sensor readings taken by reader 12. Some factors that cause inaccuracies, such as reader distance, changes in the intervening medium, orientation of the sensor towards the reader, aging of components, mechanical stress, electrical stress, outgassing, temperature, cell growth, and blood clotting, may affect the reference resonator in the same way as the sensing resonator. By understanding the relationship between the deviation of the reference resonator from its fixed frequency and the deviation of the sensing resonator from its nominal frequency, the reader can provide a correction factor for the sensing frequency based on the reference readings. In this embodiment, an additional step is introduced between steps 202 and 204 of FIG. 2, wherein the reader 10 can send an excitation pulse at the nominal resonant frequency of the reference resonator, observe any deviation in the reference ring frequency, and calculate (or obtain from a look-up table) the appropriate correction factors for the reading to be obtained in step 210. Alternatively, the reference reading may be taken after the sensing reading. While each change experienced by the sensing resonator may not affect the reference resonator in exactly the same way, this approach to "self-calibration" may improve performance by eliminating or reducing inaccuracies common to both resonators. For example, these inaccuracies may be associated with changes in distance, orientation, physiological response, intervening tissue, and other long-term changes in the behavior of the sensor 12, often collectively referred to as "sensor drift". In addition, frequency selection and other design aspects of the reference resonator must be carefully considered to avoid coupling with the original sensing resonator and common interactions with the reader.

Reader 10 includes circuitry for transmitting excitation pulses 14, receiving a ring signal 16, and processing ring signal 16 (fig. 7). For example, reader 10 includes timing and control circuitry 22 for configuring and activating other circuitry within reader 10. Solid arrows to and from the timing and control circuit 22 represent control interfaces such as digital signals or low frequency signals. The timing and control circuit 22 also generates an RF signal (shown by the dashed arrow) that is sent to the transmit circuit 24. Transmit circuitry 24 receives the RF signal and transmits excitation pulses 14 to antenna 26 to excite sensor 12. The timing and control circuitry 22 may provide an RF signal to the transmit circuitry 24 only during the intervals in which the excitation pulses are being transmitted to prevent leakage or coupling to other nodes within the system.

The reader 10 also includes an antenna 26 connected to the transmit circuit 24 and the receive circuit 28. Transmit circuit 24 transmits excitation pulse 14 using antenna 26 and receive circuit 28 receives ring signal 16 using antenna 26. In an embodiment, instead of switching between transmitting and receiving, the antenna 26 is always connected to both the transmit circuit 24 and the receive circuit 28. The design of the common antenna 26 requires special consideration to prevent damage to the receive circuitry 28. In particular, care must be taken not to overload the sensitive amplifier stage of the receiving circuit 28. In addition, the reader 10 requires a fast transition between the extreme overdrive condition present where the transmit circuitry 24 is driving the antenna 26 and the low voltage condition present at the antenna 26 during the receive and amplify phases. For example, the voltage of the antenna 26 may exceed a peak-to-peak voltage of 200 volts during the transmission of the excitation pulse, and may be a few millivolts, decaying rapidly to microvolts during the reception immediately following the excitation pulse 14. Although reader 10 is described as having a common antenna 26, it should be understood that reader 10 may include more than one antenna to perform the functions of transmitting excitation pulse 14 and receiving ringing signal 16 separately.

The reader 10 also includes a Phase Locked Loop (PLL)30 for receiving and locking to the ring signal 16. The receive circuit 28 may amplify and condition the ring signal 16 before sending the ring signal 16 to the PLL 30. The PLL30 includes a voltage controlled oscillator ("VCO") 32 (not shown in fig. 7), where the voltage controlled oscillator 32 may be operative to lock onto a frequency within a range of sensor resonant frequencies in the absence of a signal, or may be selected to prefer frequencies above or below the range of sensor resonant frequencies in the absence of a signal, thereby enhancing the lock time when the sensor resonant frequency is received. In an embodiment, a PLL is selected that performs well when the no-signal PLL locks in a range of frequencies slightly above the sensor resonant frequency. VCO32 generates an ac signal proportional to the ring signal frequency, referred to as count signal 250. The PLL30 adjusts the down-converted count signal to match the frequency of the ringing signal 16 and sends a count signal 250 to the frequency counter 34. VCO32 is connected to frequency counter 34 which determines the frequency of count signal 250 and provides a digital signal representative of that frequency to external interface circuitry 36 for communication to data interface 17. By operating the VCO32 at a higher frequency than the frequency of the ringing signal 16, the time required to count and record the frequency of the count signal 250 of the VCO32 can be greatly reduced.

The components of the reader 10 are designed to operate efficiently and reduce power consumption. To this end, the reader 10 includes a power down function. The timing and control circuit 22 controls the power state of the components using a wake-up timer 38 connected to the components (fig. 8). In the reduced power mode, some components may be completely powered down while other components may operate in a sleep mode in which power remains supplied to maintain configuration but the circuit is quiescent to minimize power consumption.

The timing and control circuit 22 may place the various components of the reader 10 in a sleep mode or a power-off mode when not in use. In addition, the integral reader 10 can be made to be placed in a low power mode at the system level for a period of time specified by the external controller. The timing and control circuit 22 may include a configuration buffer 40 for receiving timing indications from the external interface circuit 36. These indications determine the delay period before entering the power down mode and other delay periods for the wake-up timer 38. Entry/exit into the reduced power mode may be triggered by one of the on-board signals exceeding a threshold in addition to a timing indication from outside the reader 10. The firmware of the reader 10 may contain an algorithm for determining entry/exit into/out of the power reduction mode.

During reading acquisition, the wake-up timer 38 may wake-up the components of the reader 10 at the appropriate time to ensure that the components are operational when needed. In particular, the wake-up timer 38 may communicate with the transmit timer 42, the receive timer 46, the PLL timer 48, and the frequency counter timer 50 to wake-up and control various components of the reader 10. Once started, each of these timers can control and power the various components. When configured, the wake-up timer 38 may be delayed by a specified interval, which may be 0 seconds, before sending the start signal 52 to start the other timers. As shown in FIG. 8, the enable signal 52 is not shown as a continuous line from the wake-up timer 38 to the various counters, thereby preventing line crossing and minimizing aliasing.

Once started, transmit timer 42 determines the appropriate sequence and period for power control signal 54, attenuation control signal 56, Q control signal 58, and RF enable signal 60 to properly sequence transmit circuit 24 and transmit frequency generator 44. The power control signal 54 controls the power state and sleep state of the transmit circuit 24. The attenuation control signal 56 controls the activation of attenuation circuits within the transmit circuit 24 to quickly dissipate the energy of the antenna 26 at the end of the transmit period. Q control signal 58 controls switching circuitry within transmit circuit 24 to reduce Q and modify the bandwidth of antenna 26 upon receipt of ring signal 16. The RF enable signal allows the transmit frequency generator 44 to transmit the RF signal to the transmit circuitry 24. In an embodiment, the transmit frequency generator 44 provides the RF signal to the transmit circuitry 24 only during periods when the transmit circuitry 24 is transmitting the excitation pulse 14.

The receive timer 46 is configured to assert the appropriate sequence and period of the power control signal 62 to properly sequence the receive circuitry 28.

The PLL timer 48 establishes the proper sequence and period for the power control signal 64 and the S/H mode signal 66 to properly sequence the PLL 30. The power control signal 64 controls the power state and the sleep state of the PLL 30. The S/H mode signal 66 controls the sample and hold circuit in the PLL30 that is used to lock the PLL to the transmit frequency and then to the frequency of the ringing signal 16, and then to hold the frequency of the count signal 250 of the VCO32 at the lock frequency until the counter 34 measures the frequency.

The frequency counter timer 50 establishes the proper sequence and count interval for the power control signal 68 and the start/stop count signal 70 to properly sequence the frequency counter 34. The power control signal 68 controls the power state and the sleep state of the frequency counter 34. The start/stop count signal 70 controls the time at which the frequency counter 34 starts and ends measuring the frequency of the count signal 250 of the VCO 32.

Note that although fig. 8 contains signals sharing the same name, such as "start up", "configuration", and "power control", each of these signals is unique to the circuit block to which it is connected. For example, as described above, the power control signal 68 from the frequency counter timer block 50 is not the same signal as the power control signal 64 from the PLL timer block 48.

The transmit circuitry 24 is configured to transmit the excitation pulse 14 to the sensor 12 (fig. 7) via the antenna 26. The excitation pulse 14 may be a fixed frequency burst at or near the resonant frequency of the sensor 12 or a burst of rapid change in frequency. For example, the excitation pulse 14 may be a fixed frequency burst within several bandwidths of the resonant frequency of the sensor 12. Alternatively, the excitation pulse 14 may be a very short duration fixed frequency or rapidly changing frequency burst or sweep at or near a frequency harmonically related to the resonant frequency of the sensor 12. The excitation pulse 14 may also be an ultra-wideband pulse. These multiple excitation pulses 14 are possible because the ring signal 16 is received when the transmission of the excitation pulses 14 has stopped. Thus, the transmission of the excitation pulse 14 can be limited to the frequency band, amplitude and modulation scheme allowed by the regulatory body. Since the sensor 12 is a completely passive device, radio frequency regulations may not be applicable to the sensor 12.

The excitation pulse 14 does not require a significant amount of transmission time, since a short energy transmission results in a complete sample of the ringing signal 16. Power consumption can be reduced by using a lower transmission duty cycle, thereby reducing the duty cycle of the transmit, receive, count, and digital processing circuits. By reducing power consumption, battery powering is a more viable option to charge the reader 10.

The excitation pulse 14 may be configured to maximize several system parameters. For example, if a fixed frequency excitation pulse 14 is used, the frequency of the burst may be configured to maximize parameters such as maximum allowable transmit peak power, maximum freedom from in-band or near-band interference during the "receive" interval with the PLL locked to the ring signal 16, maximum wide acceptance of a particular frequency for reader transmission for the desired sensor purpose, or other such criteria.

Fig. 9 shows the transmission circuit 24. The level shifter 72 of the transmit circuit 24 receives the control signals 54, 56, 58 and the RF signal from the timing and control circuit 22. Level shifter 72 buffers these inputs and converts the control logic levels to circuit drive levels. Transmit driver 74 amplifies the RF signal to provide sufficient power to drive antenna 26. The Q control circuit 76 is activated during reception to reduce the Q of the combined antenna 26 and tuning and d.c. block 82. The attenuation circuit 78 is briefly activated immediately at the end of the transmission of the excitation pulse 14 to absorb the energy of the antenna and allow the antenna to respond to the ringing signal 16. The attenuation circuit 78 may provide different Q factors to the antenna to improve reception of the ringing signal 16. Power control circuit 80 controls the power on and sleep modes of the components within transmit circuit 24. The tuning and d.c. block 82 tunes the antenna 26 and prevents dc current from unduly biasing the attenuation circuit 78. The RF output or excitation pulse 14 from the transmit circuitry is transmitted to both the antenna 26 and the receive circuitry 28.

Once the transmit circuit 24 has transmitted the excitation pulse 14, the receive circuit 28 is configured to listen for the ring signal 16. Referring to fig. 10, the high Z buffer/clamp 84 includes a high impedance ("high Z") input device for limiting the effect of the receive circuitry 28 on the tuning and the tuning by the d.c. block 82. The high Z buffer/clamp 84 also serves to protect the amplifier stage 86 from extreme voltages present on the antenna 26 during transmission of the excitation pulse 14. The voltage of the antenna 26 may reach a peak-to-peak voltage of 200 volts or more during the transmission of the excitation pulse, requiring only about 60 picogram farads of capacitance to tune the antenna 26. In one embodiment, a 1 picofarad capacitor is used as a high impedance input current limiting circuit on a 13.56MHz transmit circuit. Low capacitance diode junctions that shunt excess voltage to the supply and undervoltage to ground may be configured on the receiver side of a 1pF capacitor so that the capacitor limits the current flowing through the diode while these diode junctions protect the receiver amplifier from the high voltage flowing through the antenna 26 during transmission.

The amplifier stage 86 amplifies the ring signal 16 to a sufficient level to drive the input of the PLL 30. The amplifier stage 86 needs to be carefully designed to achieve a proper transient response when the transmitted signal of the excitation pulse 14 is removed and attenuated, and to receive a low level ring signal 16. The output of the high Z buffer/clamp 84 may be adjusted using a Q-tuned common gate amplifier stage that reflects low drain loading, followed by several filters interspersed between the high gain amplifier stages. These filters may be resistor-capacitor ("RC") type filters or inductor-capacitor ("LC") type filters. In an embodiment, these filters may all be RC band pass filters. Another common gate amplifier stage with Q-tuned, low reactive drain loading may be used for final bandpass adjustment before feeding the signal to the input of the PLL 30. The design enables all of these amplifier types to go from very low signal input levels to very high signal input levels without signal distortion such as frequency doubling or halving due to stage saturation characteristics, and very high input impedance can be achieved with the superior transient response characteristics of the common gate amplifier stage and the RC filter interspersed between the high gain amplifier stages. Power supply and signal separation between stages must be specifically considered to prevent unnecessary oscillation due to the extreme gain associated with amplifier stage 86.

The power control circuit 88 may apply and remove power to and from the buffers in the amplifier stage 86 and the high Z buffer/clamp 84 to reduce power consumption. It should be noted that the high Z buffer/clamp 84 is designed to provide full protection even if power is removed, since the excess energy only powers the amplifier stage 86 before dissipation. The input impedance is high enough to limit the excess energy to prevent excessive power being supplied to the amplifier stage 86. In an embodiment, the receive circuit 28 operates during the transmission of the excitation pulses 14 to shorten the time required for the PLL30 to lock onto the ring signal 16.

PLL30 receives the amplified and conditioned ring signal 16 from receive circuit 28. Referring to fig. 10 and 11, the RF signal from the amplifier stage 86 of the receive circuit 28 is fed to the RF buffer 90 of the PLL 30. The RF buffer 90 may feed the RF signal to an optional RF divider 92 (fig. 11) that divides the RF signal frequency by an integer value. The RF divider 92 then feeds the RF signal to a first input of a phase frequency detector 94. The output of the phase frequency detector 94 is fed to a sample and hold (S/H) error amplifier 96. The S/H error amplifier 96 controls the frequency of the VCO 32. The count signal 250 output by the VCO32 is fed to the VCO divider 98, where the output of the VCO divider 98 is in turn fed to a second input of the phase frequency detector 94. The PLL30 may comprise an output buffer 102, wherein the output buffer 102 is used to relieve the load of the VCO32 in case the frequency of the count signal 250 is forwarded to the frequency counter 34. VCO divider 98 allows VCO32 to operate at a much higher frequency than ring frequency 16. As a result, the time required to count and record the VCO signal frequency can be greatly reduced. Furthermore, a shorter count interval reduces VCO drift during counting and allows for a higher sampling rate.

The phase frequency detector 94 is configured to determine the frequency and phase error between the divided RF signal and the divided VCO signal. This is preferably accomplished by filtering and amplifying the signal fed to the S/H error amplifier 96. In addition, the S/H function can optimally forward the filtered and amplified signal to control the VCO 32. In this manner, a closed control loop is formed that multiplies the frequency of the count signal 250 of the VCO32 by the frequency of the ring signal 16 and the integer of the VCO divider 98 divided by the integer of the RF divider 92. The PLL30 may include additional frequency dividers to optimize the circuit design and increase the frequency range of the potential VCO 32.

The PLL timer 48 sends the S/H mode control signal 66 to the S/H error amplifier 96 of the PLL 30. The S/H mode control signal 66 may place the VCO32 in a sampling mode. In an embodiment, the VCO32 is placed in the sampling mode for a predetermined length of time. In the sampling mode, the frequency of the divided VCO count signal is adjusted to match the frequency of the ring signal 16, as described above. When the S/H mode control signal 66 is placed in the hold mode, the S/H error amplifier 96 will hold its output constant, thereby making the control voltage to the VCO32 substantially constant for a length of time sufficient to determine the frequency of the count signal 250 of the VCO 32.

The power control signal 64 from the PLL timer 48 to the power control circuit 104 determines whether the PLL30 is in a power on mode or a sleep/power off mode to conserve power. Depending on the particular PLL30 used, a control and communication link (not shown) may be required to set the integer number of RF dividers 92, the integer number of VCO dividers 98, and the output and output configuration of phase frequency detector 94. The communication link may be specific to the particular PLL30 used.

As shown in fig. 12, the frequency counter 34 includes a counter stage 106, a counter buffer 108, and a power supply control circuit 110. The frequency counter timer 50 sends the start/stop control input 70 to the counter stage 106 and the counter buffer 108. The frequency counter timer 50 also sends the power control input 68 to the power control circuit 110. The counter stage 106 counts the VCO signal frequency from the output buffer 102 of the PLL 30. The counter stage 106 starts counting when the start/stop control command starts and ends counting when the start/stop control command stops. When the start/stop control command stops, the counter buffer 108 is loaded with the count value from the counter stage 106. Power control circuit 110 controls the power-on mode and sleep mode of the components in frequency counter 34. The output of counter buffer 108 may supply a count input to external interface circuit 36. The ringing frequency 16 may be determined from the frequency count and subsequently the sensed parameter.

In other embodiments, other methods for measuring the received and amplified frequencies may be employed. These other methods may include direct counting of the ringing signal, or various frequency-to-voltage conversion circuits known in the art.

In operation, reader 10 is ordered as follows. During periods when sensor 12 is not being sampled, all components of reader 10 are placed in a reduced power mode. The wake-up timer 38 in the timing and control circuit 22 is configured for a particular sampling delay or sampling interval. At a specified time, the wake-up timer 38 initiates a sampling sequence. Specifically, the wake-up timer 38 powers up or wakes up the components of the reader at the appropriate times to ensure that the components are operational when needed.

External interface circuitry 36 is typically not required in the sampling sequence, except for receiving the final data generated. The entry/exit of the external interface circuit 36 into/out of the low power mode may be handled by an internal controller or an external controller other than the timing and control circuit 22. The timing and control circuit 22 provides the RF signal to the transmit circuit 24 for a short period of time, such as about 20 milliseconds. The RF signal from the timing and control circuit 22 is then terminated and the transmit circuit 24 is controlled to rapidly attenuate the transmit signal at the antenna 26. The transmit circuit 24 is then placed in the appropriate mode to allow the ring signal 16 to be received at the antenna 26. In an embodiment, when the antenna 26 is configured to receive a ring signal 16, the attenuation of the antenna 26 is greater than the attenuation of the ring signal 16.

During the transmission of the excitation pulse 14, the receive circuit 26 receives, conditions and clamps the transmitted RF signal at the antenna 26. Once transmission of excitation pulse 14 ceases and antenna 26 is configured to receive ring signal 16, receive circuitry 28 transitions to a high gain receive mode to receive ring signal 16 from antenna 26. PLL30 is in sampling mode to allow RF buffer 90 to receive the regulated output of receive circuitry 28. When the antenna 26 begins to receive the ringing signal 16, the PLL30 transitions from locking to the frequency of the transmitted excitation pulse 14 to locking to the frequency of the ringing signal 16. After a time interval sufficient for the PLL30 to lock to the frequency of the ringing signal 16, the PLL30 is transitioned to hold mode to maintain the frequency of the count signal 250 of the VCO32 at the frequency of the ringing signal 16. The time required for locking may be predetermined or may be adapted based on the PLL lock condition. After locking, the receive circuitry 28 and transmit circuitry 24 are powered down or placed in a sleep mode as appropriate.

Once the PLL30 is in hold mode, the timing and control circuit 22 instructs the frequency counter 34 to count a controlled interval of the frequency of the count signal 250 of the VCO 32. When this counting is complete, the components of PLL30 are powered down or placed in sleep mode as appropriate and the count value is passed to external interface circuitry 36. The components of frequency counter 34 are then powered down or placed in sleep mode as appropriate, followed by the components of timing and control circuit 22 being powered down or placed in sleep mode as appropriate. If programmed to sample at intervals, the wake-up timer 38 of the timing and control circuit 22 counts until the next sample should be taken. Otherwise, the timing and control circuit 22 waits for a wake-up command from the external interface circuit 36 and any other required indication. In burst sampling mode, the power up time required for components to be ready may precede the power down time, in which case these components will remain powered up until the sampling burst is complete.

The embodiment of the PLL circuit 30 in the reader 10 shown in fig. 13 includes several features that may be added to the PLL30 to achieve functionality that is alternative to, but equivalent to, the functionality of the PLL30 circuit described above. Some or all of the variations seen between fig. 11 and 13 may be applied to enhance the operation of the PLL30 of fig. 11. Selectable input RF buffer 111 allows the RF signal from amplifier stage 86 or a reference signal generated at other components of reader 10 to be selected for input to RF divider 92. The selection is determined by the reference/receive control input of the RF buffer 111. Error amplifier 112 has been simplified and does not directly provide the sample and hold capability previously described for S/H error amplifier 96 from fig. 11.

Fig. 13 shows circuit elements including an analog-to-digital (a/D) converter 113, a digital-to-analog (D/a) converter 114, and a switch 115. These elements may be used to implement sample and hold functions. In the configuration of fig. 13, during the time that the reader 10 sends the excitation pulse 14 to the sensor 12, the reference frequency signal "Ref signal" may be selected as an input to the RF buffer 111, and the reference signal is maintained until the time that the RF signal at In a of the selectable input RF buffer 111 becomes stable and available from the receiving circuit 28. This reference signal allows the PLL30 to "pre-lock" to a stable reference signal, thus shortening the lock time when a ringing signal is available from the receiving circuit 28. The RF divider 92 divides the output of the selectable input RF buffer by any value equal to or greater than 1 and then feeds the divided buffer signal to the phase frequency detector 94. The output of the phase frequency detector 94 is fed to an error amplifier 112, where the error amplifier 112 provides the appropriate gain and frequency response needed to serve as a control signal for the VCO32 in the PLL 30. The output of the error amplifier 112 is fed to input a of switch 115. When input a is selected, switch 115 passes the signal of error amplifier 112 to both VCO32 and a/D converter 113. The a/D converter 113 is then used to sample the control voltage for the VCO to determine the control voltage level that locks the VCO32 to the frequency associated with input a of the selectable input RF buffer 111. As will be explained later, the signal of the a/D converter 113 may be used to indirectly measure the frequency of the VCO32 and may be used to determine the appropriate setting of the D/a converter 114 so that the switch 115 may be set to input B to maintain the VCO32 at the locked frequency input level for any period of time, thereby implementing the same digital sample and hold function as described for the S/H error amplifier 96 of fig. 11.

Several slight modifications to the operation of the circuit of fig. 13 as described may allow for functionally equivalent effects. One such modification is to use input B of selectable input RF buffer 111, fed at a known frequency, to calibrate the voltage of a/D converter 113 to the RF signal frequency of the particular receive circuit 28. Once calibrated so that the relationship between the signal input to the RF buffer and the digital output of the a/D converter 113 is well defined, the output of the a/D converter 113 can be used to represent the frequency of the ringing signal 16. The output of the a/D converter 113 becomes the PLL output. Operation in this manner will allow the a/D converter 113 to partially or completely replace the functions of the output buffer 102 and the frequency counter 34.

Another modification to the operation of the described circuit of fig. 13 is to perform a lock analysis on the PLL30 using data from the a/D converter 113 to reduce the lock time and improve the lock frequency accuracy. This operation is possible because the output of error amplifier 112 converges on the lock-in voltage value when signal 16 of sensor 12 is available at the output of receive circuit 28, and then diverges in a predictable manner when the level of signal 16 of sensor 12 decays past a position where lock-in may be maintained.

Another modification to the operation of the circuit of fig. 13 described is to use the D/a converter 114 to generate specific voltages at the input of the VCO32, record the output of the a/D converter at these specific voltages, and use the frequency counter 34 to determine the frequency of the signal at the output of the output buffer 102. This allows the use of frequency counter 34 to calibrate the a/D converter for one or more frequencies.

Minor modifications to the circuit of fig. 13 that will be apparent to those of ordinary skill in the electronic design art include reconfiguring the positions of the switch 115 and D/a converter 114 from those shown in fig. 13 to be between the phase frequency detector 94 and the error amplifier 112. This configuration requires the following additional steps to be performed using the a/D converter 113 or the frequency counter 34, or both: the output of the D/a converter 114 via the error amplifier 112 is calibrated to determine the appropriate scaling to achieve the desired control voltage of the VCO 32. However, instead of using the reference signal at input B of the selectable input RF buffer 111, this configuration allows the D/a converter 114 to be used for pre-locking. Combining this configuration with the calibration scheme of a/D converter 113 previously described as allowing the elimination of output buffer 102 and frequency counter 34, this may moderately reduce the power required to operate reader 10 by shortening the time required to resolve the resonant frequency of sensor 12 for each ring cycle. Another slight modification of the described embodiments is to allocate system processing load to appropriate locations based on power limits, computational complexity, time critical requirements, or other system related priorities. Such modifications may cause the designer to place the processing or analysis of the data from the a/D converter 13, D/a converter 114, or frequency counter 34 in any of the remote data system 18, reader 10, or external data interface 17.

In yet another embodiment of the circuitry of the reader 10, a digital spectrum analysis circuit replaces the PLL30 and frequency counter 34 of fig. 7, resulting in a modified block diagram shown in fig. 14. Here, the digital sampling circuit 260 replaces the PLL30, and the spectrum analyzing circuit 262 replaces the frequency counter 34. Likewise, the digital count signal 264 replaces the analog count signal 250.

Functionally, digital sampling circuit 260 extracts and digitizes information from ring signal 16 for the short ring duration of ring signal 16. Receive circuitry 28 may amplify and condition ring signal 16 before sending ring signal 16 to digital sampling circuitry 260. Digital sampling circuitry 260 may directly sample the radio frequency output of receive circuitry 28 to obtain time domain based data for further analysis.

In an embodiment, reader 10 further includes a spectral analysis circuit 262, wherein spectral analysis circuit 262 is to convert time domain data output from digital sampling circuit 260 to frequency domain data and to buffer the frequency domain data for forwarding to external interface circuit 36. Spectral analysis circuitry 262 may also include discrimination functionality to determine the ring frequency of ring signal 16. All or a portion of the functionality of the spectrum analysis circuit 262 may be readily performed by the reader 10 or by the remote data system 18, with the primary differences in implementation being in the type and amount of data sent via the external interface circuit 36, and the required processing power of the location where processing is performed, as will be apparent to those skilled in the art.

The timing and control circuit 22 controls the digital sampling circuit 260 and the spectral analysis circuit 262 in the same manner as the PLL embodiment described in fig. 8. FIG. 15 is a block diagram illustrating an alternative embodiment of the timing and control circuit 22 configured to control the alternative reader 10 circuit shown in FIG. 14. The PLL timer 48 of fig. 8 is replaced with the digital sample timer 274 of fig. 15. The timer determines the appropriate sequence and period for the power control signal 270 and the sample start signal 272 to order the digital sampling circuit 260. Power control signal 270 controls the power state and sleep state of digital sampling circuit 260. The sample start signal 272 causes the digital sampling circuit 260 to collect an appropriate number of samples in the burst sampling mode for transmission to the spectrum analysis circuit 262.

Likewise, the frequency counter timer 50 of fig. 8 is replaced with the spectrum analysis timer 280 of fig. 15. The spectrum analysis timer 280 establishes the appropriate sequence and timing for the power control signal 276 and the analysis start signal 278 to sequence the spectrum analysis circuit 262. The power control signal 276 controls the power state and sleep state of the spectral analysis circuit 262. The analysis start signal 278 controls the time at which the spectrum analysis circuit 262 begins evaluating the sample burst 264 provided by the digital sampling circuit 260.

The receive circuit 28 in the alternative embodiment of fig. 14 is functionally and architecturally identical to the receive circuit 28 of the PLL-based embodiment of fig. 7 and 10, with the only difference being that the output signal from the amplifier stage 86 is fed to an analog-to-digital converter 290 at the input of the digital sampling circuit 260, instead of the PLL 30.

Fig. 16 is a block diagram illustrating an embodiment of a digital sampling circuit 260. The RF signal from amplifier stage 86 of receive circuit 28 is fed to the input of analog-to-digital converter (ADC)290 of digital sampling circuit 260. The ADC 290 converts the RF signal into a set of time-dependent samples acquired at sufficiently close intervals with a sufficient number of samples to allow the spectrum analysis circuit 262 to achieve its required frequency accuracy. The set of time-dependent samples is referred to herein as a digital sample burst 264.

As shown in fig. 17, the digital sample burst 264 output from the ADC 290 is fed to the time-frequency domain conversion circuit 94 of the spectrum analysis circuit 262. The internal operation of the time-frequency domain transform circuit 94 is not specified here because the transform may be any of several ways including a fast fourier transform or a discrete fourier transform, a discrete wavelet transform or a continuous wavelet transform, any of several laplacian transforms, any of several Z transforms, or other transform algorithms known in the art. The internal workings of the time-frequency domain conversion circuitry 94 may be implemented in hardware or software or any combination of the two to achieve the desired conversion. Since the output of the time-frequency domain conversion circuit 94 is generated at sampling intervals and may contain a plurality of values to be passed to the external data interface 17, a results buffer 96 is shown in the spectrum analysis circuit 262 to hold the values until they can be passed to the external data interface 17.

In this digital spectrum analysis embodiment, the operating sequence of reader 10 is the same as described above for the "reader operating sequence" except that digital sampling circuit 260 and spectrum analysis circuit 262 perform functions related to the determination of the frequency of ringing signal 16. When the antenna 26 begins receiving the ring signal 16, the digital sampling circuit 260 samples quickly for a predetermined or calculated period of time to obtain a burst of digital samples 264. After digital sampling burst 264 is complete, receive circuitry 28 and digital sampling circuitry 260 are powered down or placed in a sleep mode, as appropriate. Spectral analysis circuitry 262 converts the data of digital sample burst 264 to the frequency domain and places the result in result buffer 96, and then transitions to a low power mode. The components of the timing and control circuit 22 are then powered down or placed in a sleep mode as appropriate. If programmed to sample at intervals, the wake-up timer 38 of the timing and control circuit 22 counts until the next sample should be taken. Otherwise, the timing and control circuit 22 waits for a wake-up command from the external interface circuit 36 and any other required indication. The sampled data in the results buffer 96 remains available to the external interface circuitry 36 for delivery to the remote data system 18, as controlled by the communication interface.

It will be apparent to those skilled in the art that numerous minor modifications may be made to the digital spectral analysis embodiments described to achieve functionally equivalent results. One such modification is the use of zero padding of the ADC data, which is common practice for evaluating time-frequency domain conversions of signal burst data. Another such modification is to move the physical location of spectrum analysis circuit 262 from reader 10 to remote data system 18, where the data of ADC 290 is transmitted in time domain form from reader 10 to remote data system 18. Yet another such modification is to frequency convert the ring signal 16 at some point within the reader 10 by means of a frequency doubling, dividing, summing or differencing circuit, changing the ring signal 16 to an intermediate frequency signal for any of a number of reasons related to frequency selectivity, bandwidth, sampling time, etc. Yet another such modification is the use of digital signal processing techniques to filter, shape, analyze, compare with other data, or other processing and evaluation of frequency or time domain data.

Also, those skilled in the art will readily observe that combinations of the various frequency determination methods disclosed herein may be made and that such combinations are advantageous in different applications. For example, an analog sample and hold circuit may be used in conjunction with digital spectral analysis to hold the ring signal 16 long enough to obtain enough samples for digitization.

In another embodiment, a standard RFID tag of the type known to those skilled in the art may be incorporated with the sensor 12. Such a tag may have a separate antenna and may operate at frequencies outside of the sensor operating range 220. Such a tag may be encoded with configuration information for the sensor 12.

Embodiments of the invention have been described above, and it is apparent that modifications and alterations will occur to others upon reading and understanding the preceding detailed description. It is intended that the following claims include all such modifications and changes as fall within the scope of the claims and their equivalents.

Claims (7)

1. A system for obtaining measurements from a remote location, the system comprising:

a wireless sensor for changing a resonant frequency of the wireless sensor in proportion to at least one sensed parameter; and

a reader for transmitting a plurality of excitation pulses of a fixed frequency to the wireless sensor, the fixed frequency being a predetermined fixed frequency, receiving at least one signal from the wireless sensor in response to the excitation pulses, sampling and holding the received signals, and averaging at least two of the signals received from the same wireless sensor,

wherein the transfer function of the wireless sensor has a bandwidth wide enough such that the amplitude of the product of the excitation pulse and the transfer function of the wireless sensor at the intersection of the two is greater than or equal to the detection threshold of the reader over the entire operating frequency range.

2. The system of claim 1, wherein the wireless sensor comprises at least one capacitor and at least one inductor, and wherein a value of at least one of the at least one capacitor and the at least one inductor changes in response to the at least one sensed parameter.

3. The system of claim 1, wherein the reader comprises a phase-locked loop circuit that generates a count signal and adjusts a frequency of the count signal to match a frequency of the received signal, and wherein the phase-locked loop circuit is placed in a hold mode to hold the count signal to determine the frequency of the count signal.

4. The system of claim 3, wherein the count signal is generated by a voltage controlled oscillator, and wherein the system samples a control voltage of the voltage controlled oscillator after adjusting the frequency of the count signal to determine the frequency of the received signal.

5. The system of claim 1,

the wireless sensor further comprising an additional resonant circuit having a fixed resonant frequency outside a bandwidth corresponding to a frequency change caused by the sensed parameter,

the reader also transmits a further excitation pulse at the fixed resonance frequency of the additional resonant circuit,

the reader also receives a response from the additional resonant circuit, an

The reader also determines a frequency of a response from the additional resonant circuit to calibrate the at least one signal received.

6. The system of claim 1, wherein the plurality of excitation pulses are at least one of:

a burst containing a frequency within a bandwidth of ± 20% of the frequency of the received signal;

bursts containing frequencies within a bandwidth of ± 20% of a subharmonic frequency of the received signal;

1/3 ultra-wideband pulses having a pulse width less than twice the period of the signal and spectral content no less than the frequency of the received signal;

a burst of frequency not less than 8/10 and not more than 12/10 of the frequency of the received signal comprising not less than 10 cycles and not more than 10000 cycles; and

a burst containing a frequency harmonically related to a resonant frequency of the wireless sensor.

7. The system of claim 1, wherein the wireless sensor is a passive device and the response signal is a ring signal.