EP3739601B1

EP3739601B1 - Interleaved llc converter

Info

Publication number: EP3739601B1
Application number: EP19175151.0A
Authority: EP
Inventors: Robert Weger
Original assignee: Infineon Technologies Austria AG
Current assignee: Infineon Technologies Austria AG
Priority date: 2019-05-17
Filing date: 2019-05-17
Publication date: 2022-12-28
Anticipated expiration: 2039-05-17
Also published as: EP3739601A1

Description

TECHNICAL FIELD

The present disclosure relates to the field of switching converters, in particular to an Interleaved LLC converter (ILLC converter).

BACKGROUND

LLC resonant converters are commonly used in modern power supply designs due to their ability to operate with high efficiency at moderate complexity with regard to control circuitry. Furthermore, multiple LLC resonant converters may be combined in parallel to spread power losses and to decrease the size of output filters and thus the ripple of the output voltage. Such combinations of two or more LLC converter units are referred to as Interleaved LLC converter or ILLC converter.
Additional control circuitry is usually employed in ILLC converters, for example to balance the load (i.e. the output current) among the individual LLC converter units. A misbalance/asymmetry of the output currents provided by the individual LLC converter units is, inter alia, caused by tolerances of circuit components used in the tank circuits (LC circuits) of the individual LLC converter units. Standard tolerances of, e.g. 5%, may lead to significantly differing resonance frequencies of the mentioned tank circuits and thus to the LLC converter units operating at different operating points.
One approach to balance the load among two or more LLC converter units includes using current-controlled variable inductors (CCVIs) in the tank circuits of the individual LLC converter units which allows to tune the resonance frequency of the tank circuits by adjusting the inductance of the CCVI. Usually a CCIV is composed of a coil, which is would around a magnetic core with an air gap. The coil has an inductance, which can be varied by generating a magnetic bias field in the magnetic core using a further coil, which is also would around the magnetic core. The magnitude of the magnetic bias field depends on a control current passing through the further coil.
CCIVs may be implemented in various different ways, and one problem to be solved by the embodiments described herein can be regarded as how to improve an interleaved LLC converter by improving the CCIV with regards to, for example, efficiency (low losses), electromagnetic compatibility (low disturbing stray fields) and size.
EP 2 624 260 A1 discloses a magnetic component with first, second and third U/UR-cores. The first U/UR-core and the second U/UR-core are magnetically coupled through gaps, wherein the legs of the U/UR-cores abut each other and form an O-core. The third U/UR-core is magnetically coupled through gaps to the first and the second U/UR-core. The legs of the third U/UR-core abut the legs of the first and the second U/UR-cores. A first primary winding and a first secondary winding are arranged on a body section of the first U/UR-core. A second primary winding and a second secondary winding are arranged on a body section of the second U/UR-core. Alternatively, the O-core formed by the first and second U/UR-cores may be rotated by 90°, wherein the windings on the O-core are not rotated.
EP 2 299 456 A1 discloses magnetic components with first, second, third and fourth U/UR-cores. Free ends of the first and second U/UR-cores face each other and form an O-core which is arranged between the third and fourth U/UR-cores with the free ends of the third and fourth U/UR-cores facing toward the O-core. Between each of the free ends of the third and fourth U/UR-cores and the O-core an air gap is provided. The magnetic components further include six windings: a first primary winding and a first secondary winding wound on the first U/UR-core, a second primary winding and a second secondary winding wound on the second U/UR-core, a third secondary winding wound on the third U/UR-core and a fourth secondary winding wound on the fourth U/UR-core. Alternatively, the air gaps between each of the free ends of the third and fourth U/UR-cores and the O-core may be omitted.
EP 3 349 224 A1 discloses a magnetic component with four linearly stacked U-shaped core elements: A first transformer core element, a second transformer core element, a first filter core element and a second filter core element. Each of the four core elements comprises a first and a second outer leg, and a flange. The two transformer core elements are facing each other with their first outer legs and their second outer legs and form an O-like shaped transformer core section having an air gap between their first outer legs. The O-like shaped transformer core section is arranged between the first filter core element and the second filter core element with the first and second outer legs of the first filter core element and the second filter core element facing the O-like shaped transformer core section. Between the O-like shaped transformer core section and each first outer leg of the first filter core element and the second filter core element an air gap is formed. The O-like shaped transformer core section has a window like opening which provides a transformer winding window for receiving turns of a first and a second lower current transformer winding part and of a first and a second higher current transformer winding part. The higher current transformer winding parts and the lower current transformer winding parts are arranged on the first outer legs of the transformer core elements in a way that the lower current transformer winding parts are arranged between the higher current transformer winding parts. A first filter winding is arranged on the first outer leg of the first filter core element and a second filter winding is arranged on the first leg of the second filter core element.
EP 3 133 614 A1 discloses a LLC resonant converter circuit having one resonant capacitor, one series resonant inductor, one parallel resonant inductor and a transformer comprising a primary winding, a first secondary winding and a second secondary winding.
RU 2 047 262 C1 describes a converter having a ring-shaped core with a center bar arranged between a first outer bar and a second outer bar. A first winding of a primary coil is arranged on the first outer bar, a second winding of the primary coil is arranged on the second outer bar, a first winding of a secondary coil is arranged on the first outer bar, a second winding of the secondary coil is arranged on the second outer bar, and a third winding of the secondary coil is arranged on the center bar.
US 4 563 731 A discloses a control transformer that has a variable inductance. The transformer includes a ring core with a first outer leg and a second outer leg, and central leg disposed between the outer legs. Two AC windings are provided on each of the outer legs. The central leg is provided with a DC winding. One of the AC windings on the first outer legs and one of the AC windings on the second outer legs are connected in series to thereby constitute a primary winding wound so as to cancel the magnetic flux induced in the central leg by an AC current through the primary winding. A direct-current control circuit causes a direct-current through the DC winding in order to change the inductance value across the primary winding.

SUMMARY

The above-identified problem can be solved by the inductive component according to claim 1, the multiphase LLC switching converter according to claim 9 and the method according to claim 10. Various embodiments and further developments are covered by the dependent claims.
An inductive component is described herein. In accordance with one example, the inductive component includes a magnetic core assembly comprising at a first portion and a second portion, which is magnetically separated from the first portion of the magnetic core assembly via two or more air gaps. The inductive component further includes an inductor with a variable inductance including a first coil that is arranged on the first portion of the magnetic core assembly, a second coil arranged on the second portion of the magnetic core assembly and configured to generate a bias magnetic field in the second portion of the magnetic core assembly. The first coil is configured to generate a magnetic flux in a magnetic path that includes the first portion of the magnetic core assembly, the two or more air gaps and a part of the second portion of the magnetic core assembly. Furthermore, multiphase LLC switching converter, which includes the above-mentioned inductive component is described herein. Furthermore, a multiphase LLC switching converter is described herein which includes at least two LLC switching converter units, each including a tank circuit and each tank circuit including an inductor as mentioned above.
Moreover, a method for controlling the inductance of the inductor is described herein. In accordance with one example, the method includes supplying an AC current to the first coil, which is arranged on the first portion of a magnetic core assembly. The method further includes generating the bias magnetic field in the second portion of the magnetic core assembly by supplying a DC current to the second coil that is arranged on the second portion of the magnetic core assembly. The second portion of the magnetic core assembly is magnetically separated from the first portion of the magnetic core assembly via the two or more air gaps. Further, the method includes controlling the inductance of the first coil by adjusting the DC current supplied to the second coil. Thereby, the AC current, when passing through the first coil, generates the magnetic flux in the magnetic path that includes the first portion of the magnetic core assembly, the two or more air gaps and the part of the second portion of the magnetic core assembly.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention can be better understood with reference to the following drawings and descriptions. The components in the figures are not necessarily to scale; instead emphasis is placed upon illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts. In the drawings:

Figure 1 illustrates one example of a multi-phase LLC converter including two LLC converter units whose outputs are coupled in parallel.
Figure 2 is a diagram illustrating the characteristic curve of the resonance of the tank circuit used in an LLC converter unit.
Figures 3 to 6 illustrate different configurations of a current-controlled variable inductor (CCVI) for use in the multi-phase LLC converter of Fig. 1.
Figures 7 and 8 includes magnetic circuit diagrams illustrating the equivalent magnetic paths of the examples shown in Figs. 3 and 4 and, respectively, Figs. 5 and 6.

DETAILED DESCRIPTION

Fig. 1 illustrates one example of an Interleaved LLC (ILLC) converter, also referred to as multi-phase LLC converter. In the present example, the ILLC converter is composed of two LLC converter units 10 and 20 whose outputs are coupled in parallel. However, dependent on the application more than two LLC converter units may be coupled in parallel. A balancing of the output currents of the individual LLC converter units is achieved by using a current-controlled variable inductor (CCVI) in the tank circuits of the LLC converter units. Before discussing the implementation of the CCVIs in more detail, the general construction of an ILLC converter is explained with reference to Fig. 1.
According to Fig. 1 the first LLC converter unit 10 includes a tank circuit that is composed of a capacitor C_IN1 an inductor L_V1 and a further inductor L_P1 The inductance of the inductor L_V1 can be varied by applying a control current, and the inductor L_P1, either as a whole or parts thereof, forms the primary winding of a transformer. The respective secondary winding is represented by inductor Lsi, which has a middle-tap in the present example. Diodes D₁₁ and D₁₂ form a rectifier, which rectifies the current induced in the secondary winding Lsi, wherein the diodes D₁₁ and D₁₂ are connected between the outer taps of the secondary winding Lsi and an output node of the first LLC converter unit 10. A current sense resistor Rsi may be connected between the middle tap of the secondary winding Lsi and a reference node (e.g. ground node GND) of the first LLC converter unit 10. The voltage drop across the current sense resistor Rsi may be tapped to obtain a voltage signal representative of the output current of the first LLC converter unit 10.
It is understood that other circuitry may be used to obtain information representing the output current. Alternatively, current measurement may also take place on the primary side of the transformer. The alternating input voltage V_IN1 applied across the tank circuit, which is a series resonance circuit formed by the series circuit of capacitor C_IN1, inductor Lvi, and inductor L_P1, is usually provided by a transistor H-bridge circuit or, alternatively, by a transistor half-bridge circuit (not shown in Fig. 1) dependent on the actual implementation. It is understood that the transistor bridge circuits may be implemented using MOS transistors that provide a current sensing function in order to obtain current information. In this case, the current sense resistor Rsi can be omitted.
In the example of Fig. 1, the second LLC converter unit 20 is substantially identical with the first LLC converter unit 10. According to Fig. 1, the second LLC converter unit 20 includes a tank circuit, which is a series resonance circuit formed by the series circuit of capacitor C_IN2, inductor L_V2, and inductor L_P2, wherein the latter represents the primary winding of a further transformer. The respective secondary winding is represented by inductor L_S2, which has a middle-tap, and diodes D₂₁ and D₂₂ form a rectifier, which rectifies the current induced in the secondary winding L_S2. Diodes D₁₁ and D₁₂ are connected between the outer taps of the secondary winding L_S2 and an output node of the second LLC converter unit 20. A current sense resistor R_S2 may be connected between the middle tap of the secondary winding L_S2 and a reference node (e.g. ground node GND) of the second LLC converter unit 20. As mentioned, the outputs of both LLC converter units 10 and 20 are connected in parallel, wherein an output capacitor C_OUT may be provided to reduce the ripple of the common output voltage V_OUT. The input voltage V_IN2 applied across the tank circuit may also be provided by a transistor bridge circuit. Usually, the alternating input voltages V_IN1 and V_IN2 are phase shifted with respect to each other.
According to the example of Fig. 1, the ILLC converter includes a regulator circuit 30 that is configured to provide control currents for the inductors L_V1 and L_V2 (which are CCVIs) so as to tune the respective inductances such that the two LLC converter unit 10 and 20 are at least approximately equally loaded. Accordingly, a mismatch between the inductances of inductor L_V1 and L_V2 can be compensated for. It is noted, that in case of an ILLC converter with two converter units it would be sufficient if only one of the two LLC converter units includes a CCVI whereas the corresponding inductance in the other LLC converter unit may be fixed. Generally, in case of an ILLC converter with N LLC converter units at least N-1 LLC converter units must be equipped with a CCVI in order to be able to balance the loads. However, the depicted example, in which each LLC converter has a CCVI, provides somewhat more flexibility.
The resonance frequency f_R1 of the first converter unit's tank circuit composed of capacitor C_IN1 and inductors L_V1 and L_P1 can be approximated by the following equation $f_{R}$ $1 = \frac{1}{2 π \cdot \sqrt{C_{IN 1} (L_{V 1} + L_{P 1})}},$
and, accordingly, the resonance frequency f_R1 increases with a decreasing inductance Lvi. Similarly, the resonance frequency f_R2 of the second converter unit's tank circuit is $f_{R 2} = \frac{1}{2 π \cdot \sqrt{C_{IN 2} (L_{V 2} + L_{P 2})}} .$
To balance the two LLC converter units 10 and 20, the variable inductance (L_V1 or L_V2) of that LLC converter unit, which provides the lower output current, is decreased in order to increase the resonance frequency (f_R1 or f_R2) and thus also the output current of the respective LLC converter unit and to reduce the mismatch between the output currents. In fact, the first converter unit's tank circuit as shown in Fig. 1 has two resonance frequencies, one being associated with capacitance C_IN1 and inductance L_V1, and a further one being associated with capacitance C_IN1 and inductance L_V1+L_P1. The same applies for the second converter unit's tank circuit.
The mechanism described above is further illustrated by the diagram of Fig. 2, which shows the characteristic curve of the resonance of the tank circuit used in the LLC converter unit that provides the lower output current. As mentioned, the regulator circuit 30 controls the CCVI of the respective LLC converter unit to lower values thus shifting the resonance frequencies and hence shifting the respective characteristic curve (solid line) to higher frequencies (dashed line). As consequence the operating point is shifted, for a given operating frequency f_OP from point A to point B, i.e. towards higher output voltages and hence increasing the output current. As already mentioned, the specific way the current information is obtained is not relevant for the further discussion, and there are various different ways to obtain the current information which are as such known and are thus not explained herein in more detail.
The further discussion focusses on the implementation of the CCVIs that may be used as inductor L_V1 and/or L_V2 shown in Fig. 1. Generally, a CCVI is implemented by a coil (referred to as "AC coil") wound around a magnetic core. A control current is fed into a further coil (referred to as "DC coil") that is also wound around the magnetic core. The control current passing through the DC coil is a direct current (DC), whereas the input current supplied to an LLC converter unit (e.g. using a transistor bridge) and thus to AC coil is an alternating current (AC) that alternates in accordance with the operating frequency f_OP. The control current passing through the DC coil causes a magnetic bias field in the magnetic core, and the relative permeability of the magnetic core decreases as the bias field increases and the magnetic core gradually saturates. When the magnetic core becomes fully saturated due to a sufficiently high bias field, the relative permeability of the magnetic core reaches its minimum (relative permeability of one). The inductance of the AC coil depends on (and is approximately proportional to) the relative permeability of the magnetic core. Accordingly, an increasing control current passing through the DC coil causes the inductance of the AC coil to decrease and vice versa. The inductance of the AC coil reaches its maximum at a control current of zero and its minimum when the control current is so high that the magnetic core is fully saturated. The general principle of a CCVI is as such known and thus not explained herein in more detail.
When using a CCVI in a switching converter it is desirable that the CCVI is designed to be efficient in terms of low power losses and low electromagnetic emissions (mainly caused by stray fields). Further, the CCVI should be able to be operated as high frequencies which, in turn, allows the magnetic core to be small. Some known CCVI concepts exhibit the problem that the AC currents passing through the AC coil can induce comparably high voltages in the DC coil, in particular when the DC-coil has a high number of windings in order to keep the control currents low. The induced voltages may be so high that a breakthrough may occur between neighboring windings of the DC coil.
Fig. 3 illustrates a not claimed embodiment of a CCVI that may be used to improve performance of an ILLC converter. According to Fig. 3 a so-called COC core assembly is used as magnetic core for the AC coil 6 that forms the CCVI and the DC coil 5 that is used to control the inductance of the AC coil 6 as explained above. The COC core assembly is composed of several core elements, in the present example an O-shaped core 2 (that may be formed by two opposing C-shaped cores 2a and 2b) and two C-shaped cores 1 and 4 that are arranged at opposing sides of the O-shaped core 2 so that air gaps δ remain between the vertical segments of the O-shaped core 2 and the adjacent (but magnetically separated therefrom by the air gaps) front sides of the horizontal segments of the C-shaped cores 1 and 3. In other words, the O-shaped core 2 is arranged between the two opposing C-shaped cores 1 and 3. It is understood that the terms "vertical" and "horizontal" merely refer to the depicted drawings as the actual CCVI may be mounted in an arbitrary position. The term O-shaped core basically denotes a ring core, i.e. a magnetic core that provides a closed, gap-free magnetic path for the magnetic field generated by a coil would around one or more segments of the ring core. An O-shaped core may be composed of two or more parts, e.g. by two adjoining C-cores (cf. example of Fig. 3 or 5) or by a C-core and an adjoining I-core (cf. example of Fig. 5).
According to Fig. 3, the AC coil 6 is split into two partial coils 6a and 6b which are wound around C-shaped core 1 and, respectively, C-shaped core 3, and which are electrically connected in series, wherein partial coil 6a has a winding direction (winding sense) opposite to the winding direction of partial coil 6b. The partial coils 6a and 6b may be wound on both horizontal segments of the C-shaped cores 1 and 3, respectively, with an equal number of turns around the upper horizontal segments and the lower horizontal segments. Additionally or alternatively, the vertical segments of the C-shaped cores 3 and 1 may be used to arrange the turns of the coils 6a and 6b, respectively. In Fig. 3, the arrows indicate the orientation of the resulting magnetic field obtained for positive inductor currents. The dash-dotted lines illustrate schematically the magnetic (main) paths for the magnetic flux generated by the partial coils 6a and 6b during operation. This magnetic paths, which includes the C-shaped cores 1 and 3, the air gaps and parts (vertical segments) of the O-shaped core 2, is further explained later with reference to Fig. 7.
The DC coil 5 is wound around the O-shaped core 2 which forms a closed magnetic path for the resulting bias magnetic field. The DC coil 5 has a comparably high number of turns so that even small currents will cause a magnetic field that is high enough to change the inductance of the AC coil 6. The magnetic path of coil 6a is formed by the C-shaped core 3, the two air gaps with width δ and the right vertical segment of the O-shaped core 2. Likewise, the magnetic path of coil 6b is formed by the C-shaped core 1, the two air gaps with width δ and the left vertical segment of the O-shaped core 2. When a magnetic bias field is generated in the O-shaped core by applying a DC control current i_DC to the DC coil 5, then the vertical segments of the O-shaped core 2 will increasingly cause magnetic losses with an increasing bias magnetic field. The magnetic resistance of the vertical segments of the O-shaped core 2 increases with an increasing bias magnetic field and, as a consequence, the magnetic resistance of the magnetic paths associated with coils 6a and 6b increases and the inductance of the AC coil 6 (composed of coils 6a and 6b) decreases accordingly.
As mentioned, the magnetic paths of AC coils 6a and 6b are formed basically by the C-shaped cores 1 and 3 and the adjacent vertical segments of the O-shaped core 2. However, a part of the (alternating) magnetic field generated by the AC coils 6a and 6b will also pass the horizontal segments of the O-shaped core 2. Due to the symmetric arrangement of the AC coils 6a and 6b and due to the fact that they have opposite winding directions, the AC fields in the horizontal segments of the O-shaped core 2 caused by AC coil 6a and AC coil 6b at least approximately compensate. Thus only little (and ideally no) AC currents are induced in the DC coil 5. Furthermore, the series connection of the AC-coils in combination with their opposite winding directions result in an improved linearity of the CCVI.
The example of Fig. 4 is almost identical to the example of Fig. 3. The only difference is that the widths d₂ of the left and right vertical segments of the O-shaped core 2 are smaller than the corresponding width di in the example of Fig. 3 (and thus the respective cross-section area is smaller). Particularly, the cross-section areas of the vertical segments of the O-shaped core 2 are between 0.5 to 0.9 times the cross-section area of the C-shaped cores 1 and 3. The smaller cross-section ensures that the magnetic bias field caused by the DC current i_DC passing through the DC coil 5 is substantially kept inside the horizontal segments of the O-shaped core 2 even when the vertical segments reach magnetic saturation. Consequently, the magnetic flux is better focused to the vertical segments of the O-shaped core and hence the required maximum control current i_DC can be reduced.
Fig. 5 illustrates an embodiment according to the present invention of a CCVI that may be used to improve performance of an ILLC converter. According to Fig. 5 a so-called ICIICI core assembly is used as magnetic core for the AC coil 6 that forms the CCVI and the DC coil 5 that is used to control the inductance of the AC coil 6. The magnetic arrangement of Fig. 5 is similar to the example of Fig. 3. However, the positions of AC coil and DC coil are interchanged. According to Fig. 5, the AC coil 6 is arranged between the outer coils 5a and 5b, which are connected in series and form the DC coil 5, whereas, in contrast, the outer coils are the AC coils in the previous example of Fig. 3. The outer C-shaped cores 9 and 11 are each completed by a respective vertical I-shaped core 9b, 11b and are thus effectively O-shaped cores, that are arranged side-by side with two horizontal I-shaped cores 7 and 8 in between, wherein air gaps δ remain between the right and left front sides of the I-shaped cores 7 and 8 and the adjacent C-shaped cores 9 and 11. The AC coil 6 is formed by partial coils 6a, and 6b, which are would around the I-shaped cores 8 and 7, respectively. The dash-dotted line illustrates schematically the magnetic (main) path for the magnetic flux generated by the AC coil 6 during operation. This magnetic path, which includes the I-shaped cores 7 and 8, the air gaps and parts ( segments 9a, 11a) of the outer ring cores 9, 9a, 11, 11a, is further explained later with reference to Fig. 8.
The O-shaped cores 9, 9a and 11, 11a each form a closed magnetic path for the partial coils 5b and 5a, respectively, which together form the DC coil 5.Similar as in the previous example of Fig. 3, the vertical segments 11a and 9a of the C-shaped cores 11 and 9 are part of the magnetic path of the partial DC coils 5a and 5b as well as part of the magnetic part of the AC coil 6, which is formed by the vertical segments 9a and 11a, the air gaps δ as well as the vertical I-shaped cores 7 and 8. When a control current i_DC is supplied to the DC coil 5, then the vertical segments 11a and 9a of the C-shaped cores 11 and 9 are magnetized by the resulting magnetic bias field which causes a respective increase of the magnetic resistance of the magnetic path associated with the AC coil 6 and a corresponding decrease of the inductance. Similar as in the example of Fig. 4, the width d₃ of the vertical segments 11a and 9a may be lower than the width di of the other segments (and the respective cross-sections area as well). For example, the cross-section area of the vertical segments 11a and 9a may be a factor 0.5 to 0.9 of the cross-section area of the other segments of the C-shaped cores 9 and 11.
Most of the magnetic field generated by the AC coil 6 (magnetic AC flux) is guided through the vertical segments 9a, 11a of the C-shaped cores 9 and 11. However, a minor portion of the AC-flux will also run through the DC coils 5a and 5b arranged on the horizontal segments of the C-shaped cores 9 and 11 thereby inducing an AC-voltage in the DC coils 5a and 5b. Unlike in the previous example of Fig. 3, the AC flux is not (approximately) compensated at the position of the DC coils and thus the mentioned AC voltage is induced. However, at sufficiently high operating frequencies, the mentioned AC voltage is effectively shorted by the winding capacitance of the multi-turn DC- coils 5a, 5b. In order to prevent problems at lower frequencies additional short circuit windings with a single turn can be placed on the C-shaped cores 9, 11, e.g. over the DC coils 5a an 5b or on the outer I- cores 9b and 11b. For example, the short circuit windings may be made of copper foil or the like. The example of Fig. 6 is substantially identical to the example of Fig. 5 but additionally includes the mentioned short circuit windings which are denoted with numerals 12a and 12b.
The CCVI described herein allow lower loss and low electromagnetic emissions. Furthermore, the CCVI described herein allow operation of the ILLC switching converter at a comparable high frequency and therefore the size of the CCVI can be reduced. One feature that is (at least in part) responsible for the comparably low losses is the division of the total air gap in the magnetic path of the AC flux into four smaller air gaps δ. As a consequence the parasitic stray fields are lower and also the parasitic eddy currents resulting from the stray fields. Furthermore, losses are further reduced due to the symmetric arrangement of the coils. Accordingly, only a small portion of the core segments that form the magnetic path for the AC flux is premagnetized (which entails enhanced core losses) by the magnetic bias field. These are the vertical segments 9a and 11a of C-shaped cores 9 and 11 in the embodiment of Fig. 5, and the vertical segments of O-shaped core 2 in the example of Fig. 3. The reduction of the cross-sectional area of the mentioned vertical segments (cf. Fig. 4) further helps to concentrate saturation to the relevant parts of the core assembly.
Another aspect addresses electromagnetic emissions (EMI). Particularly when the relevant core segments are strongly magnetized (up to saturation) by the magnetic bias field. Different from known CCVI implementations, which basically produce a magnetic AC flux similar to a rod magnet when magnetized up to saturation by the bias field, the ICIICI-CCVI of Fig. 5 basically produces a magnetic quadrupole field when the vertical core segments are magnetized up to saturation by the bias field. The (approximate) quadrupole field is generated because the AC coil 6 is composed of two partial coils with I-shaped cores (see Fig. 5 horizontal cores 7, 8) arranged anti-parallel closely adjacent to each other. Such an arrangement is also more efficient in terms of losses.
Figures 7 and 8 includes magnetic circuit diagrams illustrating the equivalent magnetic paths of the examples shown in Figs. 3 and 4 and, respectively, the embodiments shown in Figs. 5 and 6 Both diagrams illustrate the magnetic resistances (often referred to as reluctances) of the components (air gaps and cores) that form the magnetic paths for a magnetic flux as well as the sources of magnetomotive force (corresponding to the DC coils generating the magnetic biasing field). It can be seen that the four magnetic resistances R_δ are symmetrically arranged with respect to the cores the form the magnetic core assembly.
The magnetic circuit of Fig. 7 represents the the not claimed examples of Fig. 3 and 4. In these examples the AC coil is formed by series circuit of a two partial coils 6a, 6b (not shown in Fig. 7), which are arranged side by side with the DC coil 5 in between. In this case, the magnetic core assembly includes one ring core 2 (composed of C-shaped core elements 2a and 2b), around which the DC coil 5 is wound, and two further cores 1 and 3, around which the two partial coils 6a, 6b of the first coil are wound. The two further cores 1 and 3 are arranged on opposing sides of the ring core 2 so that two air gaps δ are formed between the ring core 2 and each one of the two further cores 1 and 3.
In Fig. 7 the air gaps are represented by magnetic resistances R_δ, and the (O-shaped) ring core 2 is represented by the magnetic resistances R_2a and R_2b. It is noted that R_2a and R_2b in fact represent only the vertical segments of the ring core 2 (cf. Fig. 4), which is a simplification because the magnetic resistance of the horizontal segments of the ring core 2 is comparably small. The DC coil 5 is represented by the source MMFs of magnetomotive force generating the DC flux (representing the magnetic bias field). The magnetic resistances R_2a and R_2b are variable and depend on the magnetic flux generated by the source MMFs of magnetomotive force (DC coil 5) as explained above. The further cores 1 and 3 are represented by magnetic resistances R₁ and R₃ which are, however, negligibly small as compared to the magnetic resistance R_δ of an air gap. The magnetic resistance of the magnetic path of the AC coils 6a and 6b, which determines the inductance of the CCVI, basically depends on the air gaps (magnetic resistances R_δ) and the magnetic resistances R_2a and R_2b that depend on the DC flux generated by the DC coil 5.
The magnetic circuit of Fig. 8 represents the embodiment of Fig. 5 and 6. In these examples the DC coil 5 is a series circuit of a two partial coils 5a and 5b, which are arranged side by side with the AC coil 6 in between. In this case, the magnetic core assembly includes two ring cores (see Fig. 5, cores 9 and 9b and cores 11 and 11b), around which the two partial coils 5a and 5b of the DC coil 5 are wound. The magnetic core assembly further includes a further core comprising two core segments (see Fig. 5, I-shaped cores 7 and 8), around which the AC coil 6 is wound. The further core segments 7 and 8 are arranged between the two ring cores leaving an air gap δ between each one of the two core segments 7, 8 and each one of the two ring cores.
As in the previous example of Fig. 7, the magnetic resistances of the outer ring cores are essentially determined by the magnetic resistances R₉ and R₁₁ of the vertical segments 9a and 11a (see Fig. 5), whose magnetic resistances depends on the magnetomotive force generated by the sources MMFsa and MMFsb (representing partial coils 5a and 5b) and on the resulting DC flux. The magnetic resistance of the magnetic path of the AC coil, which determines the inductance of the CCVI, basically depends on the air gaps (magnetic resistances R_δ) and the magnetic resistances R₉ and R₁₁. The magnetic resistances R₈ and R₇ are negligible as compared to the magnetic resistances of the air gaps.

Claims

An inductive component comprising:
a magnetic core assembly comprising a first portion (7, 8) and a second portion (9, 9a, 9b; 11, 11a, 11b), which is magnetically separated from the first portion (7, 8) of the magnetic core assembly via two or more air gaps (δ), the second portion (9, 9a, 9b; 11, 11a, 11b) of the magnetic core assembly including a gap-free first ring core (11, 11a, 11b) and a gap-free second ring core (9, 9a, 9b) that are arranged on opposing sides of the first portion (7, 8) of the magnetic core assembly,

whereby the first portion (7, 8) is separated from the second portion (11, 11a, 11b) in that the at least at two air gaps (δ) are provided between the first ring core (11, 11b) and the first portion (7, 8), and whereby the first portion (7, 8) is separated from the second portion (9, 9a, 9b) in that the at least two air gaps (δ) are provided between the second ring core (9, 9b) and the first portion (7, 8);

an inductor with a variable inductance including a first coil (6; 6a, 6b) that is arranged on the first portion (7, 8) of the magnetic core assembly;

a second coil (5, 5a, 5b) arranged on the second portion (9, 11) of the magnetic core assembly and configured to generate a bias magnetic field in the second portion (9, 9a, 9b; 11, 11a, 11b) of the magnetic core assembly,

wherein the first coil (6) is configured to generate a magnetic flux in a magnetic path that includes the first portion (7, 8) of the magnetic core assembly, the two or more air gaps (δ) and a part of the second portion (9a, 11a) of the magnetic core assembly.
The inductive component of claim 1,
wherein the first coil comprises a first partial coil (6a) and a second partial coil (6b) connected in series to the first partial coil (6a), and

wherein the first partial coil (6a) is arranged on a first core segment (7) and the second partial core (6b) is arranged on a second core segment (8) of the first portion of the magnetic core assembly.
The inductive component of claim 1 or 2,
wherein the first ring core (11, 11a, 11b) and the second ring core 9, 9a, 9b) of the second portion of the magnetic core assembly are arranged symmetrically with respect to the first portion (7, 8) of the magnetic core assembly.
The inductive component of any of the preceding claims,
wherein the second coil (5) comprises a first partial coil (5a) and a second partial coil (5b) connected in series to the first partial coil (5a), and

wherein the first partial coil (5a) of the second coil (5) is arranged on the first ring core (11, 11b) and the second partial coil (5b) of the second coil (5) is arranged on the second ring core (9, 9b).
The inductive component of any of the preceding claims,
wherein the first portion of the magnetic core assembly includes a first I-shaped core (8) and a second I-shaped core (7) arranged substantially parallel to the first I-shaped core (8).
The inductive component of any of the preceding claims,
wherein the part of the second portion of the magnetic core assembly, which is included in the magnetic path of the flux generated by the first coil (6), is formed by one segment (11a) of the first ring core 11, 11a, 11b) and one segment (9a) of the second ring core (9, 9a, 9b)
The inductive component of any of the preceding claims,
wherein a first short circuit winding (12a) is arranged on the first ring core (11, 11b) and a second short circuit winding (12b) is arranged on the second ring core (9, 9b)
The inductive component of any of the preceding claims,
wherein the part of the second portion of the magnetic core assembly, which is included in the magnetic path of the flux generated by the first coil (6), as a lower magnetic cross-section area than the remainder of the second portion of the magnetic core assembly, which is not included in the magnetic path of the flux generated by the first coil (6).
A multiphase LLC switching converter comprising:
at least two LLC switching converter units, each including a tank circuit and each tank circuit including an inductive component according to claim 1.
A method for controlling the inductive component according to claim 1 comprising:
supplying an AC current to the first coil (6, 6a, 6b) that is arranged on the first portion of a magnetic core assembly, and

generating the bias magnetic field in the second portion of the magnetic core assembly by supplying a DC current to the second coil (5, 5a, 5b) that is arranged on the second portion (2a, 2b; 9, 11) of the magnetic core;

controlling the inductance of the first coil by adjusting the DC current supplied to the second coil (5, 5a, 5b);

wherein the AC current, when passing through the first coil (6; 6a, 6b), generates a magnetic flux in a magnetic path that includes the first portion of the magnetic core assembly, the two or more air gaps (δ) and a part of the second portion of the magnetic core assembly.