CN103269200A - High speed stabilizing drive control method of satellite-borne large inertia load mechanism - Google Patents

Abstract

A high-stable speed drive control method for a star-loaded large-inertia load mechanism. (1) Apply DC power to the motor, and calculate the phase difference between the motor and the resolver after the load mechanism is stable; (2) Collect the motor at the current speed loop sampling time. (3) Collect the motor current i _A , i _B at the current sampling moment of the current loop to obtain the harmonic spectrum, amplitude, and phase of i _A , i _B ; (4) For i _A , i _B for harmonic compensation; (5) Convert the compensated motor A and B phase winding currents to the rotor coordinate system; (6) Calculate the two-axis voltage of the rotor coordinate system; (7) Determine the three-phase voltage of the PWM waveform generator Phase time value; the SVPWM waveform generator generates a voltage wave according to the three-phase time value, and loads the voltage wave on the motor winding to realize the control of the motor; (8) Enter the next current loop sampling time, go to step (3) and repeat Execute; (9) Go to the next speed loop sampling moment, and turn to step (2) to repeat.

Description

A High Steady Speed Drive Control Method for Spaceborne Large Inertia Load Mechanism

technical field

The method relates to a control driving method for a rotating mechanism with large inertia, and especially requires high speed stability precision, and belongs to the technical field of antenna control.

Background technique

The scanning mechanism of the microwave radiometer and the scanning mechanism of the scatterometer are respectively used to drive the radiometer and the scatterometer to perform conical scanning on the ground, and carry the structure of the micro-detection head and internal electronic equipment to realize the observation of the ground. The mass of the rotating part of the scatterometer is 76.2kg, and the moment of inertia of the radiometer is 7.8kgm ² (the mass of the rotating part is 61.3kg), respectively rotating at a constant speed of 95°/s.

The scanning mechanism, also known as the load mechanism, adopts the same direct drive method to realize the rigid coupling between the motor and the load to improve the responsiveness. The structural diagram of the scanning servo mechanism is shown in Figure 1. The scanning mechanism adopts a brushless DC torque motor and adopts an outer rotor structure. The main shaft in the middle is not only the supporting structure of the antenna and the high-frequency box, but also the load-bearing part that carries the entire rotating part. The top is Conductive slip ring 1, resolver 3 in the middle, motor (motor rotor 5 and electronic stator 6) and bearing 2 for supporting and carrying at the bottom. During the working process, the central axis of the scanning mechanism is fixed, the motor drives the casing 4 to rotate, and the casing drives the antenna and the high-frequency box to rotate (7 in the figure is the interface of the high-frequency box), and the resolver is used to detect the rotating position of the scanning mechanism. The slip ring is used to transmit the electrical signal of the scatterometer to the satellite cabin (8 is the installation interface of the satellite cabin board). During the rotation process, the bearing support mechanism makes relative motion, and the contact parts are slip ring and bearing. The conductive slip ring is a component of the electrical rotary joint, and the slip ring operates 24 hours a day.

The scanning mechanism control circuit is composed of DSP, FPGA, coarse and fine dual-channel resolver circuit, high-precision Hall current acquisition circuit, and three-phase bridge drive circuit, as shown in Figure 2.

Due to the strong nonlinearity of the permanent magnet synchronous motor, its parameter changes will affect the control accuracy and dynamic response speed, so it is necessary to find a suitable control method to achieve high-precision closed-loop control of torque and speed.

The greater the moment of inertia of the load, the greater the mechanical time constant of the motor control system, it is difficult to reconcile the response time and overshoot of the system, and it is difficult to guarantee the control dynamic characteristics of the motor when the speed changes; and there are disturbances in the inertial system Factors such as the cogging torque fluctuation of the motor, the quantization error of the measurement sensor, and the dead zone of the inverter, etc., these non-linear factors will generate torque fluctuations in the system, thereby causing speed fluctuations.

When the load with such a large inertia rotates, the torque will generate a large inertia torque. The instantaneous energy storage of the motor is large, and the fluctuation of its speed will inevitably have a great impact on the attitude of the whole star, which will cause damage to other objects on the star. The pointing accuracy requires that the pointing of the payload instrument cause interference; if the velocity fluctuates greatly, it may even affect the lifetime of the entire star.

The technical documents relevant to the application are described as follows:

[1] Yuan Li, Yang Lei. Research on low-speed, high-inertia, high-precision scanning mechanism technology for satellite cameras. Papers of the 12th National Academic Annual Conference on Space and Moving Body Control Technology. 2006;

[2] Han Changpei. Current loop design of low-speed, high-inertia, high-precision scanning control system. Infrared. 2005;

[3] Pang Wei. Analysis of motion characteristics of space drive mechanism with large inertia load. Master's thesis of Nanjing University of Aeronautics and Astronautics. 2009;

Literature [1] puts forward the scanning mode, lightweight design of scanning mirror, supporting technology and driving mode considerations that need to be considered for optical-mechanical scanners to achieve high-precision scanning;

Literature [2] proposes a current loop control method suitable for the scanning radiometer scanning control system of Fengyun-4 meteorological satellite, and gives the principle block diagram of the current loop;

Literature [3] analyzed the load characteristics of the solar cell array driving mechanism in the space environment, constructed the force model of the driving mechanism under the action of large inertia load, and simulated the load speed and load torque curve of the driving mechanism under various working conditions. And simulate the resonant frequency of the system.

Contents of the invention

The technical problem of the present invention is: to overcome the deficiencies of the prior art, to provide a high-steady-speed drive control method for a large-inertia load mechanism on board, using this method to ensure that the load mechanism rotates at a steady speed, satisfying that the fluctuation of the speed cannot exceed 0.2 /s index requirements.

The technical solution of the present invention is: a high-steady-speed drive control method for a star-borne large-inertia load mechanism. The load mechanism includes a motor, a rotary transformer, and a high-frequency box. The steps of the method are as follows:

(1) Apply direct current to the motor, and after the load mechanism is stable, calculate the phase difference between the motor and the resolver;

(2) Collect the current speed of the motor at the current speed loop sampling time, and use adaptive PID to calculate the torque current according to the rated speed and current speed of the motor;

(3) At the current current loop sampling time, collect the motor A and phase B winding currents i _A and i _B and perform discrete Fourier DFS calculations on them to obtain the harmonic spectrum, amplitude and phase of i _A and i _B ;

(4) Calculate the compensation coefficient according to the result of step (3) and the rotor magnetic field angle θ _R , and use the compensation coefficient to perform harmonic compensation for the winding currents i A and i B of phase _A and _B of the motor;

(5) Transform the A and B phase winding currents of the motor after compensation in step (4) into the rotor coordinate system;

(6) Use the torque current calculated in step (2) to calculate the two-axis voltage of the rotor coordinate system;

(7) Use the two-axis voltage calculated in step (6) to determine the three-phase time value of the PWM waveform generator; the SVPWM waveform generator generates a voltage wave according to the three-phase time value, and loads the voltage wave on the motor winding to realize the control of the motor control;

(8) Enter the next current loop sampling time, turn to step (3) and repeat;

(9) Enter the next speed loop sampling time, turn to step (2) and repeat; the speed loop sampling time is greater than the current loop sampling time and is an integer multiple of the current loop sampling time.

The formula for calculating the phase difference in the step (1) is as follows:

${\mathrm{<span\; class="notranslate">\; \&theta;</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; \&times;</span>\mathrm{<span\; class="notranslate">\; p</span>}}{\mathrm{<span\; class="notranslate">\; 360</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; INT</span>}<span\; class="notranslate">\; [</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; \&times;</span>\mathrm{<span\; class="notranslate">\; p</span>}}{\mathrm{<span\; class="notranslate">\; 360</span>}}<span\; class="notranslate">\; ]</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&times;</span>\mathrm{<span\; class="notranslate">\; 360</span>}$

Among them, p-the number of pole pairs of the resolver; INT-rounding; θ-the mechanical angle of the resolver.

The calculation formula of the compensation coefficient C _factor in the step (4) is as follows:

…为步骤（3）计算的相位，θ-旋转变压器的机械角度。In the formula, a ₃ , a ₅ , a ₇ , a ₉ ... are the amplitudes calculated in step (3),

... is the phase calculated in step (3), θ - the mechanical angle of the resolver.

Principle of the present invention is:

(1) First complete the initial positioning of the rotating load to obtain the phase relationship between the motor and the resolver on the mechanism. The phase difference determines the maximum output torque of the system.

(2) Due to reasons such as motor cogging torque, inverter dead zone, sensor quantization error, AB current acquisition gain inconsistency, etc., there will be harmonic components of motor output torque, which will cause torque fluctuations; according to the motor The dynamic equation of the load will cause the speed fluctuation of the load.

Inherent deviations in the phase current sensing channel (such as DC offset, gain mismatch) will generate deviation errors. Because field positioning control is based on current feedback, any current sensing error will directly affect torque performance.

Therefore, it is necessary to detect the harmonic components of the motor output torque and compensate each harmonic component. Since the current on the motor winding directly reflects the output torque of the motor, the harmonic component of the motor phase current represents the harmonic component of the output torque. Carry out DFS calculation on the motor winding current to obtain the frequency, phase and amplitude of each harmonic; calculate the compensation coefficient according to the position, current and rotor magnetic field angle, and correct the motor winding current by the compensation coefficient.

(3) Use the corrected motor winding current to calculate the space field-oriented control strategy, that is, decompose the stator current vector into two orthogonal components in the synchronous coordinate system (dq coordinate system) along the rotor field orientation, one is the rotation The moment current component (that is, the q-axis current component); the other is the excitation current component (that is, the d-axis current component). By controlling the phase and amplitude of the stator current space vector, that is, controlling the phase and amplitude of the motor torque current component and the excitation current component, the decoupling control of the magnetic field and torque is realized. If the d-axis component of the air gap flux linkage is kept constant, the torque is proportional to the q-axis current, so that the direct control of the motor torque can be realized by controlling the q-axis current. Figure 3.

Compared with the prior art, the present invention has beneficial effects as follows:

(1) The field-oriented control strategy of the magnetic field harmonic compensation of the present invention uses the harmonics of the phase current of the motor winding to calculate the torque harmonic component, and comprehensively compensates the cogging torque of the motor, the dead zone of the inverter, the quantization error of the sensor, Harmonic components generated by non-linear factors such as AB current acquisition gain inconsistency, reduce current harmonic components, improve power utilization, and can effectively reduce torque ripple.

(2) The present invention proposes an algorithm for phase calibration between the motor and the multi-pole resolver, which can automatically obtain the installation zero position relationship between the motor and the multi-pole resolver in the mechanism.

(3) The present invention effectively solves the problem that loads such as microwave radiometers and microwave scatterometers interfere with the satellite platform attitude control due to fluctuations in rotational speed.

(4) The high-speed and stable-speed scanning implementation scheme of the spaceborne scanning mechanism proposed by the present invention can be directly applied to spaceborne microwave radiometers, microwave scatterometers, microwave imagers, fully polarized microwave radiometers, and high-speed stable antennas, etc. design implementation.

This application is aimed at the high-stable scanning application of space-borne microwave radiometer and microwave scatterometer, on the basis of ensuring technological advancement and engineering feasibility, combined with the design of permanent magnet synchronous motor with small output torque fluctuation characteristics, control and drive strategy In terms of design, etc., we strive to minimize the output torque fluctuation of the system, so as to meet the high and stable speed requirements of the scanning mechanism.

Description of drawings

Fig. 1 is the driving unit of the scanning mechanism;

Figure 2 is a block diagram of the hardware platform;

Fig. 3 is a schematic block diagram of the magnetic field harmonic compensation directional control of the present invention;

Fig. 4 is a flow chart of the method of the present invention;

Fig. 5 is motor winding current waveform of the present invention;

Fig. 6 is the measurement result of the rotational speed fluctuation test of the present invention.

Detailed ways

In order to better understand the present invention, the coordinate system involved in the present invention will be described first.

Two-phase stator coordinate system (αβ coordinate system):

Two-phase symmetrical windings, passing two-phase symmetrical currents, also generate a rotating magnetic field. For a vector, it is customary to use a two-phase Cartesian coordinate system to describe it mathematically, so a two-phase coordinate system (αβ coordinate system) is defined.

The two-phase fixed windings α and β of the motor have a spatial difference of 90 degrees, and the two-phase balanced AC currents i _α and i _β have a phase difference of 90 degrees.

Three-phase stator coordinate system (abc coordinate system):

The three-phase stator windings of the permanent magnet synchronous motor have a mutual difference of 120 degrees in space, and are defined as a, b, and c axes along their axes, forming an abc coordinate system.

Rotor coordinate system (dq coordinate system):

The rotor coordinate system is fixed on the rotor, its d-axis is located on the rotor axis, and the q-axis is 90 degrees ahead of the d-axis counterclockwise. The coordinate system rotates with the rotor at the angular velocity of the rotor in space.

The high-stable-speed driving control method described in the present invention realizes automatic phase calibration between the mechanism motor and the resolver through design, and realizes starting a large inertia load with the maximum torque output; at the same time, by calculating the harmonic component of the phase current, the magnetic field harmonic Wave compensation algorithm to achieve stable torque output and achieve high and stable speed.

As shown in Figure 4, the implementation steps of the present invention are as follows:

(1) Apply direct current to the motor, and after the load mechanism is stable, calculate the phase difference between the motor and the resolver. The phase difference calculation method is as follows:

${\mathrm{<span\; class="notranslate">\; \&theta;</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; \&times;</span>\mathrm{<span\; class="notranslate">\; p</span>}}{\mathrm{<span\; class="notranslate">\; 360</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; INT</span>}<span\; class="notranslate">\; [</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; \&times;</span>\mathrm{<span\; class="notranslate">\; p</span>}}{\mathrm{<span\; class="notranslate">\; 360</span>}}<span\; class="notranslate">\; ]</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&times;</span>\mathrm{<span\; class="notranslate">\; 360</span>}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>$

p-the number of pole pairs of the resolver; INT-rounding; θ-the mechanical angle of the resolver.

(2) For the characteristics of large inertia loads, at the current speed loop sampling time, according to the rated speed and current speed, the adaptive PID is used to calculate the torque current.

(3) At the current sampling moment of the current loop, the Hall current sensor is used to collect the winding currents i A and B of the motor phase _A and _B , and perform discrete Fourier DFS calculations on them to obtain the amplitude a _i and phase of the harmonic spectrum

-DFS相位；f_i-DFS频率；M-采样数；x[n]-第n个采样点的采样值。a _i - DFS amplitude;

-DFS phase; f _i -DFS frequency; M-sample number; x[n]-sample value of the nth sample point.

(4) Carry out harmonic compensation to I _A and I _B according to the input I _A , I _B , and rotor magnetic field angle θ. Calculated as follows:

I _A *=C _factor ×I _A (2)

I _B *=C _factor ×I _B (3)

C _factor is the compensation coefficient, and I _A ^* and I _B ^* are the results of harmonic compensation.

a ₃ , a ₅ , a ₇ , a ₉ ... are the amplitude of DFS, …is the phase of DFS, θ is the rotor magnetic field angle. Due to the non-linearity of the hardware and the cogging harmonic torque of the motor mainly generating odd harmonics, the above coefficients are only compensated for the odd harmonics.

(5) Transform the compensated phase winding current values I _A ^* , I _B ^* into the two-phase stator coordinate system I _alfa , I _beta , and then into i _d , i _q of the dq axis of the rotor coordinate system. This completes the transformation from the abc coordinate system to the dq axis coordinate system.

I _alfa =I _A

I _beta =(I _A +2*I _B )*0.577350269189625764509

I _d =I _alfa *cos(θ-θ ₀ )+I _beta *sin(θ-θ ₀ )

I _q =I _beta *cos(θ-θ ₀ )-I _alfa *sin(θ-θ ₀ )

(6) According to the torque current calculated in step (2), use the proportional integral (PI) algorithm to calculate the dq-axis voltage; the q-axis current PI regulator outputs the q-axis voltage V _q , and the d-axis current PI regulator outputs the d-axis voltage V _d .

q-axis voltage PI calculation:

e _qCurErr =I _qref -I _q

Δu=K _p [e _qCurErr -e _qLastErr ]+K _i e _qCurErr

V _q =Δu+u _qLast

u _qLast =V _q

e _qLastErr =e _qCurErr

d-axis voltage PI calculation:

e _dCurErr =0-I _d

Δu=K _p [e _dCurErr -e _dLastErr ]+K _i e _dCurErr

V _d =Δu+u _dLast

u _dLast = V _d

e _dLastErr =e _dCurErr

I _qref is the torque current; I _q is the q-axis current; I _d is the d-axis current; V _q is the q-axis voltage; V _d is the d-axis voltage;

e _qCurErr is the error value of the q-axis current at the current moment; e _qLastErr is the error value of the q-axis current at the previous moment;

e _dCurErr is the error value of the d-axis current at the current moment; e _dLastErr is the error value of the d-axis current at the previous moment;

K _p is the proportional coefficient; K _i is the integral coefficient.

(7) Convert V _d and V _q of the dq axis into V _alfa and V _beta of the two-phase stator coordinate system (that is, Vα, Vβ in Figure 3), and complete the conversion from the dq coordinate system to the αβ coordinate system.

V _alfa =[cos(θ-θ ₀ )×V _d -sin(θ-θ ₀ )×V _q ]

V _beta =[sin(θ-θ ₀ )×V _d +cos(θ-θ ₀ )×V _q ]

(8) Determine the sector where V _alfa and V _beta are located

V _a = V _beta

V _b =0.5×(1.732051×V _alfa -V _beta )

V _c =0.5×(-1.732051×V _alfa -V _beta )

The sector is determined by V _a , V _b , and V _c look-up table.

If V _a >0, then a=1, otherwise a=0; if V _b >0, then b=1, otherwise b=0; if V _c >0, then c=1, otherwise c=0. Assuming N=4c+2b+a, the corresponding relationship between N and sectors is shown in the table.

Correspondence between Table 1N and sectors

N 1 2 3 4 5 6 sector 1 5 0 3 2 4

(9) Allocate the action time t ₁ and t ₂ of the V _alfa and V _beta vectors according to the sectors

X=V _beta

Y=0.5×(1.732051×V _alfa +V _beta )

Z=0.5×(-1.732051×V _alfa +V _beta )

The sectors of V _alfa and V _beta are determined from Table 1, and time t ₁ and t ₂ are calculated according to Table 2. t ₁ and t ₂ represent the time when V _alfa and V _beta act. For example: in sector 0, the value of time t ₁ of V _alfa is -Z, and the value of time t ₁ of V _beta is X.

Table 2 t ₁ and t ₂ times

sector 0 1 2 3 4 5 t ₁ -Z Z x -X -Y Y t ₂ x Y -Y Z -Z -X

(10) Calculate the time value of the voltage space vector pulse width modulation (SVPWM) waveform generator according to t ₁ and t ₂ :

t _aon =(1.0-t ₁ -t ₂ )*0.5;

t _bon =t _aon +t ₁ ;

t _con =t _bon +t ₂ ;

Write t _aon , t _bon , t _con into the time register in the SVPWM waveform generator, and generate the SVPWM voltage wave through the inverter.

The SVPWM voltage wave is loaded on the motor winding to form a sine wave current, as shown in Figure 4.

(11) When the current loop sampling time is up, return to step 3, and repeat steps 3 to 10 in turn to form a current closed-loop control.

(12) When the sampling time of the speed loop is up, return to the second step, and recalculate the torque current according to the current speed to form a closed-loop control of the speed. The measured value of the speed fluctuation is shown in Fig. 5.

The sampling time of the speed loop is required to be greater than the sampling time of the current loop, and the sampling time of the speed loop is an integer multiple of the sampling time of the current loop.

The invention can ensure that the scanning mechanism of the microwave radiometer and the scanning mechanism of the scatterometer rotate at a high and stable speed, meeting the index requirement that the fluctuation of the speed cannot exceed 0.2/s.

Parts not described in detail in the present invention belong to the common knowledge of those skilled in the art.

Claims (3)

1. the high speed stabilizing of spaceborne big inertia mechanism loading drives control method, and described mechanism loading comprises motor, resolver and high frequency case, it is characterized in that method step is as follows:

(1) motor is passed to direct current, treat that mechanism loading is stable after, calculate the phase difference of motor and resolver;

(2) at the current rotating speed of present speed ring sampling instant collection motor, rated speed and current rotating speed according to motor utilize self-adaptive PID calculating torque electric current;

(3) gather motor A, B phase winding current i in current electric current loop sampling instant _A, i _BAnd it is carried out discrete fourier DFS calculate, obtain i _A, i _BHarmonic spectrum, amplitude, phase place;

(4) according to result and angle, the rotor field θ of step (3) _RCalculate penalty coefficient, utilize penalty coefficient to motor A, B phase winding current i _A, i _BCarry out harmonic compensation;

(5) the motor A after step (4) compensation, B phase winding current conversion are arrived under the rotor coordinate system;

(6) torque current that utilizes step (2) to calculate calculates two shaft voltages of rotor coordinate system;

(7) two shaft voltages that utilize step (6) to calculate are determined the three-phase time value of PWM waveform generator; The SVPWM waveform generator produces voltage wave according to the three-phase time value, this voltage wave is carried in the control that realizes on the motor winding motor;

(8) enter next electric current loop sampling instant, change step (3) over to and repeat;

(9) enter next speed ring sampling instant, change step (2) over to and repeat; The described speed ring sampling time is greater than the electric current loop sampling time and be the integral multiple in electric current loop sampling time.

2. the high speed stabilizing of spaceborne big inertia mechanism loading according to claim 1 drives control method, and it is characterized in that: the phase difference calculating formula of described step (1) is as follows:

${\mathrm{\&theta;}}_0=(\frac{\mathrm{\&theta;}\&times;p}{360}-\mathrm{INT}[\frac{\mathrm{\&theta;}\&times;p}{360}\left]\right)\&times;360$

Wherein, p-resolver number of pole-pairs; INT-rounds; The mechanical angle of θ-resolver.

3. the high speed stabilizing of spaceborne big inertia mechanism loading according to claim 1 drives control method, it is characterized in that: the penalty coefficient C in the described step (4) _FactorComputing formula is as follows:

In the formula, a ₃, a ₅, a ₇, a ₉Be the amplitude of step (3) calculating,

Be the phase place that step (3) is calculated, the mechanical angle of θ-resolver.

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