CA1295712C - Time domain radio transmission system - Google Patents
Time domain radio transmission systemInfo
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- CA1295712C CA1295712C CA000565712A CA565712A CA1295712C CA 1295712 C CA1295712 C CA 1295712C CA 000565712 A CA000565712 A CA 000565712A CA 565712 A CA565712 A CA 565712A CA 1295712 C CA1295712 C CA 1295712C
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Abstract
Abstract of the Disclosure A communications system wherein there is employed a signal mixer in which a received signal is multiplied by a template signal, and then the output of the mixer is integrated. By this process, usable signals are obtained which would be otherwise obscured by noise.
Description
~.Z~3~5~ ~ ~
TIME DOMhIN RADIO TRANSMISSION SY~TEM
Field of the Invent.ion The invention relates generally to radlo transmission systems, and particularly to a time domain syst~m wherein spaced sin~le cycles, or near single cycles sometimes referred to a~ monocycles, of electromagnetic energy are transmitted in ~pace and where discrete frequency signal components are generally below noise level and are thus not discernabl0 by conventional radio receiving equipment.
Ba k~ound of the Inven_ion The radio transmission of communlcations signals, for example, audio signals, i5 normally effected by one of two methods. In one, referred to a3 an amplitude modulation sy~tem, a sinusoidal radio frequency carrier is modulated in amplitude in terms of the intelligence or communications signal, and when the signal i~ received ~t a receiving location, the reverse process, that i5, demodulation of the carrier, is ef~ected to recover the communications signal. The other ~ystem employs what is termed frequency modulation, and ln.stead oP amplitude modulation of the carrier signal, i t is frequency modulated. When an FM frequency modulation or FM signal i8 received, circuitry is employed which performs what is termed discrimination wherein chan~es in freguency are changed to changes in amplitude in accordance with the original modulation, and thereby a communlcation si~nal i~ recovered. In both systems, there is as a basis a sinusoidal carrier which is assigned and occupies a distinctive frequency band width, or channel, and this channel occupies ~pectrum space which cannot be utilized by other transmissions within the range of its employment.
At this time, almost every nook and cranny of spectrum space X
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is being utilized, and there lg a tremendous need for some method of expanding the availability of communications channels. In consideration of thi~, it has been suggested that instead of the use o~ di~crete frequency channels for radio communications links, which is the con~entional approach, a radio tran3mission link employing a wider frequency spectrum which ~ay extend over a range of lO to 100 times the intelligence band width being transmitted, but wherein the energy o~ any single fre~uency making up that spectrum be very low, typlcally below normal noise level3. Thus, it would be obvious that thl~ type o~
transmission would be essentially non interferin~ with other services.
Additionally, and as well expre~sed in an article entitled "Time Domain Electromagnetics and Its Application,"
Proceeding~ of the IEEE, Vol. 6~, No. 3, (March 19~8), it has been suggested that baseband ~ignals generated ~rom pulses o~ a short duration, e.g., in the pico second range employed for such applications ~s basehand radar.
Ranges on the order of 5 to 5,000 feet were sugg~sted.
This article appeared in 1978, and recent discussions with radar engineers as to subsequent development o~ baseband radar su~gest that little has been achie~ed, p~rticularly in the development of the widely use medium range ~ield of up to 10 kilometers. The reasons given for the lack of success in this area appear to be si~nal reception ~rom such systems cannot adequately combat noi~e. It i3 to be kept in mind that the signal-to-noise ratio problem is made enormous by the fact that the baseband radar signal received must compete with all of the electromagnetic noi~e occurring within the entire spectrum of the band of the radar signal, which i5, ~or example, from 100 MHz to 1.5 GHz, or higher. E~en where there is no intentional jamming of energy present, there is an enormous amount of electromagnetic energy present in X
addition to the poor r~dar ~ignal at the input of a B~R
BAseband Radar. On it5 face, the problem seems rather hopeless, and it is believed that this is about where the problem lies.
In accordance with this invention, a pulse slgnal of a fixed or programmed rate i5 generated, and it is varied or modulated as to tAe time of occurrence when it is eraployed as a basis of an intelligence signal. The re~ultant pulse slgnals affect the turn--on or turn-off o~ a fast electronic switch ~uch as an avalanche mode o~erated semiconductor or ~park cap which turns on or of~ power input to a broadband transmission system. The resultant output, a carrierles~ signal burst, i5 coupled to the atmosphere or space and thus transmission. Reception of the transmission is effected by a receiver which is timed selectively to effect detection e~ploying an analog multiplier which multiplies the received signal by a locally ger.erated signal which bears a polarity-time relationship to the transmitted ~ignal. The multiplication produces a correlation signal which rather uniquely corresponds to the actual transmitted signal where the target is normal and planar to the direction of transmission and reception. Thus, even in the presence of an essentially randomly changing voltage from noise, upon which the transmitted signal i8 typically ridin~, it has beeh found that si~nal recognition over the noise can be achieved despite substantially higher levels of noise than the level of the radar signal. Essentially what happens i8 that in the absence of a received ~ignal which is not closely in phase from a polarity point of view with the internally senerated signal, the output, particularly after inte~rative processing~ will be vastly lower in level than with a correlated si~nal present.
To enhance reception even further, the output of the mixer, or correlator, i~ sampled at a number o~ polnts X
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over .its effective length or duration, and it then becomes pos~ible to relate this serie3 o~ points with a series of anticipated values and a decision a~ to whether there is a target present can be made in terms of a degree of resemblance between a sltandard and the waveform bit pattern of the received signal. This process thus is a two-stage one, wherein one would choose a known template, representative, at least time-polarit~wise, of a particular type of target for th~ in~ection signal.
There is then provided éa detected sign~l which will provide target identication with an extremely high probability just from its level. Then, beyond this, by virtue of discrete samplings a~ ~elected time points and comparison with what would be expected if a targct were present, a still further element of pasitiveness in resolution i5 enabled. All in all, it is bslieved that the present invention solves the grea test inhibition to the satisfactory development of medium and, in fact, long distance radar.
Brief Descr~ on of the Drawin~s Fig. 1 is a combination block-~chematic diagram of a time domain transmitter.
Fi~. la is a schematic diagram oP an alternate form of output stage for the transmitter shown in Fig. 1.
Fig. 2 is a combination block-schematic diagram of a time domain receiver as contemplated by this invention.
~ig. 2a is a combined block-schematic electrical diagram of an alternate form of synchronous detector to the one shown in Fig. 2.
Fig. 3 i~ an electrical block diagram of an alternate embodiment of a time domain receiver.
Fig. 4 is a set of electrical wa~eforms illustr~tive of aspects of the circuitry shown in Fi~s. 1 X
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and 2.
Fig. 5 is an electrical block diagram of a time domain r~dar system.
Fig. 6 ls a schematic illustration of a surveillance system as contemplated by the present lnvention.
Fig. 7 is a schematilc illustration of a phase~ array radar system as contemplated by this invention.
Description of _he Preferred _mbodiment Referring to F~g. 1, and initi~lly to transmitter 10, a base frequency of 100 KHz is generated by o~cillator 12, typically being a crystal controlled oscillator which includes conventional circuitry ~or providing a~ an output square wave pulse~ at 100 KHz rate. Thi8 pulse signal is applied to divide-by-4 divider 14 to provide at its output a square wave 25 KHz, 0-5 volt, signal shown in wave~orm A of Fig. 4. Further references to waveforms will simply identify them by their letter identity and will not further refer to the figure, which is Fig. 4 in all cas~s. This output is employed as a general transmission signal and as 20 an input to power supply 16. The latter is regulated, one which supplies a 300-volt D.C. bias on a non-inter~ering basis for the output stage 18 of transmitter 10, whlch is also keyed at the 25 KHz rate.
The output of divide-by-4 divider 14 is employed as a signal base and as such is supplied through capacitor to pulse po~ition modulator ~2. Pulse position modulator 22 includes in its input an RC circuit consi3ting of resi~tor 24 and capacitor 26 which convert the ~uare wave input to an approximately trian~ular wave as 3hown in waveform B, it being applied across re~istor 25 to the non-inverting input of comparator 28. A sslected or reference positive voltage, ~iltered by capacitor 27, is also applied to the non~invertins input o~ comparator 28, it being supplied from ~5 volt terminal 29 of D.C. bias ~ 3~
supply 30 through resistor 32. Accordingly, Por example, there would actually appear at the non-inverting input a triangular wave biased upward positively as illustrated by waveform C.
The actual cGnduction level of comparator 28 is deter~ined by an audio signal input ~rom microphone 34 supplied through capacitor 36, across resistor 37, to the invert~ng input of comparator 28, as biased from supply 30 throu~h resistor 38 and across resistor 32.
The combined audio signal and bia~ i~ illu~trated in waveform D. By virtue of the thu~ described input combination, the output of comparator 28 would ri~e to a positive saturation level when triangular wave slgnal 40 (waveform E~ i8 of a higher value than modulat~on signal 42 a~d drop to a negatlve saturation level when modulation signal 42 i8 of a greater value than the trl~ngular wave signal 40. The output signal of comparator 28 is shown in waveform F.
In the present case, we are interested in employing the negative going or trailing edge 44 (waveform F) of the output o~ comparator 28, and it is to be noted that this trailing edge will vary in its time position as a function of the signal modulation. This trailing edge of the waveform in wave~orm F triggers "on" mono, or monostable multivibrator, 46 havin~ an "on" time of approximately 50 nanoseconds, and its output is shown in waveform ~. For purposes of illustration, ~hile the pertinent leading or trailing edges of related waveforms are properly aligned, pulse widths and spacings (as indicated by break lines, spacings are 40 microseconds) are not related in scale.
Thus, the leading edge of pulse waveform G corresponds in time to the trailing edge 44 ~wave~orm FJ and its time position within an avera~e time between pulses o~ waveform G is varied as a function of the input audio modulation ~ignal to ~omparator 28.
X
~f~?',~7~ 2 The output of mono 46 i~ applied through dlode 48 across resistor 50 to the base input of NPN transistor S2 operated as a triggering ampli~ier. It i8 conventionally biased through resistor 54, e.g., 1.5K
ohms, ~rom +5 volt terminal 29 of 5 volt power supply 30 to its collector. Capacitor 56 havlng an approxlmate capacitance of 0.1 mf is connected between the collector and ground of transistor 52 to enable ~ull bias potential to appear across the translstor for lts brief turn-on interval, 60 nanoseconds. The output o~
transistor 52 is coupled between its emitter and 0round to the primary 58 of trigg~r transformer 60.
Additionally, transistor 52 may drive trans~ormer 60 via an avalanche transistor connected in a rommon emitter con~iguration via a collector load resistor. In order to drive transformer 60 with a steep wave ~ront, the avalanche mode operated transistor i8 ideal. Like secondary windings 62 and 64 of trigger transformer 60 separately supply base-emitt~r inputs of NPN avalanche, or avalanche mode operated, transistors 66 and 63 o~ power output sta~e 18. Although two are shown, one or more than two may be employed when approprlately couple~.
Avalanche mode operated transistors 66 and 68, many type 2N2222 with a met21 can, have the characteristic that when they are triggered "on," their resistance goes low (e.gO, approximately 30 ohms for each) and st~y~ at this state until collector current drops sufficiently to cut o~ conduction (at a few microamperes~. Certain other transistors, such as a type ~N440l, also display reliable avalanche characteristics. Their collector-emitter circuits are connected in series, and collector bias of ~300 volts is applied to them from power supply 16, across filter capacitor 72, and throu~h resistor 74 to one end 76 of parallel connected delay lines DL. While three sections S1-53 are shown, typically five to ten would be X
5~
employed. They may be constructed of type R558 coaxial cable, and each being approximately three inche~ in length as required to totally effect an approximately 3 nanosecond pulse. ~5 shown, the positive input potential ~rom resistor 74 is connected to the center conductor of each of the delay lines, and the outer conductors are connected to ground.
Re~istor ~4 is on the order of 50R ohms and is ad~usted to allow a current flow through transistors ~6 and 68 o~
about 0.2 MA which i~ a zener current which places both transistors in a near sel~-triggerin~ state. It has been found that under this condition, the transistors will self-ad~u~t to an av;alanche voltage which may be different for the two. Normally, resistor 74 will still be of value which will enable charging of the delay lines ~ between pulses. Delay lines DL ar~ charged to 300 volts bias durin~ the period when transistors 66 and 68 are turned o~f, between input pul~es. When the inputs to transi~tors 66 and 68 are triggered "on" by a triggering pulse they begin to conduct within 0.5 nanoseconds, and by virtue of the low vultage drop across them (when operated in an avalanche mode as they are), about 120 volts appears as a pul5e across output resistor 78, e.g., 50 ohms.
Significantly, the turn-on or leading edge of this pulse is effeoted by the trigger pulse applied to the inputs of transi~tors 66 and 68, and the trailing edge of this output pulse is determined by the dischar~e time of delay llnes DL. By this technique, and by choice of length and Q of the delay lines, a well-shaped, very short pul~e, on the order of 3 nanoseconds and with a peak power o~ appro~imately 300 watts, i5 generated.
Following turn-o~f, delay lines DL are recharged through resistor 74 before the arrival of the next triggering pulse. As will be apparent, pow~r stage 18 is extremely simple and is constructed of quite inexpensive circuit elements. For example, transistors 66 and 68 are X
,5~
available at a cost of approximately $0.1Z.
The output of power output sta~e 18 appears across resistor 78 and is supplied through coaxial cable 80 to a time domain shaping fi:Lter 82 which would be employed to affix a selected signature to the output a9 a form o~ encoding or recognitton signal. Alternately, filter 82 may be omitted where such security ~easures are not deemed necessary; and, el9 indicative o~ thi~, a bypa~s line 84 including a switch 86 diagrammatically illu~trates such omission.
The signal outpu't of filter 82, or directly the output o~ power ~tage 1~, i8 supplied through coaxial cable 88 to discone antenna 90, which is an aresonant antenna. This type of antenna relatively uniformly 16 radiates all ~lgnals of a frequency above its cut-off frequency, which is a function of size, for example, signals above approximately 50 MHz for a relatively small unit. In any event, antenna 90 radlate~ a wide spectrum signal, an example being shown ln the time domain in waveform ~ of Fig. ~, ess~ntially a mono~ycle, this waveform being the composite of the shaping effect~ of filter 82, i~ used, and, to an extent, discone antenna 90.
Fig. la illustrates an alternate and simplified output s~age. As illustrated, biconical antenna 200, a~ a broadband antenna, is charged by a D.C. source 65 through resistors 6~ and 69 to an overall voltag~ which is the sum of the avalanche voltage o~ transistors 66 and 68 as discu-~sed above. Resistors 67 and 69 together have a resistance value which will enable transistors 66 and 68 to be biased as descri~ed above. Resistors ~1 and 73 are of relatively low value and are adjusted to receive energy below the frequency of cut-off o~ the antenna and also to prevent ringing. In operation, when a pul~e is applied to the primary 58 of pulse transformer 60, transi~tors ~6 and 68 are turned on, ef~ectively shorting, through X
t~
resistors 71 and ~3, biconical cmtenna elementæ 20~ and 206. This action occurs essentially at the speed of light, with the result that a signal, es~entially a monocycle or about one and a half cycles a~ Rhown in Fig.
4H, is transmitted as described above for the transmitter output system shown in Fig. 1.
The output o~ disc:one antenna 90, or blcone antenna, is typically transl~itted over a di3crete space and would typically be received by a like discone antenna g2 of receiver 96 at a second location. Although transmission e~fects may distort the waveform some, for purpo~es of illustration, it will be as3umed that the wave~orm received will be a replica o~ wave~orm H. The received signal is amplified by broad band ampli~ier 94, having a broad band frequency response over the range of the transmitted signal. In instances where a filter 82 is employed in transmitter 10, a reciprocally con~iyured filter 98 would be employed. As illustrative of instances where no matched filter would be employed, there i5 diagrammatically illustrated a switch 100 connecting the input and output of filter 98, denoting that by closing it, filter 98 would be bypassed. Assumlng that no match fil~er is employed, the output of broad band ampli~ier as an amplified replica of waveform H is illustrated in wa~e~orm I. In either case, it appears across re~istor 101 .
5ignal waveform I iY~ applied to synchronous detector 102. Basically, it has two functional UAi ts, avalanche transistor 104 and adjustable mono 106. Mono lQ6 is driven from an input acro~s emitter-resi~tor llO, connected between the em~tter of avalanche transistor 104 and ground. Avalanche transistor 104 i5 biased from variable voltag~ D.C. source 112, e.g., 100 to 130 volts, through variable resistor 114, e.g., lOOK to lM ohms. A delay line 116 is connected between the collector and ~round of X
%
transistor 104 and provides the effe!ctive operatingbla~ for translstor 104, it being char~ed between conduction period~
as will be described.
A~suming now that a chargin~ interval ha~ occurred, avalanche transistor 104 will be turned on, or trig~ered, by a si~nal applied to its base ~rom across resistor 101. It will be further assumed that this triggering is enabled by the Q output, waveform J, o~ mono 106 bein~ hi~h. Upon being trig~ered, the conduction of avalanche transistor 104 will produce a rising volta~e across emitter re~istor 110, waveform K, ~nd this volta~e ~ill in turn trig~er mono 106 to cause it~ Q output to go low. This in turn causes diode 108 to conduct and thus effectively shorting out the input to avalanche transistor 104, this occurring within 2 to 20 nanoseconds from the positive leadin~ edge of the input ~ignal, waveform I.
The conduction period of transi~tor 104 i8 precisely set by the charge capacity of delay line 116. With a delay line formed of 12" of R~58 coaxial cable, and with a charging voltage o~ approximately 110 ~olts, this period is set, for example, a~ approximately 2 nanoseconds. One to 25 sections of coaxial cable having lengths of from 0.25"
to 300" may be employed, with appropriat~ variation in on-time.
Mono 106 is adjustable to set a switching time for its Q output to return high at a selected time, following it being a tri~gered as described. When it does, diode 108 would again be blocked and thu~ tha shorted condition on the base input o~ avalanch~ transistor 104 removed, enabling it to be sensitive ts an incoming signal. For example, this would occur at time T1 of waveform J. The period of delay before switching by mono 106 i8 set such that renewed sensitivity for avalanche amplifier 104 occurq at time psint T1, ju~t before it i9 anticipated that a signal of interest will occur. As will be noted, X
this will be just before the occurrenc0 of a signal pulse of waveform I. Thus, with a repetition rate of 25 KHz for the si~nal of interest, as described, mono 105 would be set to switch the Q output from low to high a~ter an essentially 40 microsecond, or 40,000 nanosecond, period.
Considerin~ that the width of the positive port.ion oP the input pulse is only about 20 nanoseconds, t'nus, durin~ most o~ the time, synchronous detector 102 i8 insensitive.
The window of sensitlvity is illu~trated as existing from time Tl to T2 and i5 tunable in duration by conventional timing adjustment of mono 106. Typically, it would be ~irst tuned fairly wide to provide a suEficient window for rapid locking onto a signal and then be tuned to provide a narrower w~ndow for a rnaximum compresslon ratio.
The output signal of avalanche transistor 104, wave~orm K, is a train of constant width pulses havin~ a leadin~
ed~e varyin~ as a ~unction of modulation. Thus, we have a form of pulse position modulat1on present. It appears across emitter-resistor 110, and it is fed from the emitter of transistor 104 to an act.ive type low pass filter 117.
Low pass filter 117 translates, demodulates, this thus varying pulse signal to a base band intell~gence signal, and this is fed to, and amplified by, audio amplifier 119.
~5 Then, assuming a voice transmissio~ a~ illustrated here, the output of audio amplifi~r ll9 is fed to and reproduced by loud speaker 12~. If the intelli~AncQ signal were otherwise, appropriate demodulation would be employ~d to detect the modulation present.
It is to be particularly noted that recelver 96 has two tuning features: sensitivity and window duration.
Sensitivity is adjusted by adjustment of variable volta~e source 112, and signal 'llock on" is effected by tuning of the period o~ high output state of mono 106 as described.
Typically, this period would be adjusted to the ~Z~
- 13 - .
minimum necessary to capture the range of excursion of the position modulated ~ignal pulses of lnterest.
Fig. 3 illustrates an alternate form of detector ~or receiver 96, it being designated detector 122. In it a form of synchronou~ siynal detlection i8 effected employing ring demodulator 124, formed of four matched diodes D1-D4.
In essence, it is operated as a sin~le pole, single throw swi tch, or simply a gate, ~with an input appearing across resistor 101 and applied to its input terminal I. Its gated output appears at terminal O and is ~ed through capacitor 113 and across resi~tor 115 to the lnput of demodulating, active type, low pas3 filter 117. Ring demodulator 124 i8 gated by a pulse PG illustrated in da hed lines in waveform L of Fig. 4 and applied across terminal ; 15 G. Pulse PG i8 generated by mono (monostable multivibrator) 126 as controlled by VC0 (voltage controlled oscillator) 121. VC0 127 is in turn controlled to effect synchronization with the average rate of the incoming signals shown in solid lines in waveform L. To accomplish this, the output voltage from ring demodulator 124 is fed through resistor 128 and across a (averaging) capacitor 130, connected to the con~rol input of VC0 12~. The thus controlled signal frequency output of VC0 127 is fed to the input of mono 126 which then provides as an output gating pulse PG. This pulse is rectangular as shown and having a selected pulse width, typically from 2 to 20 nanoseconds, being selected in ter~s of the time modulation of the transmitted pulse. It is fed to the primary winding of pulse transformer 132, and the secondary o~ this transformer i8 coupled across gate ter~inals G of rin~
demodulator 124. Diode 134 is connected acrocs the secondary of trans~or~er 132 and functions to effectively short out the ne~ative transition which would otherwis~
occur by virtue o~ the application of the pulse output of J\
7:L~
mono 126 to trans~ormer 132. In this manner, the gating pulse PG operates to bias all of the diodes of rlng demodulat4r 124 conductive for its duratlon ~nd thereby gating through the signal input ~rom terminal I to terminal 5 0. A~ stated above, this signal lnput is appli0d through capacitor 113 and across resistor 115 to the input o~ low pa~s filter 11~.
The function of detector 122 i5 to provide to low ~a~s filter 117 that portion of th~ input sig~al shown in waveform L of Fig. 4 appearing within the confines o~
gating pulse PG. The tinne position o~ ~ating pulse P~
is set by the timing of the pulse outputs o~ VC0 127, and the rate of the output of VC~ 127 is determined by the voltage input to VC0 127 as appearing across capacitor 130. Capacitor 130 i8 chosen to have a time constant which is just below that corresponding to the lowest frequency of modulation to be demodulated. Thus, the output pulse rate of VC0 12~ will be such as not to vary the pulse posltion of gating pulse PG during modulation ~0 induced time positions of the input signal (as shown in solid lines in wave~orm ~). As a result, the average value to the signal. This a~erage value is translated into an amplitu~e type int~lligence signal by passing it throuyh low pass filter 11~. It is then amplified, as desired, by audio amplifier 119 and then reproduced by loud speaker 120.
Fig. 3 illustrates an alternate embodiment o~ the receiv~r shown in Fig. 2. First, the antenna shown, bicone a~tenna 115, i~ employed as a directional a~tenna.
Second, a mixer 117 is ln the form of a double balance modulator, and it multiplies the ampli~i~d out-put o~
broadband ampli~ier 94 by a replica of the transmitted signal (Fi~. 4H~ generated by template generator 119 which may be an avalanche transistor. As will be noted, a monostable unit 126 is omitted, and the output of VC0 12~
31;Y~ L2 pr-ovide~ an injection voltage to mixer 117, Capacltor 129 and re~i~tor 131 function as a low pass fllter to control VC0 12~, which i5 an oscillator whlch c~n be varied by one or two cycles by voltage eontrol to e~fect a pha~e lock loop.
From the fore~oing, it should be appreciated that the applicant has provided a both inexpensive and practical time dornain syste~m o~ co~municat~ons. It employs the combination o~ an avalanche mode gated transistor 10 char~ed from a delay line; and when fed with a modulation induced variable! posltion pul~e, provi~es, as an output, a variable posi1:ion pul~e having a width of one to three nanoseconds. This in turn, of course, enables a large spectrum commencing at about 50 megacycles and 1~ extending downward to on the order of 500 megacycles.
Thus, with an audio frequence of, say, 5,000 ~z, the energy radiated to transmit this si~nal is d~spersed or spread an almost unbelievable 100,000 times. As a result, interferenoe with a conventional restricted 20 bandwidth signal is es~entially eliminated~ As an example of the effectiveness of such a -cy~tem, and employing 20-cent transistors in an avalanche ~ode, an audio modulated audio leading edge modulated pulse was provided as an output having a peak power of approximately 25 280 watts. The si~nal recelved at a distance o~ 200 feet had a peak vol~age of approximately 1 volt into a 50 ohm load. Actually, the power le~el necessary to receive has been found to be approximately a few micro-watts, thu~ the effective ranse with this power level is cons~derable. At 30 the same time, a spectrum analy~er at the reGeiving point failed to reveal any signal present or thus posslbility of ~ntere~erence with other services. ~ctually, in view of the distribution o~ the spectrum of the tran~mitted siynal, th~ level present which mi~ht interfere with a 35 standard signal, for example, a 5 K~z band width ~i~nal, ~t~
would be on the order of 2.8 micro-watts at the antenna.
One way of descrlbing the advantage that this type of transmission has over more conv2ntional ones is to note tha-t power appears in the example during an essentially 3-nanosecond period and appears only every 100,000 nanoseconds. Thus, it has a natural power ratio o~
33,000:1. Then, by limiting the listening period for that signal at essentially its pulse width, the receiving circuitry i8 only concern~ed with its appearance within a tiny window. Accordlngly, the overall signal-to-noise ratio i8 tremendous. It iLs to be ~urther appreciated that a vast number of users, employing slightly different repet.ttion rates, may be acco~modated and even this may be expanded by discrete patterns of pulse timing.
Either analo~ or digital patterns may be Qmployed which, for example, may ef~ect a dithering of the modulated pulse base, with a like or complementary dithering employed on the receivin~ end. In fact, with little degrees of sophistication, extremely con~idential communirations can be achieved even as against a receptor who has general knowledge of the presence of this type transmis3ion.
Beyond this, itR application to radar and motion detectors is essentially unllmited, enabling detection without delays typically re~uired ~or signal intQr~ration as often re~uired.
Fig. 5 illustrate~ a radar application of the pres~nt invention and particularly to a radar system, or that part of radar system, involved in determining range from transmitting antenna 200, a broadband, bicone antenna, to a like type receiving antenna 202. Of course, appropriate ~eans may also be employed to effect utilization o~ one antenna for both purposes. In general, signal bursts as illustrated in Fig. 4~ (disregardlng the tlme spacing which may vary as selected) is ~ssumed to be transmltted by bicone antenna ~00 having antenna elements X
204 and 206, and the output stage being in gen~r~l like that shown in Fig. la with switch 208 comprising the combination o~ the avalanche transistors 66 and 68 and pul~e transformer 60.
The transmitter is biasically controlled by control 210. It includes a transmit sequence co~trol portio~ 212 ~hich determines the timin~ of transmitted si~nal bursts, which sequence may be random, changing, or at a constant rate, say, for example, 10,000 bursts per second, in which case transmit sequence control 212 would generate and provide as .an output a 10,000 Hz pulse output on lead 214.
Oscillator 216 ~ operated at a higher rate, for example, 20 MHz, and the signal output of transmit se~uence control 212 i9 employed to select particular pul e outputs of oscillator 216 to be the actual pul~e which is used as a ma~ter pulse for controlling both the outpllt of transmitter 218 and the timing of receiver functions as will be further described. In ord~r to unambiguously and repetitively select an operative pulse with low timing uncertainty from oscillator 216, the ~election is on~ and some fraction o~ an oscillator pulse interval after an initial signal from control ~12. The selection is made via a control sequence employing D-type flip-flops 218, 220, and 222. Thus, the transmit se~uence control pulse on lead 214 is applied to the clock input of flip-~lop 218. This causes the Q output oP flip-flop 218 to transition to a high state, and this i~ applied to a D
input of ~lip-flop 220. Subsequently, the output of oscillator 216 impose6 a risin~ ed~e on the clock input of flip-flop 220. At that time, the high level of the D
input of this flip-flop i~ transferred to the Q output.
Similarly, the Q output of flip-flop 220 i~ provided to the D input of flip-~lop 222, and the next rising ed~e of the pulse from oscillator 216 will cause the not Q output to go low and th~3 inltiate the beginning of the transmit-receive cycle.
The not Q output of flip-flop 222 is fed to delay 224 which, in this embodiment, delays thi~ pulse by 200 nanoseconds, and the latter provides an output which triggers ~switch 228, cau~ing a transmitt~r burst by bicone antenna 200.
Receiv~r 226 receives echoes or returns via bicone antenna 202, and this output i8 amplified by ampllfier 228 and fed to mixer 230. Mi~r 230 include~ a double balanced modulator, and it functlons to multiply th~
signals instantaneously present, æay, a signal burat as illustrated by Fig. 4H by a signal which i~ polarity related and, for example, mlght be a signal like that o~
wave~orm 4H. In our illustration here, we are assuming that we will look at a single period in t.ime for a return : following a single output ~rom transmitter 218, and template generator 232 would generate a signal like, for example, waveform 4H and apply it to mi~er 230 at a precise time which would be a time which it is po~ible there would be a signal return for a target. In order to trigger template generator 232 to create a template like waveform 4H at an appropriate time, it is, of course, necessary to ef~ect a delay from a known time related to the transmission of a burst; and in thi~ instance, that signal information is on lead 234. In order to determine the precise time to ~e examined during a single cycle of operation of the system, encompassing one transmitted pulse, two pulse delay units are employed, course delay down counter 235 and fine programmable delay line 235. Down count~r 235 counts down the number of pulæe outputs from o~cillator 216 which occur subsequent to a control input to it on lead 238. The number of cuch pulses is pro~rammable in down counter 235 by an output ~ fro~
load count 241 on lead 240 of control 210, a conventional device wherein a binary count is generated X
in control 210 which i5 loaded into down counter 235. As an example, we will assume that it is desired to look at a return which occurs 1~5 nano~econds after the transmission of a signal ~rom antenna 200. To 5 accomplish this, we load imto down counter 235 the number "7," which means it will count seven of the pulse outputs of oscillator 216, each b0ing spa~ed at 50 nanoseovnds. At the same time, it is to be noted that delay 224 accomplishes a fixed delay of 200 nanoseconds.
So there is aohieved a 350-nanosecond delay in down counter 234, but subtracting 200 nanoseconds, we will have really an output of down counter 234 occurrin~ 150 nanosecond~ af~er the transmis~ion of a burst by transmitting antenna 200. In order to obtain the precise timing of 175 nanoseconds, an additlonal delay, or fine delay control, is effected by pro~rammable delay line 236, which is triggered by the output o~ down counter 234 when its seven count is concluded. It is programmed in a conventional manner hy load delay 2~2 of control 210 on lead Y and, thus in the example described, would have programmed programmable delay line 236 to delay an input pulse provided to it by 25 nanoseconds. In this manner, programmable delay llne 236 provides a pulse output to template generator 232, 175 nano6econds after it i ~ransmit~ed by bicon~ transmitting antenn~ 200. Template generator 232, which may be an avalanche transistor generating a waveform 4~, is then supplied to mixer 230 to be mixed, multiplied, with the amplified received output of receiving bicone antenna 202. The output of mixer 230 is fed to analGg integrator 250. A-~suming that there is identity in time between the identical waveforms~ a D.C.
signal level, for example, a positive signal value, will be provided ~8 an outpu~ o~ analog inte~rator 250.
This is ampli~ied by amplifier 252 and supplied to sample and hold unit 254. The output of sample and hold - 20 ~
unit 254 is fed to A-D converter 256, whlch then digltizes the summed values, ef~ecting this after a ~ixed delay of 40 microsecond3 provided by dPlay unit 258 which takes into account the processing time required by sample and hold unit 254. Where de~ired, a ~umber o~
transmisslons described would be effected in seque~ce, ~or example, 10, wherein the same signal transit tlme of reception would be observed, and any slgnals occurring during like ~ransmlssions would then be integrated in digital integrator 262, and in this way enable recovery of signals from ambient noise. The OlltpUt of digital integrator 2~2 would be displayed on display 264, synchronized ln time by an appropriate s.lgnal from delay line 236 (and delay 256) which would thus enable the time or distance position of a signal return to be displayed in terms o~ distance from the radar unit.
Fig,. 6 illustrates an application of applicant's radar to a surveillance operatio~ whieh might cover a radius of anywhere Prom 20 or 30 feet to several thousand Peet.
In this illustration, it is assumed that there is positioned at a selected central location a transmit bicone antenna, in this case a non-directional, or onmidirectional, antenna 300, and positioned at 120 degree points around it are receivsd bicone antennas 302, 304, and 306. Antenna 300 is powered by a transmitter 200 ~Fig. la). Assuming that a single signal burst is transmitted from transmit antenna 300, it would be radiated around 360 degrees and into space. At some selected time as discussed above, receivers 308, 310, and 311 would be supplied a template signal as described above to thu~ in effect cau3e the receivers to sample a signal echo being received at that precise instant. Thi~ process would be repeated for incrementally increasiny or decreasing times, and thus there would be stored in the memory's units 312, 314, and 316 si~nal~ representative o~ a range oP transit time~.
57~2 Then by selection o~ a ~omb~nation of transit t1mes for each of the receivers in terms oP triangularizations, lt i~
possible to select stored signals from the memory unit~
representative of a particular location in space. For surveillance purpose~, the result of signals derived from one scan and a la~er occurrin0 scan would be digitally subtracted, and thus where an object at some point within the range of the unit has moved to a new location, there will there be a difference in the scan information.
This thus would signal that something may have entered the area. This process ln general wottld be controlled by a read-write control 318 which would control the memory's units 312, 314, and 316 and would control a comparator 320 which would receive selected values X, Y, and Z ~rom memory units 312, 314, and 316 to make the subtraction.
Display 322, such as an oRcillo~cope, may be employed to display the relative position of an obJect change with respect to a radar location.
Fig. 7 illustrates an application of applicant's invention to a radar system wAerein there is one transmitting antenna located in a discrete planar posi~ion with respect to the direction of observation, three receiving antennas spaced in a plane parallel to the first plane, and a fourth receiving antenna positioned in a third plane. Thus, radiation from transmitting antenna 404, which is re~lected by a target, i5 received by the four receiving antennas at varying times by virtue of the di~ference in path length. Because of the unique characteristic of applicant's system in that lt can be emp~loyed to resolve literally inches, extreme detail can be resolved from the returns. Referring to Fig. 7, control 400 directs a tran mission by transmitter 402 which supplies a signal burst to tran~mitting antenna 404.
Signal returns are received by antenna3 406, 408, and 410 ~5 located, for example, in a plane generally normal to the X
direction of view and separate from the plane in which transmit antenna 404 is located. A fourth receiving antenna 412 i5 located in still a third plane which is normal to the direction of view and thus in a plane separated from the plane in which th~ other receiving antennas are located. By virtue of this, there is provided means for locating, via triangularixation, a target in space, and thus there is derived sufficient signal information to enable three-dimensional information di~plays. The received signals ~rom receivers 412, 414, 416, and 418 are separately supplied to signal processor and comparator 420, which includes a memory f~r storing all sample~ received and in term~ of their time of receipt.
From this data, one can compute po~ition information by an appropriate comparison as well as target characteri~tics, Ruch as size and reflectivity.
TIME DOMhIN RADIO TRANSMISSION SY~TEM
Field of the Invent.ion The invention relates generally to radlo transmission systems, and particularly to a time domain syst~m wherein spaced sin~le cycles, or near single cycles sometimes referred to a~ monocycles, of electromagnetic energy are transmitted in ~pace and where discrete frequency signal components are generally below noise level and are thus not discernabl0 by conventional radio receiving equipment.
Ba k~ound of the Inven_ion The radio transmission of communlcations signals, for example, audio signals, i5 normally effected by one of two methods. In one, referred to a3 an amplitude modulation sy~tem, a sinusoidal radio frequency carrier is modulated in amplitude in terms of the intelligence or communications signal, and when the signal i~ received ~t a receiving location, the reverse process, that i5, demodulation of the carrier, is ef~ected to recover the communications signal. The other ~ystem employs what is termed frequency modulation, and ln.stead oP amplitude modulation of the carrier signal, i t is frequency modulated. When an FM frequency modulation or FM signal i8 received, circuitry is employed which performs what is termed discrimination wherein chan~es in freguency are changed to changes in amplitude in accordance with the original modulation, and thereby a communlcation si~nal i~ recovered. In both systems, there is as a basis a sinusoidal carrier which is assigned and occupies a distinctive frequency band width, or channel, and this channel occupies ~pectrum space which cannot be utilized by other transmissions within the range of its employment.
At this time, almost every nook and cranny of spectrum space X
.
is being utilized, and there lg a tremendous need for some method of expanding the availability of communications channels. In consideration of thi~, it has been suggested that instead of the use o~ di~crete frequency channels for radio communications links, which is the con~entional approach, a radio tran3mission link employing a wider frequency spectrum which ~ay extend over a range of lO to 100 times the intelligence band width being transmitted, but wherein the energy o~ any single fre~uency making up that spectrum be very low, typlcally below normal noise level3. Thus, it would be obvious that thl~ type o~
transmission would be essentially non interferin~ with other services.
Additionally, and as well expre~sed in an article entitled "Time Domain Electromagnetics and Its Application,"
Proceeding~ of the IEEE, Vol. 6~, No. 3, (March 19~8), it has been suggested that baseband ~ignals generated ~rom pulses o~ a short duration, e.g., in the pico second range employed for such applications ~s basehand radar.
Ranges on the order of 5 to 5,000 feet were sugg~sted.
This article appeared in 1978, and recent discussions with radar engineers as to subsequent development o~ baseband radar su~gest that little has been achie~ed, p~rticularly in the development of the widely use medium range ~ield of up to 10 kilometers. The reasons given for the lack of success in this area appear to be si~nal reception ~rom such systems cannot adequately combat noi~e. It i3 to be kept in mind that the signal-to-noise ratio problem is made enormous by the fact that the baseband radar signal received must compete with all of the electromagnetic noi~e occurring within the entire spectrum of the band of the radar signal, which i5, ~or example, from 100 MHz to 1.5 GHz, or higher. E~en where there is no intentional jamming of energy present, there is an enormous amount of electromagnetic energy present in X
addition to the poor r~dar ~ignal at the input of a B~R
BAseband Radar. On it5 face, the problem seems rather hopeless, and it is believed that this is about where the problem lies.
In accordance with this invention, a pulse slgnal of a fixed or programmed rate i5 generated, and it is varied or modulated as to tAe time of occurrence when it is eraployed as a basis of an intelligence signal. The re~ultant pulse slgnals affect the turn--on or turn-off o~ a fast electronic switch ~uch as an avalanche mode o~erated semiconductor or ~park cap which turns on or of~ power input to a broadband transmission system. The resultant output, a carrierles~ signal burst, i5 coupled to the atmosphere or space and thus transmission. Reception of the transmission is effected by a receiver which is timed selectively to effect detection e~ploying an analog multiplier which multiplies the received signal by a locally ger.erated signal which bears a polarity-time relationship to the transmitted ~ignal. The multiplication produces a correlation signal which rather uniquely corresponds to the actual transmitted signal where the target is normal and planar to the direction of transmission and reception. Thus, even in the presence of an essentially randomly changing voltage from noise, upon which the transmitted signal i8 typically ridin~, it has beeh found that si~nal recognition over the noise can be achieved despite substantially higher levels of noise than the level of the radar signal. Essentially what happens i8 that in the absence of a received ~ignal which is not closely in phase from a polarity point of view with the internally senerated signal, the output, particularly after inte~rative processing~ will be vastly lower in level than with a correlated si~nal present.
To enhance reception even further, the output of the mixer, or correlator, i~ sampled at a number o~ polnts X
%
over .its effective length or duration, and it then becomes pos~ible to relate this serie3 o~ points with a series of anticipated values and a decision a~ to whether there is a target present can be made in terms of a degree of resemblance between a sltandard and the waveform bit pattern of the received signal. This process thus is a two-stage one, wherein one would choose a known template, representative, at least time-polarit~wise, of a particular type of target for th~ in~ection signal.
There is then provided éa detected sign~l which will provide target identication with an extremely high probability just from its level. Then, beyond this, by virtue of discrete samplings a~ ~elected time points and comparison with what would be expected if a targct were present, a still further element of pasitiveness in resolution i5 enabled. All in all, it is bslieved that the present invention solves the grea test inhibition to the satisfactory development of medium and, in fact, long distance radar.
Brief Descr~ on of the Drawin~s Fig. 1 is a combination block-~chematic diagram of a time domain transmitter.
Fi~. la is a schematic diagram oP an alternate form of output stage for the transmitter shown in Fig. 1.
Fig. 2 is a combination block-schematic diagram of a time domain receiver as contemplated by this invention.
~ig. 2a is a combined block-schematic electrical diagram of an alternate form of synchronous detector to the one shown in Fig. 2.
Fig. 3 i~ an electrical block diagram of an alternate embodiment of a time domain receiver.
Fig. 4 is a set of electrical wa~eforms illustr~tive of aspects of the circuitry shown in Fi~s. 1 X
.~2~57~Z
and 2.
Fig. 5 is an electrical block diagram of a time domain r~dar system.
Fig. 6 ls a schematic illustration of a surveillance system as contemplated by the present lnvention.
Fig. 7 is a schematilc illustration of a phase~ array radar system as contemplated by this invention.
Description of _he Preferred _mbodiment Referring to F~g. 1, and initi~lly to transmitter 10, a base frequency of 100 KHz is generated by o~cillator 12, typically being a crystal controlled oscillator which includes conventional circuitry ~or providing a~ an output square wave pulse~ at 100 KHz rate. Thi8 pulse signal is applied to divide-by-4 divider 14 to provide at its output a square wave 25 KHz, 0-5 volt, signal shown in wave~orm A of Fig. 4. Further references to waveforms will simply identify them by their letter identity and will not further refer to the figure, which is Fig. 4 in all cas~s. This output is employed as a general transmission signal and as 20 an input to power supply 16. The latter is regulated, one which supplies a 300-volt D.C. bias on a non-inter~ering basis for the output stage 18 of transmitter 10, whlch is also keyed at the 25 KHz rate.
The output of divide-by-4 divider 14 is employed as a signal base and as such is supplied through capacitor to pulse po~ition modulator ~2. Pulse position modulator 22 includes in its input an RC circuit consi3ting of resi~tor 24 and capacitor 26 which convert the ~uare wave input to an approximately trian~ular wave as 3hown in waveform B, it being applied across re~istor 25 to the non-inverting input of comparator 28. A sslected or reference positive voltage, ~iltered by capacitor 27, is also applied to the non~invertins input o~ comparator 28, it being supplied from ~5 volt terminal 29 of D.C. bias ~ 3~
supply 30 through resistor 32. Accordingly, Por example, there would actually appear at the non-inverting input a triangular wave biased upward positively as illustrated by waveform C.
The actual cGnduction level of comparator 28 is deter~ined by an audio signal input ~rom microphone 34 supplied through capacitor 36, across resistor 37, to the invert~ng input of comparator 28, as biased from supply 30 throu~h resistor 38 and across resistor 32.
The combined audio signal and bia~ i~ illu~trated in waveform D. By virtue of the thu~ described input combination, the output of comparator 28 would ri~e to a positive saturation level when triangular wave slgnal 40 (waveform E~ i8 of a higher value than modulat~on signal 42 a~d drop to a negatlve saturation level when modulation signal 42 i8 of a greater value than the trl~ngular wave signal 40. The output signal of comparator 28 is shown in waveform F.
In the present case, we are interested in employing the negative going or trailing edge 44 (waveform F) of the output o~ comparator 28, and it is to be noted that this trailing edge will vary in its time position as a function of the signal modulation. This trailing edge of the waveform in wave~orm F triggers "on" mono, or monostable multivibrator, 46 havin~ an "on" time of approximately 50 nanoseconds, and its output is shown in waveform ~. For purposes of illustration, ~hile the pertinent leading or trailing edges of related waveforms are properly aligned, pulse widths and spacings (as indicated by break lines, spacings are 40 microseconds) are not related in scale.
Thus, the leading edge of pulse waveform G corresponds in time to the trailing edge 44 ~wave~orm FJ and its time position within an avera~e time between pulses o~ waveform G is varied as a function of the input audio modulation ~ignal to ~omparator 28.
X
~f~?',~7~ 2 The output of mono 46 i~ applied through dlode 48 across resistor 50 to the base input of NPN transistor S2 operated as a triggering ampli~ier. It i8 conventionally biased through resistor 54, e.g., 1.5K
ohms, ~rom +5 volt terminal 29 of 5 volt power supply 30 to its collector. Capacitor 56 havlng an approxlmate capacitance of 0.1 mf is connected between the collector and ground of transistor 52 to enable ~ull bias potential to appear across the translstor for lts brief turn-on interval, 60 nanoseconds. The output o~
transistor 52 is coupled between its emitter and 0round to the primary 58 of trigg~r transformer 60.
Additionally, transistor 52 may drive trans~ormer 60 via an avalanche transistor connected in a rommon emitter con~iguration via a collector load resistor. In order to drive transformer 60 with a steep wave ~ront, the avalanche mode operated transistor i8 ideal. Like secondary windings 62 and 64 of trigger transformer 60 separately supply base-emitt~r inputs of NPN avalanche, or avalanche mode operated, transistors 66 and 63 o~ power output sta~e 18. Although two are shown, one or more than two may be employed when approprlately couple~.
Avalanche mode operated transistors 66 and 68, many type 2N2222 with a met21 can, have the characteristic that when they are triggered "on," their resistance goes low (e.gO, approximately 30 ohms for each) and st~y~ at this state until collector current drops sufficiently to cut o~ conduction (at a few microamperes~. Certain other transistors, such as a type ~N440l, also display reliable avalanche characteristics. Their collector-emitter circuits are connected in series, and collector bias of ~300 volts is applied to them from power supply 16, across filter capacitor 72, and throu~h resistor 74 to one end 76 of parallel connected delay lines DL. While three sections S1-53 are shown, typically five to ten would be X
5~
employed. They may be constructed of type R558 coaxial cable, and each being approximately three inche~ in length as required to totally effect an approximately 3 nanosecond pulse. ~5 shown, the positive input potential ~rom resistor 74 is connected to the center conductor of each of the delay lines, and the outer conductors are connected to ground.
Re~istor ~4 is on the order of 50R ohms and is ad~usted to allow a current flow through transistors ~6 and 68 o~
about 0.2 MA which i~ a zener current which places both transistors in a near sel~-triggerin~ state. It has been found that under this condition, the transistors will self-ad~u~t to an av;alanche voltage which may be different for the two. Normally, resistor 74 will still be of value which will enable charging of the delay lines ~ between pulses. Delay lines DL ar~ charged to 300 volts bias durin~ the period when transistors 66 and 68 are turned o~f, between input pul~es. When the inputs to transi~tors 66 and 68 are triggered "on" by a triggering pulse they begin to conduct within 0.5 nanoseconds, and by virtue of the low vultage drop across them (when operated in an avalanche mode as they are), about 120 volts appears as a pul5e across output resistor 78, e.g., 50 ohms.
Significantly, the turn-on or leading edge of this pulse is effeoted by the trigger pulse applied to the inputs of transi~tors 66 and 68, and the trailing edge of this output pulse is determined by the dischar~e time of delay llnes DL. By this technique, and by choice of length and Q of the delay lines, a well-shaped, very short pul~e, on the order of 3 nanoseconds and with a peak power o~ appro~imately 300 watts, i5 generated.
Following turn-o~f, delay lines DL are recharged through resistor 74 before the arrival of the next triggering pulse. As will be apparent, pow~r stage 18 is extremely simple and is constructed of quite inexpensive circuit elements. For example, transistors 66 and 68 are X
,5~
available at a cost of approximately $0.1Z.
The output of power output sta~e 18 appears across resistor 78 and is supplied through coaxial cable 80 to a time domain shaping fi:Lter 82 which would be employed to affix a selected signature to the output a9 a form o~ encoding or recognitton signal. Alternately, filter 82 may be omitted where such security ~easures are not deemed necessary; and, el9 indicative o~ thi~, a bypa~s line 84 including a switch 86 diagrammatically illu~trates such omission.
The signal outpu't of filter 82, or directly the output o~ power ~tage 1~, i8 supplied through coaxial cable 88 to discone antenna 90, which is an aresonant antenna. This type of antenna relatively uniformly 16 radiates all ~lgnals of a frequency above its cut-off frequency, which is a function of size, for example, signals above approximately 50 MHz for a relatively small unit. In any event, antenna 90 radlate~ a wide spectrum signal, an example being shown ln the time domain in waveform ~ of Fig. ~, ess~ntially a mono~ycle, this waveform being the composite of the shaping effect~ of filter 82, i~ used, and, to an extent, discone antenna 90.
Fig. la illustrates an alternate and simplified output s~age. As illustrated, biconical antenna 200, a~ a broadband antenna, is charged by a D.C. source 65 through resistors 6~ and 69 to an overall voltag~ which is the sum of the avalanche voltage o~ transistors 66 and 68 as discu-~sed above. Resistors 67 and 69 together have a resistance value which will enable transistors 66 and 68 to be biased as descri~ed above. Resistors ~1 and 73 are of relatively low value and are adjusted to receive energy below the frequency of cut-off o~ the antenna and also to prevent ringing. In operation, when a pul~e is applied to the primary 58 of pulse transformer 60, transi~tors ~6 and 68 are turned on, ef~ectively shorting, through X
t~
resistors 71 and ~3, biconical cmtenna elementæ 20~ and 206. This action occurs essentially at the speed of light, with the result that a signal, es~entially a monocycle or about one and a half cycles a~ Rhown in Fig.
4H, is transmitted as described above for the transmitter output system shown in Fig. 1.
The output o~ disc:one antenna 90, or blcone antenna, is typically transl~itted over a di3crete space and would typically be received by a like discone antenna g2 of receiver 96 at a second location. Although transmission e~fects may distort the waveform some, for purpo~es of illustration, it will be as3umed that the wave~orm received will be a replica o~ wave~orm H. The received signal is amplified by broad band ampli~ier 94, having a broad band frequency response over the range of the transmitted signal. In instances where a filter 82 is employed in transmitter 10, a reciprocally con~iyured filter 98 would be employed. As illustrative of instances where no matched filter would be employed, there i5 diagrammatically illustrated a switch 100 connecting the input and output of filter 98, denoting that by closing it, filter 98 would be bypassed. Assumlng that no match fil~er is employed, the output of broad band ampli~ier as an amplified replica of waveform H is illustrated in wa~e~orm I. In either case, it appears across re~istor 101 .
5ignal waveform I iY~ applied to synchronous detector 102. Basically, it has two functional UAi ts, avalanche transistor 104 and adjustable mono 106. Mono lQ6 is driven from an input acro~s emitter-resi~tor llO, connected between the em~tter of avalanche transistor 104 and ground. Avalanche transistor 104 i5 biased from variable voltag~ D.C. source 112, e.g., 100 to 130 volts, through variable resistor 114, e.g., lOOK to lM ohms. A delay line 116 is connected between the collector and ~round of X
%
transistor 104 and provides the effe!ctive operatingbla~ for translstor 104, it being char~ed between conduction period~
as will be described.
A~suming now that a chargin~ interval ha~ occurred, avalanche transistor 104 will be turned on, or trig~ered, by a si~nal applied to its base ~rom across resistor 101. It will be further assumed that this triggering is enabled by the Q output, waveform J, o~ mono 106 bein~ hi~h. Upon being trig~ered, the conduction of avalanche transistor 104 will produce a rising volta~e across emitter re~istor 110, waveform K, ~nd this volta~e ~ill in turn trig~er mono 106 to cause it~ Q output to go low. This in turn causes diode 108 to conduct and thus effectively shorting out the input to avalanche transistor 104, this occurring within 2 to 20 nanoseconds from the positive leadin~ edge of the input ~ignal, waveform I.
The conduction period of transi~tor 104 i8 precisely set by the charge capacity of delay line 116. With a delay line formed of 12" of R~58 coaxial cable, and with a charging voltage o~ approximately 110 ~olts, this period is set, for example, a~ approximately 2 nanoseconds. One to 25 sections of coaxial cable having lengths of from 0.25"
to 300" may be employed, with appropriat~ variation in on-time.
Mono 106 is adjustable to set a switching time for its Q output to return high at a selected time, following it being a tri~gered as described. When it does, diode 108 would again be blocked and thu~ tha shorted condition on the base input o~ avalanch~ transistor 104 removed, enabling it to be sensitive ts an incoming signal. For example, this would occur at time T1 of waveform J. The period of delay before switching by mono 106 i8 set such that renewed sensitivity for avalanche amplifier 104 occurq at time psint T1, ju~t before it i9 anticipated that a signal of interest will occur. As will be noted, X
this will be just before the occurrenc0 of a signal pulse of waveform I. Thus, with a repetition rate of 25 KHz for the si~nal of interest, as described, mono 105 would be set to switch the Q output from low to high a~ter an essentially 40 microsecond, or 40,000 nanosecond, period.
Considerin~ that the width of the positive port.ion oP the input pulse is only about 20 nanoseconds, t'nus, durin~ most o~ the time, synchronous detector 102 i8 insensitive.
The window of sensitlvity is illu~trated as existing from time Tl to T2 and i5 tunable in duration by conventional timing adjustment of mono 106. Typically, it would be ~irst tuned fairly wide to provide a suEficient window for rapid locking onto a signal and then be tuned to provide a narrower w~ndow for a rnaximum compresslon ratio.
The output signal of avalanche transistor 104, wave~orm K, is a train of constant width pulses havin~ a leadin~
ed~e varyin~ as a ~unction of modulation. Thus, we have a form of pulse position modulat1on present. It appears across emitter-resistor 110, and it is fed from the emitter of transistor 104 to an act.ive type low pass filter 117.
Low pass filter 117 translates, demodulates, this thus varying pulse signal to a base band intell~gence signal, and this is fed to, and amplified by, audio amplifier 119.
~5 Then, assuming a voice transmissio~ a~ illustrated here, the output of audio amplifi~r ll9 is fed to and reproduced by loud speaker 12~. If the intelli~AncQ signal were otherwise, appropriate demodulation would be employ~d to detect the modulation present.
It is to be particularly noted that recelver 96 has two tuning features: sensitivity and window duration.
Sensitivity is adjusted by adjustment of variable volta~e source 112, and signal 'llock on" is effected by tuning of the period o~ high output state of mono 106 as described.
Typically, this period would be adjusted to the ~Z~
- 13 - .
minimum necessary to capture the range of excursion of the position modulated ~ignal pulses of lnterest.
Fig. 3 illustrates an alternate form of detector ~or receiver 96, it being designated detector 122. In it a form of synchronou~ siynal detlection i8 effected employing ring demodulator 124, formed of four matched diodes D1-D4.
In essence, it is operated as a sin~le pole, single throw swi tch, or simply a gate, ~with an input appearing across resistor 101 and applied to its input terminal I. Its gated output appears at terminal O and is ~ed through capacitor 113 and across resi~tor 115 to the lnput of demodulating, active type, low pas3 filter 117. Ring demodulator 124 i8 gated by a pulse PG illustrated in da hed lines in waveform L of Fig. 4 and applied across terminal ; 15 G. Pulse PG i8 generated by mono (monostable multivibrator) 126 as controlled by VC0 (voltage controlled oscillator) 121. VC0 127 is in turn controlled to effect synchronization with the average rate of the incoming signals shown in solid lines in waveform L. To accomplish this, the output voltage from ring demodulator 124 is fed through resistor 128 and across a (averaging) capacitor 130, connected to the con~rol input of VC0 12~. The thus controlled signal frequency output of VC0 127 is fed to the input of mono 126 which then provides as an output gating pulse PG. This pulse is rectangular as shown and having a selected pulse width, typically from 2 to 20 nanoseconds, being selected in ter~s of the time modulation of the transmitted pulse. It is fed to the primary winding of pulse transformer 132, and the secondary o~ this transformer i8 coupled across gate ter~inals G of rin~
demodulator 124. Diode 134 is connected acrocs the secondary of trans~or~er 132 and functions to effectively short out the ne~ative transition which would otherwis~
occur by virtue o~ the application of the pulse output of J\
7:L~
mono 126 to trans~ormer 132. In this manner, the gating pulse PG operates to bias all of the diodes of rlng demodulat4r 124 conductive for its duratlon ~nd thereby gating through the signal input ~rom terminal I to terminal 5 0. A~ stated above, this signal lnput is appli0d through capacitor 113 and across resistor 115 to the input o~ low pa~s filter 11~.
The function of detector 122 i5 to provide to low ~a~s filter 117 that portion of th~ input sig~al shown in waveform L of Fig. 4 appearing within the confines o~
gating pulse PG. The tinne position o~ ~ating pulse P~
is set by the timing of the pulse outputs o~ VC0 127, and the rate of the output of VC~ 127 is determined by the voltage input to VC0 127 as appearing across capacitor 130. Capacitor 130 i8 chosen to have a time constant which is just below that corresponding to the lowest frequency of modulation to be demodulated. Thus, the output pulse rate of VC0 12~ will be such as not to vary the pulse posltion of gating pulse PG during modulation ~0 induced time positions of the input signal (as shown in solid lines in wave~orm ~). As a result, the average value to the signal. This a~erage value is translated into an amplitu~e type int~lligence signal by passing it throuyh low pass filter 11~. It is then amplified, as desired, by audio amplifier 119 and then reproduced by loud speaker 120.
Fig. 3 illustrates an alternate embodiment o~ the receiv~r shown in Fig. 2. First, the antenna shown, bicone a~tenna 115, i~ employed as a directional a~tenna.
Second, a mixer 117 is ln the form of a double balance modulator, and it multiplies the ampli~i~d out-put o~
broadband ampli~ier 94 by a replica of the transmitted signal (Fi~. 4H~ generated by template generator 119 which may be an avalanche transistor. As will be noted, a monostable unit 126 is omitted, and the output of VC0 12~
31;Y~ L2 pr-ovide~ an injection voltage to mixer 117, Capacltor 129 and re~i~tor 131 function as a low pass fllter to control VC0 12~, which i5 an oscillator whlch c~n be varied by one or two cycles by voltage eontrol to e~fect a pha~e lock loop.
From the fore~oing, it should be appreciated that the applicant has provided a both inexpensive and practical time dornain syste~m o~ co~municat~ons. It employs the combination o~ an avalanche mode gated transistor 10 char~ed from a delay line; and when fed with a modulation induced variable! posltion pul~e, provi~es, as an output, a variable posi1:ion pul~e having a width of one to three nanoseconds. This in turn, of course, enables a large spectrum commencing at about 50 megacycles and 1~ extending downward to on the order of 500 megacycles.
Thus, with an audio frequence of, say, 5,000 ~z, the energy radiated to transmit this si~nal is d~spersed or spread an almost unbelievable 100,000 times. As a result, interferenoe with a conventional restricted 20 bandwidth signal is es~entially eliminated~ As an example of the effectiveness of such a -cy~tem, and employing 20-cent transistors in an avalanche ~ode, an audio modulated audio leading edge modulated pulse was provided as an output having a peak power of approximately 25 280 watts. The si~nal recelved at a distance o~ 200 feet had a peak vol~age of approximately 1 volt into a 50 ohm load. Actually, the power le~el necessary to receive has been found to be approximately a few micro-watts, thu~ the effective ranse with this power level is cons~derable. At 30 the same time, a spectrum analy~er at the reGeiving point failed to reveal any signal present or thus posslbility of ~ntere~erence with other services. ~ctually, in view of the distribution o~ the spectrum of the tran~mitted siynal, th~ level present which mi~ht interfere with a 35 standard signal, for example, a 5 K~z band width ~i~nal, ~t~
would be on the order of 2.8 micro-watts at the antenna.
One way of descrlbing the advantage that this type of transmission has over more conv2ntional ones is to note tha-t power appears in the example during an essentially 3-nanosecond period and appears only every 100,000 nanoseconds. Thus, it has a natural power ratio o~
33,000:1. Then, by limiting the listening period for that signal at essentially its pulse width, the receiving circuitry i8 only concern~ed with its appearance within a tiny window. Accordlngly, the overall signal-to-noise ratio i8 tremendous. It iLs to be ~urther appreciated that a vast number of users, employing slightly different repet.ttion rates, may be acco~modated and even this may be expanded by discrete patterns of pulse timing.
Either analo~ or digital patterns may be Qmployed which, for example, may ef~ect a dithering of the modulated pulse base, with a like or complementary dithering employed on the receivin~ end. In fact, with little degrees of sophistication, extremely con~idential communirations can be achieved even as against a receptor who has general knowledge of the presence of this type transmis3ion.
Beyond this, itR application to radar and motion detectors is essentially unllmited, enabling detection without delays typically re~uired ~or signal intQr~ration as often re~uired.
Fig. 5 illustrate~ a radar application of the pres~nt invention and particularly to a radar system, or that part of radar system, involved in determining range from transmitting antenna 200, a broadband, bicone antenna, to a like type receiving antenna 202. Of course, appropriate ~eans may also be employed to effect utilization o~ one antenna for both purposes. In general, signal bursts as illustrated in Fig. 4~ (disregardlng the tlme spacing which may vary as selected) is ~ssumed to be transmltted by bicone antenna ~00 having antenna elements X
204 and 206, and the output stage being in gen~r~l like that shown in Fig. la with switch 208 comprising the combination o~ the avalanche transistors 66 and 68 and pul~e transformer 60.
The transmitter is biasically controlled by control 210. It includes a transmit sequence co~trol portio~ 212 ~hich determines the timin~ of transmitted si~nal bursts, which sequence may be random, changing, or at a constant rate, say, for example, 10,000 bursts per second, in which case transmit sequence control 212 would generate and provide as .an output a 10,000 Hz pulse output on lead 214.
Oscillator 216 ~ operated at a higher rate, for example, 20 MHz, and the signal output of transmit se~uence control 212 i9 employed to select particular pul e outputs of oscillator 216 to be the actual pul~e which is used as a ma~ter pulse for controlling both the outpllt of transmitter 218 and the timing of receiver functions as will be further described. In ord~r to unambiguously and repetitively select an operative pulse with low timing uncertainty from oscillator 216, the ~election is on~ and some fraction o~ an oscillator pulse interval after an initial signal from control ~12. The selection is made via a control sequence employing D-type flip-flops 218, 220, and 222. Thus, the transmit se~uence control pulse on lead 214 is applied to the clock input of flip-~lop 218. This causes the Q output oP flip-flop 218 to transition to a high state, and this i~ applied to a D
input of ~lip-flop 220. Subsequently, the output of oscillator 216 impose6 a risin~ ed~e on the clock input of flip-flop 220. At that time, the high level of the D
input of this flip-flop i~ transferred to the Q output.
Similarly, the Q output of flip-flop 220 i~ provided to the D input of flip-~lop 222, and the next rising ed~e of the pulse from oscillator 216 will cause the not Q output to go low and th~3 inltiate the beginning of the transmit-receive cycle.
The not Q output of flip-flop 222 is fed to delay 224 which, in this embodiment, delays thi~ pulse by 200 nanoseconds, and the latter provides an output which triggers ~switch 228, cau~ing a transmitt~r burst by bicone antenna 200.
Receiv~r 226 receives echoes or returns via bicone antenna 202, and this output i8 amplified by ampllfier 228 and fed to mixer 230. Mi~r 230 include~ a double balanced modulator, and it functlons to multiply th~
signals instantaneously present, æay, a signal burat as illustrated by Fig. 4H by a signal which i~ polarity related and, for example, mlght be a signal like that o~
wave~orm 4H. In our illustration here, we are assuming that we will look at a single period in t.ime for a return : following a single output ~rom transmitter 218, and template generator 232 would generate a signal like, for example, waveform 4H and apply it to mi~er 230 at a precise time which would be a time which it is po~ible there would be a signal return for a target. In order to trigger template generator 232 to create a template like waveform 4H at an appropriate time, it is, of course, necessary to ef~ect a delay from a known time related to the transmission of a burst; and in thi~ instance, that signal information is on lead 234. In order to determine the precise time to ~e examined during a single cycle of operation of the system, encompassing one transmitted pulse, two pulse delay units are employed, course delay down counter 235 and fine programmable delay line 235. Down count~r 235 counts down the number of pulæe outputs from o~cillator 216 which occur subsequent to a control input to it on lead 238. The number of cuch pulses is pro~rammable in down counter 235 by an output ~ fro~
load count 241 on lead 240 of control 210, a conventional device wherein a binary count is generated X
in control 210 which i5 loaded into down counter 235. As an example, we will assume that it is desired to look at a return which occurs 1~5 nano~econds after the transmission of a signal ~rom antenna 200. To 5 accomplish this, we load imto down counter 235 the number "7," which means it will count seven of the pulse outputs of oscillator 216, each b0ing spa~ed at 50 nanoseovnds. At the same time, it is to be noted that delay 224 accomplishes a fixed delay of 200 nanoseconds.
So there is aohieved a 350-nanosecond delay in down counter 234, but subtracting 200 nanoseconds, we will have really an output of down counter 234 occurrin~ 150 nanosecond~ af~er the transmis~ion of a burst by transmitting antenna 200. In order to obtain the precise timing of 175 nanoseconds, an additlonal delay, or fine delay control, is effected by pro~rammable delay line 236, which is triggered by the output o~ down counter 234 when its seven count is concluded. It is programmed in a conventional manner hy load delay 2~2 of control 210 on lead Y and, thus in the example described, would have programmed programmable delay line 236 to delay an input pulse provided to it by 25 nanoseconds. In this manner, programmable delay llne 236 provides a pulse output to template generator 232, 175 nano6econds after it i ~ransmit~ed by bicon~ transmitting antenn~ 200. Template generator 232, which may be an avalanche transistor generating a waveform 4~, is then supplied to mixer 230 to be mixed, multiplied, with the amplified received output of receiving bicone antenna 202. The output of mixer 230 is fed to analGg integrator 250. A-~suming that there is identity in time between the identical waveforms~ a D.C.
signal level, for example, a positive signal value, will be provided ~8 an outpu~ o~ analog inte~rator 250.
This is ampli~ied by amplifier 252 and supplied to sample and hold unit 254. The output of sample and hold - 20 ~
unit 254 is fed to A-D converter 256, whlch then digltizes the summed values, ef~ecting this after a ~ixed delay of 40 microsecond3 provided by dPlay unit 258 which takes into account the processing time required by sample and hold unit 254. Where de~ired, a ~umber o~
transmisslons described would be effected in seque~ce, ~or example, 10, wherein the same signal transit tlme of reception would be observed, and any slgnals occurring during like ~ransmlssions would then be integrated in digital integrator 262, and in this way enable recovery of signals from ambient noise. The OlltpUt of digital integrator 2~2 would be displayed on display 264, synchronized ln time by an appropriate s.lgnal from delay line 236 (and delay 256) which would thus enable the time or distance position of a signal return to be displayed in terms o~ distance from the radar unit.
Fig,. 6 illustrates an application of applicant's radar to a surveillance operatio~ whieh might cover a radius of anywhere Prom 20 or 30 feet to several thousand Peet.
In this illustration, it is assumed that there is positioned at a selected central location a transmit bicone antenna, in this case a non-directional, or onmidirectional, antenna 300, and positioned at 120 degree points around it are receivsd bicone antennas 302, 304, and 306. Antenna 300 is powered by a transmitter 200 ~Fig. la). Assuming that a single signal burst is transmitted from transmit antenna 300, it would be radiated around 360 degrees and into space. At some selected time as discussed above, receivers 308, 310, and 311 would be supplied a template signal as described above to thu~ in effect cau3e the receivers to sample a signal echo being received at that precise instant. Thi~ process would be repeated for incrementally increasiny or decreasing times, and thus there would be stored in the memory's units 312, 314, and 316 si~nal~ representative o~ a range oP transit time~.
57~2 Then by selection o~ a ~omb~nation of transit t1mes for each of the receivers in terms oP triangularizations, lt i~
possible to select stored signals from the memory unit~
representative of a particular location in space. For surveillance purpose~, the result of signals derived from one scan and a la~er occurrin0 scan would be digitally subtracted, and thus where an object at some point within the range of the unit has moved to a new location, there will there be a difference in the scan information.
This thus would signal that something may have entered the area. This process ln general wottld be controlled by a read-write control 318 which would control the memory's units 312, 314, and 316 and would control a comparator 320 which would receive selected values X, Y, and Z ~rom memory units 312, 314, and 316 to make the subtraction.
Display 322, such as an oRcillo~cope, may be employed to display the relative position of an obJect change with respect to a radar location.
Fig. 7 illustrates an application of applicant's invention to a radar system wAerein there is one transmitting antenna located in a discrete planar posi~ion with respect to the direction of observation, three receiving antennas spaced in a plane parallel to the first plane, and a fourth receiving antenna positioned in a third plane. Thus, radiation from transmitting antenna 404, which is re~lected by a target, i5 received by the four receiving antennas at varying times by virtue of the di~ference in path length. Because of the unique characteristic of applicant's system in that lt can be emp~loyed to resolve literally inches, extreme detail can be resolved from the returns. Referring to Fig. 7, control 400 directs a tran mission by transmitter 402 which supplies a signal burst to tran~mitting antenna 404.
Signal returns are received by antenna3 406, 408, and 410 ~5 located, for example, in a plane generally normal to the X
direction of view and separate from the plane in which transmit antenna 404 is located. A fourth receiving antenna 412 i5 located in still a third plane which is normal to the direction of view and thus in a plane separated from the plane in which th~ other receiving antennas are located. By virtue of this, there is provided means for locating, via triangularixation, a target in space, and thus there is derived sufficient signal information to enable three-dimensional information di~plays. The received signals ~rom receivers 412, 414, 416, and 418 are separately supplied to signal processor and comparator 420, which includes a memory f~r storing all sample~ received and in term~ of their time of receipt.
From this data, one can compute po~ition information by an appropriate comparison as well as target characteri~tics, Ruch as size and reflectivity.
Claims (19)
1. A time domain radar system comprising:
a transmitter comprising:
signal generating means for generating a series of trigger signals at spaced times, a wideband transmitting antenna positioned for transmission into free space, a source of D.C. power, switching means responsive to said trigger signals and coupled to said source of D.C. potential and coupled to said transmitting antenna for abruptly switching between different states of a potential on said transmitting antenna, and transmitting a series of spaced, A.C., carrierless burst signals, each of which is generally monocyclic, into free space; and a radio receiver comprising:
receiving means for receiving and providing an output responsive to wideband signals received from space between the times of transmission of said A.C. carrierless burst signals from said transmitting antenna, detection signal generating means responsive to timing signals for locally generating time spaced local signals, each local signal including a single polarity up to the duration of one polarity of a said transmitted monocyclic burst signal as received, timing means responsive to the times of transmissions of said series of said burst signals for generating, as a set, successive said timing signals and coupling them to said detection signal generating means, each said timing signal of a set occurring at a selected like time after the transmission of a said burst signal, wherein a said selected time is representative of the transit time from said transmitting antenna to a target at a selected distance and back to said receiving means, output signal mixing and integration means for transmission into free space, a source of D.C. power, switching means responsive to said trigger signals and coupled to said source of D.C. potential and coupled to said transmitting antenna for abruptly switching between different states of a potential on said transmitting antenna, and transmitting a series of spaced, A.C., carrierless burst signals, each of which is generally monocyclic, into free space; and a radio receiver comprising:
receiving means for receiving and providing an output responsive to signals received from space between the times of transmission of said A.C. carrierless burst signals from said transmitting antenna, detection signal generating means responsive to timing signals for locally generating time spaced local signals, each local signal including a single polarity up to the period of one polarity of a said transmitted monocyclic burst signal as received, timing means responsive to said sequence of discrete signals from said signal means for generating as a set, successive said timing signals and coupling them to said detection signal generating means, each said timing signal of a set occurring at a selected like time after the transmission of a said burst signal, wherein a said selected time is representative of the transit time from said transmitting antenna to a target at a selected distance and back to said receiving means, output signal mixing and integration means responsive to a said output from said receiving means and a said local burst signal for providing an output signal which is a function signal from mixing a said output and said receiving means and a local burst signal and integrating this function signal for the discrete period of said local signal, and integration means responsive to a successive set of output signals from said signal mixing and responsive to a said output from said receiving means and a said local burst signal for providing an output signal which is a function signal from mixing a said output of said receiving means and a local burst signal and integrating this function signal for the discrete period of said local signal, and integration means responsive to a successive set of output signals from said signal mixing and integration means responsive to a series of transmissions from said transmitting antenna, each of said last-named output signals being for an identical time of transmitted burst signal travel, for providing an integrated signal, said integrated signal being indicative of the presence of signals having been reflected from a target at a selected distance.
a transmitter comprising:
signal generating means for generating a series of trigger signals at spaced times, a wideband transmitting antenna positioned for transmission into free space, a source of D.C. power, switching means responsive to said trigger signals and coupled to said source of D.C. potential and coupled to said transmitting antenna for abruptly switching between different states of a potential on said transmitting antenna, and transmitting a series of spaced, A.C., carrierless burst signals, each of which is generally monocyclic, into free space; and a radio receiver comprising:
receiving means for receiving and providing an output responsive to wideband signals received from space between the times of transmission of said A.C. carrierless burst signals from said transmitting antenna, detection signal generating means responsive to timing signals for locally generating time spaced local signals, each local signal including a single polarity up to the duration of one polarity of a said transmitted monocyclic burst signal as received, timing means responsive to the times of transmissions of said series of said burst signals for generating, as a set, successive said timing signals and coupling them to said detection signal generating means, each said timing signal of a set occurring at a selected like time after the transmission of a said burst signal, wherein a said selected time is representative of the transit time from said transmitting antenna to a target at a selected distance and back to said receiving means, output signal mixing and integration means for transmission into free space, a source of D.C. power, switching means responsive to said trigger signals and coupled to said source of D.C. potential and coupled to said transmitting antenna for abruptly switching between different states of a potential on said transmitting antenna, and transmitting a series of spaced, A.C., carrierless burst signals, each of which is generally monocyclic, into free space; and a radio receiver comprising:
receiving means for receiving and providing an output responsive to signals received from space between the times of transmission of said A.C. carrierless burst signals from said transmitting antenna, detection signal generating means responsive to timing signals for locally generating time spaced local signals, each local signal including a single polarity up to the period of one polarity of a said transmitted monocyclic burst signal as received, timing means responsive to said sequence of discrete signals from said signal means for generating as a set, successive said timing signals and coupling them to said detection signal generating means, each said timing signal of a set occurring at a selected like time after the transmission of a said burst signal, wherein a said selected time is representative of the transit time from said transmitting antenna to a target at a selected distance and back to said receiving means, output signal mixing and integration means responsive to a said output from said receiving means and a said local burst signal for providing an output signal which is a function signal from mixing a said output and said receiving means and a local burst signal and integrating this function signal for the discrete period of said local signal, and integration means responsive to a successive set of output signals from said signal mixing and responsive to a said output from said receiving means and a said local burst signal for providing an output signal which is a function signal from mixing a said output of said receiving means and a local burst signal and integrating this function signal for the discrete period of said local signal, and integration means responsive to a successive set of output signals from said signal mixing and integration means responsive to a series of transmissions from said transmitting antenna, each of said last-named output signals being for an identical time of transmitted burst signal travel, for providing an integrated signal, said integrated signal being indicative of the presence of signals having been reflected from a target at a selected distance.
2. A system as set forth in claim 1 wherein said timing means includes means for selectively delaying the production of timing signals of a discrete said set of successive said timing signals, whereby sensitivity of said radio receiver for different target ranges may be selected.
3. A system as set forth in claim 2 wherein said integration means includes means for sampling discrete output signals from said mixing and integration means and integrating the discrete samples.
4. A system a set forth in claim 3 wherein said integration means includes:
an analog-to-digital converter responsive to said output signals of said mixing and integration means for providing digital signal values of successive said output signals; and digital integration means far digitally integrating said digital signal values and providing said integrated signal.
an analog-to-digital converter responsive to said output signals of said mixing and integration means for providing digital signal values of successive said output signals; and digital integration means far digitally integrating said digital signal values and providing said integrated signal.
5. A system as set forth in claim 1 wherein said signal generating means includes means for providing said trigger signals at varingly spaced times.
6. A system as set forth in claim 1 wherein said switching means is positioned generally adjacent to said transmitting antenna.
7. A system as set forth in claim 6 wherein said switching means includes an impedance coupled in circuit with said transmitting antenna and switching means.
8. A system as set forth in claim 7 wherein said impedance is electrical resistance.
9. A system as set forth in claim 8 wherein said transmitting antenna comprises two elements, and said switching means includes means in series with said electrical resistance and said two elements for discharging said elements.
10. A system as set forth in claim 2 wherein said timing means is responsive to said signal generating means.
11. A time domain radar system comprising:
a transmitter comprising:
control means for generating a first series of signals, oscillator means for generating a second series of signals at a higher rate than said first series, signal means responsive to said first and second series of signals for providing as an output a sequence of discrete signals of said second series of signals which are related to signals of said first series of signals, and said sequence of discrete signals provide trigger signals at spaced times, a wideband transmitting antenna positioned integration means and the occurrence of a series of transmissions from said transmitting antenna, each of said last-named output signals being for an identical time of transmitted burst signal travel, for providing an integrated signal, said integrated signal being indicative of the presence or absence of signals having been reflected from a target at a selected distance.
a transmitter comprising:
control means for generating a first series of signals, oscillator means for generating a second series of signals at a higher rate than said first series, signal means responsive to said first and second series of signals for providing as an output a sequence of discrete signals of said second series of signals which are related to signals of said first series of signals, and said sequence of discrete signals provide trigger signals at spaced times, a wideband transmitting antenna positioned integration means and the occurrence of a series of transmissions from said transmitting antenna, each of said last-named output signals being for an identical time of transmitted burst signal travel, for providing an integrated signal, said integrated signal being indicative of the presence or absence of signals having been reflected from a target at a selected distance.
12. A time domain radar system as set forth in claim 11 wherein said signal means comprises means for effecting as an output signal of said signals of said last-named means the next signal from said oscillator means following a said signal of said first series of signals from said control means.
13. A system as set forth in claim 12 wherein said timing means includes mean for selectively delaying the production of timing signals of a discrete said set of successive said timing signals, whereby sensitivity of said radio receiver for different target ranges may be selected.
14. A system as set forth in claim 12 wherein said integration means includes means for sampling discrete output signals from said mixing and integration means and integrating the discrete samples.
15. A system as set forth in claim 14 wherein said integration means includes:
an analog-to-digital converter responsive to said output signals of said mixing and integration means for providing digital signal values of successive said output signals; and digital integration means for digitally integrating said digital signal values and providing said integrated signal.
an analog-to-digital converter responsive to said output signals of said mixing and integration means for providing digital signal values of successive said output signals; and digital integration means for digitally integrating said digital signal values and providing said integrated signal.
16. A system as set forth in claim 11 wherein said switching means is positioned generally adjacent to said transmitting antenna.
17. A system as set forth in claim 16 wherein said switching means includes an impedance coupled in circuit with said transmitting antenna and switching means.
18. A system as set forth in claim 17 wherein said impedance is electrical resistance.
19. A system as set forth in claim 18 wherein said transmitting antenna comprises two elements, and aid switching means includes means in series with said electrical resistance and said two elements for discharging power between said elements.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA000565712A CA1295712C (en) | 1988-05-02 | 1988-05-02 | Time domain radio transmission system |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA000565712A CA1295712C (en) | 1988-05-02 | 1988-05-02 | Time domain radio transmission system |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1295712C true CA1295712C (en) | 1992-02-11 |
Family
ID=4137954
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000565712A Expired - Lifetime CA1295712C (en) | 1988-05-02 | 1988-05-02 | Time domain radio transmission system |
Country Status (1)
| Country | Link |
|---|---|
| CA (1) | CA1295712C (en) |
-
1988
- 1988-05-02 CA CA000565712A patent/CA1295712C/en not_active Expired - Lifetime
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