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WO2001095508A2 - Appareil, systeme et procede de modulation a une position parmi plusieurs dans un systeme de communications radio a impulsions - Google Patents

Appareil, systeme et procede de modulation a une position parmi plusieurs dans un systeme de communications radio a impulsions Download PDF

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Publication number
WO2001095508A2
WO2001095508A2 PCT/US2001/018332 US0118332W WO0195508A2 WO 2001095508 A2 WO2001095508 A2 WO 2001095508A2 US 0118332 W US0118332 W US 0118332W WO 0195508 A2 WO0195508 A2 WO 0195508A2
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Prior art keywords
signal
impulse
impulse radio
radio signal
positions
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PCT/US2001/018332
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English (en)
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WO2001095508A3 (fr
Inventor
James L. Richards
Vernon R. Brethour
Jack T. Metheney
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Time Domain Corporation
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Priority to AU2001265381A priority Critical patent/AU2001265381A1/en
Publication of WO2001095508A2 publication Critical patent/WO2001095508A2/fr
Publication of WO2001095508A3 publication Critical patent/WO2001095508A3/fr

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B1/7163—Spread spectrum techniques using impulse radio
    • H04B1/71637—Receiver aspects
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
    • H04L25/40—Transmitting circuits; Receiving circuits
    • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
    • H04L25/4902—Pulse width modulation; Pulse position modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/02—Amplitude-modulated carrier systems, e.g. using on-off keying; Single sideband or vestigial sideband modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B1/7163—Spread spectrum techniques using impulse radio
    • H04B1/7176—Data mapping, e.g. modulation

Definitions

  • the present invention relates generally to apparatus, systems and methods for wireless communication. More particularly, the present invention relates to apparatus, systems and methods for modulation in an impulse radio communications system. The present invention also relates to apparatus, systems and methods for transmitting and receiving modulated impulse radio signals.
  • the radio transmission of both analog and digital communications intelligence has normally been effected by one of two methods.
  • a continuous sinusoidal radio frequency carrier is modulated in amplitude according to an intelligence or communications signal.
  • the reverse process that is, demodulation of the carrier
  • the other method employs what is termed frequency modulation.
  • frequency modulation instead of amplitude modulation ofthe carrier signal, the carrier signal is frequency modulated according to the intelligence.
  • circuitry When a frequency modulated signal is received, circuitry is employed which performs what is termed discrimination wherein changes in frequency are changed to changes in amplitude in accordance with the original modulation, and thereby a communications signal is recovered.
  • discrimination wherein changes in frequency are changed to changes in amplitude in accordance with the original modulation, and thereby a communications signal is recovered.
  • a continuous sinusoidal carrier is assigned to and occupies a distinctive frequency bandwidth, or channel.
  • this channel occupies spectrum space which, if interference is to be avoided, cannot be utilized by other transmissions.
  • Today almost every nook and cranny of spectrum space also referred to as the frequency spectrum) is being utilized. Accordingly, there is a tremendous need for some method of expanding the availability of medium for communications. In consideration of this, new methods and systems of communications have been developed that employ a wider frequency spectrum, rather than discrete frequency channels, for radio communications links.
  • impulse radio communications was first fully described in a series of patents, including U.S. Patent Nos. 4,641,317 (issued February 3, 1987), 4,813,057 (issued March 14, 1989), 4,979,186 (issued December 18, 1990) and 5,363,108 (issued November 8, 1994) to Larry W. Fullerton.
  • a second generation of impulse radio patents include U.S. Patent Nos. 5,677,927 (issued October 14, 1997), 5,687,169 (issued November 11, 1997) and 5,832,035 (issued
  • Basic impulse radio transmitters emit short pulses approaching a Gaussian monocycle with tightly controlled pulse-to-pulse intervals.
  • Impulse radio systems typically use pulse position modulation (also referred to as digital time shift modulation), which is a form of time modulation where the value of each instantaneous sample of a modulating signal is caused to modulate the position of an impulse in time. More specifically, in pulse position modulation, the pulse- to-pulse interval is typically varied on a pulse-by-pulse basis by two components: a pseudo-random code component and an information component.
  • each impulse is shifted by a coding amount, and information modulation is accomplished by shifting the coded time position by an additional amount (that is, in addition to PN code dither) in response to an information signal.
  • This additional amount (that is, the information modulation dither) is typically very small relative to the PN code shift.
  • the PN code may command pulse position variations over a range of 100 nsec; whereas, the information modulation may only deviate the impulse position by 150 ps (which is typically less than V. the width of an impulse).
  • the pulse position deviation due to information modulation modulated has been typically less than Vi the width of an impulse so that a single correlator can be used to receive the modulated impulse radio signal.
  • the present invention relates to apparatus, systems and methods for modulation in an impulse radio communications system.
  • the present invention also relates to apparatus, systems and methods for transmitting and receiving modulated impulse radio signals.
  • the present invention is directed to transmitting and receiving one-of-many positions modulated impulse radio signals in an impulse radio communications system.
  • One -of-many positions modulation is also referred to as one-of-N positions modulation or multiple position waveform (MPW) modulation.
  • an impulse is placed within one of a plurality of widely separated positions within a time frame. If two widely separated positions are used within a time frame, then each position can represent one of two data states (e.g., a 0 bit, or a 1 bit). If four widely separated positions are used within a time frame, then four data states can be represented (e.g., each position can represent two bits, i.e., 00, 01, 10, or 11). If eight widely separated positions are used within a time frame, then each position can represent three bits (e.g., 000, 001, 010, 011, 100, 101, 110 , or 111), and so on.
  • the term "widely separated position" minimally means that the positions within a time frame do not overlap.
  • each impulse is dithered by at least VT. the impulse width, i.e., at least 0.25 nsec for this example, based on information modulation.
  • the impulse width i.e., at least 0.25 nsec for this example, based on information modulation.
  • the dither of each impulse based on information modulation is significantly more than (e.g., by a multiple of 10) the width ofthe impulse.
  • the width ofthe impulse For example, where an impulse width is 0.5 nsec, each of the various positions where an impulse can be located within a time frame are preferably separated by at least 5.0 nsec.
  • an impulse radio receiver for demodulating a received impulse radio signal that is modulated according to a one-of-N positions modulation scheme, where N is the number of different possible positions where an impulse can be located within each time frame of the impulse radio signal, includes a timing generator, one or more samplers and a data detector.
  • the timing generator generates N timing signals, wherein each of the N timing signals is separated in time by more than Vi the width of received impulses ofthe received impulse radio signal.
  • the one or more samplers are triggered to sample the received impulse radio signal in accordance with the N timing signals and to provide a first to Nth sampler outputs.
  • the data detector produces one or more demodulation decisions based on the first to Nth sampler outputs.
  • a receiver includes an adjustable precision timing generator, a data correlator, a threshold comparitor, a data sample and hold, a counter, a latch and a data detector.
  • the adjustable precision timing generator generates N timing signals, wherein each of the N timing signals is separated in time from one other by more than Vi the width of received impulses of the received impulse radio signal.
  • the data correlator samples the received impulse radio signal in accordance with the N timing signals to provide a first sampler output through an Nth sampler output.
  • the threshold comparitor compares each ofthe first sampler output through the Nth sampler output to a threshold and outputs a threshold trigger signal when the threshold is exceeded.
  • the data sample and hold (S/H) samples at least one of the first sampler output through the Nth sampler output in response to the threshold trigger signal and outputs one or more corresponding sample values that exceed the threshold.
  • the counter increments a count value in response to receiving each of the N timing outputs, and resets every N timing outputs.
  • the latch stores the count value in response to the threshold trigger signal.
  • the data detector produces a demodulation decision based on at least the count value received from the latch and the corresponding sample value.
  • Impulse radios have typically been resistant to the effects of delayed multipath reflections. This is because delayed multipath reflections typically arrive outside the correlation time and thus have generally been ignored. However, this is not necessarily the case when receiving impulses that have been modulated using a one-of-many positions modulation scheme. Rather, in a one- of-many positions modulation scheme, it is very probable that delayed multipath reflections associated with an impulse placed in a first location will arrive during the correlation times (also referred to as sampling times) of downstream correlations (also referred to as downstream samples).
  • Delayed multipath reflections are one example of what is referred to collectively as ringing or downstream artifacts, which are those signal attributes associated with an impulse that are located later in time than (i.e., downstream from) the intended (or expected) waveform of a received impulse.
  • ringing can be caused by a number of other things, such as by components within an impulse radio transmitter and/or by components within an impulse radio receiver.
  • This ringing can cause demodulation decision errors if the ringing plus noise is greater than the signal (i.e., impulse) plus noise.
  • a receiver used in a one-of-four positions modulation scheme samples a received signal at least four times per frame in an attempt to determine which data state was received. If the sample value (i.e., correlation output) associated with a downstream artifact plus noise (e.g., a sample taken at the second position ofthe four positions) is greater than the sample value ofthe actual impulse plus noise (e.g., taken at the first position), then the receiver can make a wrong demodulation decision regarding which data state (also referred to as, symbol) is associated with the frame ofthe receive signal.
  • a downstream artifact plus noise e.g., a sample taken at the second position ofthe four positions
  • a feature ofthe present invention is the use these downstream artifacts to increase the confidence of demodulation decisions.
  • Another feature ofthe present invention is to adjust the downstream positions (e.,g., the second, third and fourth positions) used during transmission of impulses and to correspondingly adjust the downstream sampling positions used during reception of impulses, so that the disruptive effects of downstream artifacts are reduced.
  • a further feature ofthe present invention is to combine the above features such that downstream positions are adjusted to maximize the confidence of demodulation decisions that include consideration of downstream artifact measurements.
  • downstream artifacts are very useful in environments where ringing (i.e., downstream artifacts) remains somewhat constant over periods of time.
  • ringing i.e., downstream artifacts
  • use of such knowledge can actual corrupt demodulation decisions rather than improve them. This can occur, for example, in environments having constant motion (e.g., movement of a fan blade or the like).
  • the locations of downstream positions are shifted (i.e., adjusted) according to a pattern known by both a transmitter and a receiver.
  • An advantage of this embodiment is that it can improve demodulation decisions made by receivers that are in environments where downstream artifacts unacceptably corrupt demodulation decisions. This is because the shifting of downstream locations breaks up the effects of downstream artifacts.
  • FIG. 1 A illustrates a representative Gaussian Monocy cle waveform in the time domain
  • FIG. IB illustrates the frequency domain amplitude of the Gaussian
  • FIG. 2A illustrates an impulse train comprising pulses as in FIG. 1A;
  • FIG. 2B illustrates the frequency domain amplitude ofthe waveform of FIG. 2A
  • FIG. 3 illustrates the frequency domain amplitude of a sequence of time coded pulses
  • FIG. 4 illustrates a typical received signal and interference signal
  • FIG. 5A illustrates a typical geometrical configuration giving rise to multipath received signals
  • FIG. 5B illustrates exemplary multipath signals in the time domain
  • FIGS. 5C-5E illustrate a signal plot of various multipath environments
  • FIG. 5F illustrates the Rayleigh fading curve associated with non-impulse radio transmission in a multipath environment
  • FIG. 5G illustrates a plurality of multipaths with a plurality of reflectors from a transmitter to a receiver
  • FIG. 5H graphically represents signal strength as volts vs. time in a direct path and multipath environment
  • FIG. 6 is a functional diagram of an exemplary ultra wide band impulse radio transmitter
  • FIG. 7 is a functional diagram of an exemplary ultra wide band impulse radio receiver
  • FIG. 8A illustrates a representative received pulse signal at the input to the correlator
  • FIG. 8B illustrates a sequence of representative impulse signals in the correlation process
  • FIG. 8C illustrates the potential locus of results as a function of the various potential template time positions
  • FIG. 9 illustrates signal waveforms that are useful in explaining a modulation scheme according to an embodiment of the present invention.
  • FIG. 10 is a functional diagram of an impulse radio receiver, according to an embodiment ofthe present invention.
  • FIGS. 11A and 1 IB illustrate correlation functions associated with the receiver of FIG. 10;
  • FIG. 12 is a functional diagram ofthe max value selector ofthe receiver of FIG. 10, according to an embodiment ofthe present invention;
  • FIGS. 13 A and 13B illustrate signal waveforms that are useful in explaining an example of subcarrier modulation
  • FIG. 14 is a functional diagram of an impulse radio receiver, according to an alternative embodiment ofthe present invention.
  • FIG. 15 illustrates signal waveforms that are useful in explaining a one-of four-positions modulation scheme, according to an embodiment of the present invention;
  • FIG. 16 is a functional diagram of an impulse radio receiver, according to an embodiment of the present invention.
  • FIGS. 17A and 17B illustrate signal waveforms that are useful in explaining subcarrier modulation
  • FIG. 18 is a functional diagram of an impulse radio receiver, according to another embodiment ofthe present invention.
  • FIGS. 19 and 20 are functional diagrams of data detectors used in the receiver of FIG. 18, according to embodiments ofthe present invention;
  • FIG. 21 illustrates four possible positions that an impulse may be located in received signal that was modulated using a one-of-four positions modulation scheme
  • FIG. 22 shows an example of correlator output associated with the receiver of FIG. 18;
  • FIGS. 23A - 23D illustrate waveforms that are useful for explaining downstream artifacts that are used during demodulation decisions in an embodiment ofthe present invention
  • FIG. 24 illustrates an example of an artifact table for use in a receiver that receives one-of-four-positions modulated signals
  • FIGS. 25 A and 25B illustrate possible positions that an impulse may be located in two different frames of a received signal that was modulated using a one-of-four positions modulation scheme where downstream positions are shifted, according to an embodiment ofthe present invention.
  • the present invention relates to new types of modulation schemes for use in impulse radio communications systems. Additionally, the present invention relates to the transmitters and receivers that can be used to transmit and receive signals that have been modulated using these new types of modulation schemes.
  • a first data state corresponds to a first position in time of an impulse signal and a second data state corresponds to a second position in time of an impulse signal.
  • two additional data states are created using third and fourth position is time (i.e., in a "one-of-four positions" modulation scheme).
  • teachings ofthe present invention can be used to develop modulation schemes that include even more data states, while still being within the spirit and scope ofthe present invention.
  • the modulation schemes of the present invention provide for increased data speeds in impulse radio communications systems because they enable additional date states to be represented by an impulse or impulse train. Additionally, the modulation schemes of the present invention provide for increased signal to noise ratio and decreased bit error rates over conventional impulse radio modulation schemes.
  • This section is directed to technology basics and provides the reader with an introduction to impulse radio concepts, as well as other relevant aspects of communications theory.
  • This section includes subsections relating to waveforms, impulse trains, coding for energy smoothing and channelization, modulation, reception and demodulation, interference resistance, processing gain, capacity, multipath and propagation, distance measurement, and qualitative and quantitative characteristics of these concepts. It should be understood that this section is provided to assist the reader with understanding the present invention, and should not be used to limit the scope ofthe present invention.
  • Impulse radio refers to a radio system based on short, low duty cycle pulses.
  • An ideal impulse radio waveform is a short Gaussian monocycle. As the name suggests, this waveform attempts to approach one cycle of radio frequency (RF) energy at a desired center frequency. Due to implementation and other spectral limitations, this waveform may be altered significantly in practice for a given application. Most waveforms with enough bandwidth approximate a Gaussian shape to a useful degree.
  • RF radio frequency
  • Impulse radio can use many types of modulation, including AM, time shift (also referred to as pulse position) and M-ary versions.
  • the time shift method has simplicity and power output advantages that make it desirable.
  • the time shift method is used as an illustrative example.
  • the pulse-to-pulse interval can be varied on a pulse-by-pulse basis by two components: an information component and a pseudo-random code component.
  • an information component e.g., a digital signal processor
  • a pseudo-random code component e.g., a pseudo-random code component
  • conventional spread spectrum systems make use of pseudo-random codes to spread the normally narrow band information signal over a relatively wide band of frequencies.
  • a conventional spread spectrum receiver correlates these signals to retrieve the original information signal.
  • the pseudorandom code for impulse radio communications is not necessary for energy spreading because the monocycle pulses themselves have an inherently wide bandwidth. Instead, the pseudo-random code is used for channelization, energy smoothing in the frequency domain, resistance to interference, and reducing the interference potential to nearby receivers.
  • each data bit typically time position modulates many pulses of the periodic timing signal. This yields a modulated, coded timing signal that comprises a train of identically shaped pulses for each single data bit.
  • the impulse radio receiver integrates multiple pulses to recover the transmitted information.
  • Impulse radio refers to a radio system based on short, low duty cycle pulses.
  • the resulting waveform approaches one cycle per impulse at the center frequency.
  • each impulse consists of a burst of cycles usually with some spectral shaping to control the bandwidth to meet desired properties such as out of band emissions or in-band spectral flatness, or time domain peak power or burst off time attenuation.
  • desired properties such as out of band emissions or in-band spectral flatness, or time domain peak power or burst off time attenuation.
  • One such waveform model that has been useful is the Gaussian monocycle as shown in FIG. 1A. This waveform is representative of the transmitted impulse produced by a step function into an ultra- ideband antenna.
  • the basic equation normalized to a peak value of 1 is as follows:
  • ⁇ is a time scaling parameter
  • t is time
  • fmondf i s me waveform voltage
  • e is the natural logarithm base
  • the center frequency (f c ), or frequency of peak spectral density is:
  • Impulse radio systems can deliver one or more data bits per impulse; however, impulse radio systems more typically use impulse trains, not single pulses, for each data bit. As described in detail in the following example system, the impulse radio transmitter produces and outputs a train of pulses for each bit of information.
  • FIGS. 2A and 2B are illustrations of the output of a typical 10 Mpps system with uncoded, unmodulated, 0.5 nanosecond (nsec) pulses 102.
  • FIG. 2A shows a time domain representation of this sequence of pulses 102.
  • Fig 2B which shows 60 MHz at the center ofthe spectrum for the waveform of FIG. 2A, illustrates that the result of the impulse train in the frequency domain is to produce a spectrum comprising a set of comb lines 204 spaced at the frequency of the 10 Mpps pulse repetition rate.
  • the envelope ofthe line spectrum follows the curve ofthe single impulse spectrum 104 of FIG. IB.
  • the power ofthe impulse train is spread among roughly two hundred comb lines. Each comb line thus has a small fraction ofthe total power and presents much less of an interference problem to receiver sharing the band.
  • impulse radio systems typically have very low average duty cycles resulting in average power significantly lower than peak power.
  • the duty cycle ofthe signal in the present example is 0.5%, based on a 0.5 nsec impulse in a 100 nsec interval.
  • any aspect of the waveform can be modulated to convey information.
  • Amplitude modulation, phase modulation, frequency modulation, time shift modulation and M-ary versions of these have been proposed. Both analog and digital forms have been implemented.
  • digital time shift modulation has been demonstrated to have various advantages and can be easily implemented using a correlation receiver architecture.
  • Digital time shift modulation can be implemented by shifting the coded time position by an additional amount (that is, in addition to PN code dither) in response to the information signal. This amount is typically very small relative to the PN code shift. In a 10 Mpps system with a center frequency of 2 GHz., for example, the PN code may command pulse position variations over a range of
  • the information modulation may only deviate the impulse position by 150 ps.
  • each impulse is delayed a different amount from its respective time base clock position by an individual code delay amount plus a modulation amount, where n is the number of pulses associated with a given data symbol digital bit.
  • Modulation further smooths the spectrum, minimizing structure in the resulting spectrum.
  • impulse radios are able to perform in these environments, in part, because they do not depend on receiving every impulse.
  • the impulse radio receiver performs a correlating, synchronous receiving function (at the RF level) that uses a statistical sampling and combining of many pulses to recover the transmitted information.
  • Impulse radio receivers typically integrate from 1 to 1000 or more pulses to yield the demodulated output.
  • the optimal number of pulses 'over which the receiver integrates is dependent on a number of variables, including impulse rate, bit rate, interference levels, and range.
  • the PN coding also makes impulse radios highly resistant to interference from all radio communications systems, including other impulse radio transmitters. This is critical as any other signals within the band occupied by an impulse signal potentially interfere with the impulse radio. Since there are currently no unallocated bands available for impulse systems, they must share spectrum with other conventional radio systems without being adversely affected.
  • the PN code helps impulse systems discriminate between the intended impulse transmission and interfering transmissions from others.
  • FIG.4 illustrates the result of a narrow band sinusoidal interference signal 402 overlaying an impulse radio signal 404.
  • the input to the cross correlation would include the narrow band signal 402, as well as the received ultrawide-band impulse radio signal 404.
  • the input is sampled by the cross correlator with a PN dithered template signal 406. Without PN coding, the cross correlation would sample the interfering signal 402 with such regularity that the interfering signals could cause significant interference to the impulse radio receiver.
  • the transmitted impulse signal is encoded with the PN code dither (and the impulse radio receiver template signal 406 is synchronized with that identical PN code dither)
  • the correlation samples the interfering signals pseudo-randomly.
  • the samples from the interfering signal add incoherently, increasing roughly according to square root of the number of samples integrated; whereas, the impulse radio samples add coherently, increasing directly according to the number of samples integrated.
  • integrating over many pulses overcomes the impact of interference.
  • processing gain which quantifies the decrease in channel interference when wide-band communications are used, is the ratio ofthe bandwidth ofthe channel to the bit rate ofthe information signal.
  • processing gain For example, a direct sequence spread spectrum system with a 10 kHz information bandwidth and a 10 MHz channel bandwidth yields a processing gain of 1000 or 30 dB.
  • far greater processing gains are achieved with impulse radio systems, where for the same 10 kHz information bandwidth is spread across a much greater 2 GHz channel bandwidth, the theoretical processing gain is 200,000 or 53 dB.
  • V 2 ,,, is the total interference signal to noise ratio variance
  • N is the number of interfering users
  • ⁇ 2 is the signal to noise ratio variance resulting from one of the interfering signals with a single impulse cross correlation
  • Z is the number of pulses over which the receiver integrates to recover the modulation.
  • impulse radio is its resistance to multipath fading effects.
  • Conventional narrow band systems are subject to multipath tlirough the Rayleigh fading process, where the signals from many delayed reflections combine at the receiver antenna according to their seemingly random relative phases. This results in possible summation or possible cancellation, depending on the specific propagation to a given location.
  • This situation occurs where the direct path signal is weak relative to the multipath signals, which represents a major portion of the potential coverage of a radio system. In mobile systems, this results in wild signal strength fluctuations as a function of distance traveled, where the changing mix of multipath signals results in signal strength fluctuations for every few feet of travel.
  • FIG. 5A three propagation paths are shown.
  • the direct path representing the straight line distance between the transmitter and receiver is the shortest.
  • Path 1 represents a grazing multipath reflection, which is very close to the direct path.
  • Path 2 represents a distant multipath reflection.
  • elliptical (or, in space, ellipsoidal) traces that represent other possible locations for reflections with the same time delay.
  • FIG. 5B represents a time domain plot ofthe received waveform from this multipath propagation configuration. This figure comprises three doublet pulses as shown in FIG.
  • the direct path signal is the reference signal and represents the shortest propagation time.
  • the path 1 signal is delayed slightly and actually overlaps and enhances the signal strength at this delay value. Note that the reflected waves are reversed in polarity.
  • the path 2 signal is delayed sufficiently that the waveform is completely separated from the direct path signal. If the correlator template signal is positioned at the direct path signal, the path 2 signal will produce no response. It can be seen that only the multipath signals resulting from very close reflectors have any effect on the reception of the direct path signal.
  • the multipath signals delayed less than one quarter wave are the only multipath signals that can attenuate the direct path signal. This region is equivalent to the first Fresnel zone familiar to narrow band systems designers. Impulse radio, however, has no further nulls in the higher Fresnel zones. The ability to avoid the highly variable attenuation from multipath gives impulse radio significant performance advantages.
  • FIG. 5A illustrates a typical multipath situation, such as in a building, where there are many reflectors 5A04, 5A05 and multiple propagation paths 5A02, 5A01.
  • a transmitter TX 5A06 transmits a signal which propagates along the multiple propagation paths 5A02, 5A04 to receiver RX 5 A08, where the multiple reflected signals are combined at the antenna.
  • FIG. 5B illustrates a resulting typical received composite pulse waveform resulting from the multiple reflections and multiple propagation paths 5A01, 5 A02.
  • the direct path signal 5 A01 is shown as the first pulse signal received.
  • the multiple reflected signals (“multipath signals", or "multipath" comprise the remaining response as illustrated.
  • FIGs. 5C, 5D, and 5E represent the received signal from a TM-UWB transmitter in three different multipath environments. These figures are not actual signal plots, but are hand drawn plots approximating typical signal plots.
  • FIG. 5C illustrates the received signal in a very low multipath environment. This may occur in a building where the receiver antenna is in the middle of a room and is one meter from the transmitter. This may also represent signals received from some distance, such as 100 meters, in an open field where there are no objects to produce reflections. In this situation, the predominant pulse is the first received pulse and the multipath reflections are too weak to be significant.
  • FIG. 5D illustrates an intermediate multipath environment. This approximates the response from one room to the next in a building.
  • FIG. 5E approximates the response in a severe multipath environment such as: propagation through many rooms; from corner to corner in a building; within a metal cargo hold of a ship; within a metal truck trailer; or within an intermodal shipping container.
  • the main path signal is weaker than in FIG. 5D.
  • the direct path signal power is small relative to the total signal power from the reflections.
  • An impulse radio receiver in accordance with the present invention can receive the signal and demodulate the information using either the direct path signal or any multipath signal peak having sufficient signal to noise ratio.
  • the impulse radio receiver can select the strongest response from among the many arriving signals.
  • dozens of reflections would have to be cancelled simultaneously and precisely while blocking the direct path - a highly unlikely scenario.
  • This time separation of multipath signals together with time resolution and selection by the receiver permit a type of time diversity that virtually eliminates cancellation of the signal.
  • performance is further improved by collecting the signal power from multiple signal peaks for additional signal to noise performance. Wl ere the system of FIG.
  • the received signal is a sum of a large number of sine waves of random amplitude and phase.
  • the resulting envelope amplitude has been shown to follow a Rayleigh probability distribution as follows:
  • r is the envelope amplitude ofthe combined multipath signals
  • 2s 2 is the RMS power ofthe combined multipath signals
  • impulse radio In a high multipath environment such as inside homes, offices, warehouses, automobiles, trailers, shipping containers, or outside in the urban canyon or other situations where the propagation is such that the received signal is primarily scattered energy, impulse radio, according to the present invention, can avoid the Rayleigh fading mechanism that limits performance of narrow band systems.
  • FIG. 5G and 5H in a transmit and receive system in a high multipath environment 5G00, wherein the transmitter 5G06 transmits to receiver 5G08 with the signals reflecting off reflectors 5G03 which form multipaths 5G02.
  • the direct path is illustrated as 5G01 with the signal graphically illustrated at 5H02, with the vertical axis being the signal strength in volts and horizontal axis representing time in nanoseconds.
  • Multipath signals are graphically illustrated at 5H04. 11.10. Distance Measurement
  • Impulse systems can measure distances to extremely fine resolution because of the absence of ambiguous cycles in the waveform.
  • Narrow band systems are limited to the modulation envelope and cannot easily distinguish precisely which RF cycle is associated with each data bit because the cycle-to-cycle amplitude differences are so small they are masked by link or system noise. Since the impulse radio waveform has no multi-cycle ambiguity, this allows positive determination of the waveform position to less than a wavelength - potentially, down to the noise floor ofthe system.
  • This time position measurement can be used to measure propagation delay to determine link distance, and once link distance is known, to transfer a time reference to an equivalently high degree of precision.
  • the inventors of the present invention have built systems that have shown the potential for centimeter distance resolution, which is equivalent to about 30 ps of time transfer resolution. See, for example, commonly owned, co-pending applications 09/045,929, filed March 23,
  • An exemplary embodiment of an impulse radio transmitter 602 of an impulse radio communication system having one subcarrier channel will now be described with reference to Fig. 6.
  • the transmitter 602 comprises a time base 604 that generates a periodic timing signal 606.
  • the time base 604 typically comprises a voltage controlled oscillator (VCO), or the like, having a high timing accuracy and low jitter, on the order of picoseconds (ps).
  • VCO voltage controlled oscillator
  • the voltage control to adjust the VCO center frequency is set at calibration to the desired center frequency used to define the transmitter's nominal impulse repetition rate.
  • the periodic timing signal 606 is supplied to a precision timing generator 608.
  • the precision timing generator 608 supplies synchronizing signals 610 to the code source 612 and utilizes the code source output 614 together with an internally generated subcarrier signal (which is optional) and an information signal 616 to generate a modulated, coded timing signal 618.
  • the code source 612 comprises a storage device such as a random access memory (RAM), read only memory (ROM), or the like, for storing suitable PN codes and for outputting the PN codes as a code signal 614.
  • RAM random access memory
  • ROM read only memory
  • maximum length shift registers or other computational means can be used to generate the PN codes.
  • An information source 620 supplies the information signal 616 to the precision timing generator 608.
  • the information signal 616 can be any type of intelligence, including digital bits representing voice, data, imagery, or the like, analog signals, or complex signals.
  • a pulse generator 622 uses the modulated, coded timing signal 618 as a trigger to generate output pulses.
  • the output pulses are sent to a transmit antenna 624 via a transmission line 626 coupled thereto.
  • the output pulses are converted into propagating electromagnetic impulses by the transmit antenna 624.
  • the electromagnetic pulses are called the emitted signal, and propagate to an impulse radio receiver 702, such as shown in Fig. 7, through a propagation medium, such as air, in a radio frequency embodiment.
  • the emitted signal is wide-band or ultrawide-band, approaching a monocycle impulse as in Fig. 1 A.
  • the emitted signal can be spectrally modified by filtering of the pulses. This filtering will usually cause each monocycle impulse to have more zero crossings (more cycles) in the time domain.
  • the impulse radio receiver can use a similar waveform as the template signal in the cross correlator for efficient conversion.
  • the receiver 702 comprises a receive antenna 704 for receiving a propagated impulse radio signal 706.
  • a received signal 708 from the receive antenna 704 is coupled to a cross correlator or sampler 710 to produce a baseband output 712.
  • the cross correlator or sampler 710 includes multiply and integrate functions together with any necessary filters to optimize signal to noise ratio.
  • the baseband ⁇ output 712 can be applied to a digitizing logic block 713 to produce a digitized or digital baseband output 713a.
  • Digitizing logic block 712 can include, for example, a Sample-and-Hold (S/H) stage followed by an Analog- to-Digital (A/D) converter.
  • Digital baseband output 713a includes digital words representing sampled amplitudes of digital baseband output 712.
  • An advantage of digitizing baseband output 712 is that all subsequent signal processing of digital baseband output 713a can be implemented using digital techniques in a digital baseband architecture.
  • Such a digital baseband architecture can be implemented using, for example, digital logic in a gate array, a digital signal processor, and/or a microprocessor.
  • the digital baseband architecture is inherently immune to adverse effects arising from stressful environmental factors, such as impulse radio operating temperature variations and mechanical vibration.
  • the digital baseband architecture has manufacturing advantages over an analog architecture, such as improved manufacturing reproducibility and reliability.
  • the receiver 702 also includes a precision timing generator 714, which receives a periodic timing signal 716 from a receiver time base 718. This time base 718 is adjustable and controllable in time, frequency, or phase, as required by the lock loop in order to lock on the received signal 708.
  • the precision timing generator 714 provides synchronizing signals 720 to the code source 722 and receives a code control signal 724 from the code source 722.
  • the precision timing generator 714 utilizes the periodic timing signal 716 and code control signal 724 to produce a coded timing signal 726.
  • the template generator 728 is triggered by this coded timing signal 726 and produces a train of template signal pulses 730 ideally having waveforms substantially equivalent to each pulse ofthe received signal 708.
  • the code for receiving a given signal is the same code utilized by the originating transmitter 602 to generate the propagated signal 706.
  • the timing of the template pulse train 730 matches the timing of the received signal pulse train 708, allowing the received signal 708 to be synchronously sampled in the correlator 710.
  • the correlator 710 ideally comprises a multiplier followed by a short-term integrator to sum the multiplier product over the pulse interval. Further examples and details of correlation and sampling processes can be found in the above-reference commonly owned patents and commonly owned and copending U.S. Patent Application No. 09/356,384, filed July 16, 1999, entitled "Baseband Signal Converter Device for a Wideband
  • the digitized output of the correlator 710 also called digital baseband signal 713a, is coupled to a subcarrier demodulator 732, which demodulates the subcarrier information signal from the subcarrier. If digitizing logic block 713 is not used in the receiver, then baseband output 712 is provided directly from correlator 712 to the input of subcarrier demodulator 732.
  • the purpose ofthe optional subcarrier process, when used, is to move the information signal away from DC (zero frequency) to improve immunity to low frequency noise and offsets.
  • the output of the subcarrier demodulator 732 is then filtered or integrated in a pulse summation stage 734.
  • the pulse summation stage produces an output representative of the sum of a number of pulse signals comprising a single data bit.
  • the output ofthe pulse summation stage 734 is then compared with a nominal zero (or reference) signal output in a detector stage 738 to determine an output signal 739 representing an estimate of the original information signal 616.
  • the digital baseband signal 713 a is also input to a lowpass filter 742 (also referred to as lock loop filter 742).
  • a control loop comprising the lowpass filter 742, time base 718, precision timing generator 714, template generator 728, and correlator 710 is used to generate a filtered error signal 744.
  • the filtered error signal 744 provides adjustments to the adjustable time base 718 to time position the periodic timing signal 726 in relation to the position ofthe received signal 708.
  • substantial economy can be achieved by sharing part or all of several ofthe functions ofthe transmitter 602 and receiver 702. Some of these include the time base 718, precision timing generator 714, code source 722, antenna 704, and the like.
  • an impulse is placed within one of a plurality of widely separated positions within a time frame. If two widely separated positions are used within a time frame, then each position can represent one of two data states (e.g., a 0 bit, or a 1 bit). If, for example, four widely separated positions are used within a time frame, then four data states can be represented (e.g., each position can represent two bits, i.e., 00, 01 , 10, or 11). If eight widely separated positions are used within a time frame, then each position can represent three bits (e.g., 000, 001, 010, 011, 100, 101, 110 , or 111).
  • each impulse is dithered by at least Vi the impulse width, i.e., at least 0.25 nsec for this example, based on information modulation.
  • the dither of each impulse based on information modulation is significantly more than the impulse width ofthe impulse, e.g., by 5.0 nsec for this example.
  • an impulse waveform 902 (or a plurality of impulse waveforms 902) is used to represent a binary "0" symbol
  • an impulse waveform 904 (or a plurality of impulse waveforms 904) is used to represent a binary "1" symbol.
  • waveforms 902 and 904 can be described mathematically by:
  • ⁇ is a time scaling parameter
  • t is time
  • f m o m ft is me waveform voltage
  • e is the natural logarithm base.
  • Impulses 902 and 904 are exemplary waveforms associated with transmitted signals (e.g., signals transmitted through the air from a transmitter to a receiver). Once impulses 902 and 904 are received by an antenna of a receiver, their waveforms typically resemble waveform 906 and waveform 908, respectively. More specifically, waveform 906 is approximately the first derivative of waveform 902, and waveform 908 is approximately the first derivative of waveform 904. This occurs due to the receive antenna response. Because waveforms 906 and 908 resemble a "w”, they shall be referred to as "w- pulses" or "triplets".
  • w- pulse 906 (or a plurality of w-pulses 906) corresponds to a binary "0" and w-pulse 908 (or a plurality w-pulses 908) corresponds to a binary "1". It is noted that a receive antenna does not necessarily differentiate a received signal. Thus, if a receive antenna does not differentiate a received signal, then the pulse waveforms of a received signal should resemble the pulse waveforms of a transmitted signal.
  • impulse radio systems can deliver one or more data bits per impulse.
  • impulse radio systems more typically use impulse trains, not single pulses, for each data bit.
  • a train of pulses 902 e.g., 100 pulses 902
  • a train of pulses 904 e.g., 100 pulses 904
  • impulse trains are often used because ofthe additional benefits that can be obtained by using more than one impulse to represent one digital information bit.
  • the received signal from the ensemble of pulses associated with each bit is combined in a process referred to as integration gain.
  • the combination process is basically the summation ofthe received signal plus noise energy associated with each impulse over the number of pulses for each bit.
  • the voltage signal-to-noise ratio improves roughly by the square root ofthe number of pulses summed. Proper summation requires that the timing be stable and accurate over the entire integration (summing) time.
  • a transmitter that is substantially similar to transmitter 602, described above in the discussion of FIG. 6, can be used to transmits impulses that are modulated using the above described one-of-many positions modulation scheme (e.g., to transmit impulses 902 and 904). What is important is that precision timing generator 608 produces timing signal 618 (which, may or may not be coded, depending on implementation) based on the one-of-many positions modulation scheme that has been chosen for implementation.
  • FIG. 10 is a block diagram of an exemplary impulse radio receiver 1002 for receiving one-of-many positions modulated signals, according to an embodiment of the present invention. More specifically, receiver 1002 is for receiving one-of-two positions modulated signals. An example of a one-of-two positions modulation scheme was described above in connection with FIG. 9.
  • receiver 1002 includes an antenna 1004 for receiving a propagated impulse radio signal.
  • antenna 1004 is designed such that it differentiates the received propagated impulse radio signal.
  • received signal 1006 resembles the first derivative of the propagated impulse radio signal.
  • waveform 906 is the first derivative of impulse 902
  • waveform 908 is the first derivative of impulse 904.
  • antenna 1004 does not differentiate the received propagated impulse radio signal.
  • Received signal 1006 is input to a first data correlator 1008 (also called first sampler 1008).
  • first data correlator 1008 By correlating received signal 1006 with a template signal 1074 (also referred to as a reference signal 1074), discussed in more detail below, correlator 1008 produces a first baseband output signal 1010 (also referred to as first correlator output signal 1010, or first sample 1010).
  • first data correlator 1008 By correlating received signal 1006 with a template signal 1074 (also referred to as a reference signal 1074), discussed in more detail below, correlator 1008 produces a first baseband output signal 1010 (also referred to as first correlator output signal 1010, or first sample 1010).
  • 1008 ideally comprises a multiplier followed by a short term integrator to sum the multiplied product over the pulse interval (as shown in FIGS. 11 A and 1 IB).
  • Received signal 1006 is also input to a second data correlator 1026 (also referred to as second sampler 1026).
  • second data correlator 1026 By correlating received signal 1006 with a delayed template signal 1082, correlator 1026 produces a second baseband output signal 1032 (also referred to as second correlator output signal 1032, or second sample 1032).
  • Second data correlator 1026 ideally comprises a multiplier followed by a short term integrator to sum the multiplied product over the pulse interval (as shown in FIGS. 11 A and 1 IB).
  • Received signal 1006 is also input to lock loop correlator 1086 that is used in a lock loop that corrects drifts in a receiver time base 1054. It is important to correct drifts in time base 1054 so that first data correlator 1008 and second data correlator 1026 sample received signal 1006 at the appropriate times.
  • the lock loop function is described in additional detail below.
  • Receiver 1002 also includes a precision timing generator 1060, which receives a periodic timing signal 1056 from receiver time base 1054.
  • Time base 1054 is adjustable and controllable in time, frequency, and/or phase, as required by the lock loop (described below) in order to lock on the received signal 1006.
  • Precision timing generator 1060 provides a synchronization signal 1066 to an optional code generator 1064 and receives a code control signal 1062 (also referred to as coding signal 1062) from optional code generator 1064.
  • Precision timing generator 1060 utilizes periodic timing signal 1056 and optional code control signal 1062 to produce a (coded) timing signal 1070.
  • Template generator 1072 (also referred to as pulse generator 1072, or reference signal generator 1072) is triggered by (coded) timing signal 1070 and produces a train of template signal pulses 1074 (also referred to as reference signal pulses 1074) ideally having waveforms substantially equivalent to each impulse of received signal
  • template signal 1074 ideally 1074 consists of pulses that are substantially equivalent to the first derivative ofthe propagated pulses. More likely, template signal 1074 consists of square pulses, because square pulses are much easier to generate. Where template signal 1074 consists of square pulses, template generator 1072 is not necessary if precision timing generator 1060 outputs square pulses having the appropriate shape to be used by correlators 1008 and 1026. Further, where template signal 1074 consists of square pulses, the width of each square pulse is preferably somewhat less than Vi the pulse width of a received impulse and centered about the center peak of the received impulse. For example, where received impulses are approximately 0.5 nsec wide, the square pulses of template signal are preferably approximately 0.15 nsec wide.
  • Template signal 1074 is used by first data correlator 1008 to sample received signal 1006, as discussed above. Template signal 1074 is also delayed by an amount of time (e.g., 5.0 nsec), and the delayed template signal 1082 is used by second data correlator 1026 to sample received signal 1006, as discussed above.
  • the amount of time that template signal 1074 is delayed is the amount of separation that is between the two different information modulation states in the one-of-two positions modulation scheme. In the example shown in FIG. 9, the two different impulse modulations states are 5.0 nsec apart. Thus, a 5.0 nsec delay 1080 can be used to produce the appropriate delay.
  • time base 1054, precision timing generator 1060, template generator 1072 and delays 1080 and 1076 can be combined into a single sampling/timing generator that provides the appropriate reference signals to first data correlator 1008, second data correlator 1026 and lock loop correlator 1086 at the precise times to appropriately sample received signal 1006.
  • received signal 1006 is sampled at each time position that an impulse may exist within each frame.
  • received signal 906 is sampled at a point in time (e.g., a zero crossing of a received impulse) that enables corrections of timing offsets.
  • the sampling used to correct timing offsets does not need to occur every frame, only enough times to track oscillator instability and potential motion between a transmitter and receiver.
  • the inventors have found that a IKHz lock bandwidth is suitable for many applications.
  • code generator 1064 If code generator 1064 is used, then the code for receiving a given signal is the same code utilized by the originating transmitter (e.g., used by code generator 612 of transmitter 602) to generate the propagated signal.
  • the timing of template impulse train 1074 also referred to as template signal 1074 or reference signal 1074 matches the timing of received signal impulse train 1006, allowing received signal 1006 to be synchronously sampled by correlators 1008 and 1026.
  • Baseband output 1010 of first data correlator 1008 is preferably provided to an analog to digital converter (A/D) 1012, which outputs a digital signal 1014 representative of output 1010 of first data correlator 1008.
  • baseband output 1032 of second data correlator 1026 is provided to an analog to digital converter (A/D) 1034, which outputs a digital signal 1036 representative of output 1032 of second data correlator 1026.
  • Digital signals 1014 and 1036 are provided to optional subcarrier demodulator 1016, if subcarrier modulation was used by the transmitter that generated received signal 1006. Otherwise, digital signals 1014 and 1036 are provided directly to summing accumulators 1020 and 1040, respectively. Additional details of subcarrier demodulator 1016 are discussed below.
  • An output 1024 of summing accumulator 1020 and an output 1044 of summing accumulator 1040 are both provided to a max value selector 1027.
  • Max value selector 1027 and subcarrier demodulator 1016 are discussed in more detail below. However, additional details ofthe correlation process are provided first.
  • data detector 1012 and 1034 can be thought of as being components of a data detector 1003 (shown by dotted lines).
  • data detector 1003 shows the exact structure of data detector 1003 can be modified and simplified while still being within the spirit and scope of the present invention.
  • data detector 1026 produces a data signal based on outputs 1010 and 1032 of first and second correlators 1008 and 1026.
  • FIGS. 11A and 11B show results of an exemplary correlation process performed by first data correlator 1008 and second data correlator 1026.
  • first data correlator 1008 is shown as consisting of a multiplier 1106 followed by a pulse integrator 1108 that sums the multiplied product over at least a portion of the pulse interval.
  • second data correlator 1026 is shown as consisting of a multiplier 1116 followed by a pulse integrator 1118.
  • a received impulse 1102a e.g., of received signal
  • first data correlator 1008 is provided to first data correlator 1008 and second data correlator 1026, as was discussed above in connection with FIG. 10.
  • a reference pulse 1104a i.e., of template signal 1074 is provided to first data correlator 1008. Notice that since the received impulse 1102a and the reference pulse 1104a are offset in time (i.e., they do not overlap in time), output 1010 of first data correlator 1008 is substantially zero volts (as shown by signal 1110a).
  • a reference pulse 1112a (e.g., of delayed template signal 1082) is provided to second data correlator 1026. Notice that since the received impulse 1102a and the reference pulse 1112a are substantially aligned in time, output 1032 of second data correlator 1026 is a positive voltage (as shown by signal 1120a).
  • a different received impulse 1102b (e.g., of received signal 1006) is provided to first data correlator 1008 and second data correlator 1026.
  • a reference pulse 1104b (e.g., of template signal 1074) is also provided to first data correlator 1008. Notice that since the received impulse 1102b and the reference pulse 1104b are substantially aligned in time, output 1010 of first data correlator 1008 is a positive voltage (as shown by signal 1110b).
  • a reference pulse 1112b (e.g., of delayed template signal 1082) is provided to second data correlator 1026. Notice that since the received impulse 1102b and the reference pulse 1112b are offset in time (i.e., they do not overlap in time), output 1032 of second data correlator 1026 is substantially zero volts (as shown by signal 1120b).
  • Max value selector 1027 determines the data states (e.g., bit or bits) that an impulse, or a plurality of pulses (e.g., 100 pulses), represent. For example, assuming that 100 pulses of received signal 1006 are used to represent each data bit, max value selector 1027 makes a decision whether each 100 pulses represent a "0" bit or "1 " bit.
  • max value selector 1027 comprises a comparitor 1202.
  • the signal applied to the (+) input terminal i.e., signal 1044
  • the signal applied to the (-) input terminal i.e., signal 1024
  • output signal 1046 assumes a HIGH output state, which for example corresponds to a " 1 " bit.
  • the signal applied to the (+) input terminal i.e., signal 1044
  • the signal applied to the (+) input terminal i.e., signal 1044
  • output signal 1046 assumes a LOW output state, which for example corresponds to a "0" bit.
  • max value selector 1027 receives a value associated with a "0" bit (e.g., signal 1044)
  • max value selector 1027 is essentially a digital comparitor that compares two values and outputs a "0" or a "1", depending on which ofthe two values is greater.
  • received signal 1006 consists of 100 pulses 1102a (i.e., 100 frames each with an impulse 1102a) .
  • A/D converter 1012 converts each voltage to a corresponding substantially zero value.
  • received signal 1006 was not modulated by a subcarrier, and subcarrier demodulator 1016 is not used.
  • signal 1014 is identical to 1018 (referred to collectively as signal
  • signal 1014/1018 and signal 1036 is identical to signal 1038 (referred to collectively as signal 1036/1039).
  • Accumulator 1020 adds the 100 substantially zero values (signal 1014/1018) and provides the sum (1024) to max value selector 1027.
  • second data correlator 1026 receives the same 100 impulses 1102a. This causes signal 1032 (output from second data correlator 1026, and input to A/D converter 1034) to consist of 100 positive voltage values (i.e., signal 1120a). A/D converter 1034 converts each positive voltage to a corresponding positive value. Accumulator 1040 adds the 100 values (signal 1036/1038) and provides the sum (signal 1044) to max value selector 1027. In this example, max value selector 1027 will determine that sum 1044 is greater than sum 1024, and thus that the 100 pulses 1102a represent a "1 " bit. As a result, max value selector 1027 outputs a data signal 1046 that signifies a "1" bit.
  • received signal 1006 consists of 100 pulses 1102b (i.e., 100 frames each with an impulse 1102b).
  • signal 1010 output from first data correlator 1008, and input to A/D converter 1012 to consists of 100 positive voltage values (i.e., signal 1110b).
  • A/D converter 1012 converts each positive voltage to a corresponding positive value.
  • received signal 1006 was not modulated by a subcarrier, and thus that subcarrier demodulator 1016 is not used.
  • signal 1014 is identical to 1018 and signal 1036 is identical to signal 1038.
  • Accumulator 1020 adds the 100 positive values (signal 1014/1018) and provides the sum (1024) to max value selector 1027.
  • second data correlator 1026 receives the same 100 impulses 1102b. This causes signal 1032 (output from second data correlator 1032, and input to A/D converter 1040) to consist of 100 substantially zero voltage values (i.e., signal 1120b). A/D converter 1040 converts each substantially zero voltage to a corresponding substantially zero value. Accumulator 1040 adds the 100 values (signal 1036/1038) and provides the sum (signal 1044) to max value selector 1027. In this example, max value selector
  • a max value selector can be designed to distinguish between states other than a "0" bit and a " 1 " bit.
  • a max value selector 1627 of a receiver 1602 shown in FIG. 16, and discussed below
  • receives one-of-four positions modulated signals can distinguish between four data states (e.g., bits "00", "01", "10” and "11").
  • a lock loop (also referred to as a control loop) is used to generate an error signal 1052 that corrects any drifts in time base 1054. More specifically, a control loop including lock loop filter 1050, time base 1054, precision timing generator 1060, template generator 1072, delay 1076, lock loop correlator 1086, A/D converter 1090, accumulator 1094 and lock path switch 1048, is used to generate error signal 1052. Error signal 1052 provides adjustments to the adjustable time base 1054 to time position periodic timing signal 1056 in relation to the position of received signal 1006. The function of the lock loop is described in more detail, below.
  • Received signal 1006 is input to a lock loop correlator 1086.
  • lock loop correlator 1086 correlates received signal 1006 with a slightly delayed template signal 1078 (generated by delay 1076) and outputs a lock loop correlator output 1088.
  • the delay caused by delay 1074 is precisely selected such that an output of lock loop correlator 1086 is theoretically zero when received signal 1006 and non-delayed template signal 1074 are synchronized.
  • delay 1076 is precisely selected such that lock loop correlator 1086 samples received signal
  • delay 1076 delays template signal 1074 by a quarter of an impulse width.
  • the width of each impulse is 0.5 nsec (as shown in FIG. 9)
  • output 1088 of lock loop correlator 1086 will be a positive or negative value that is used to correct time base 1054.
  • A/D converter 1090 the correction of time base 1054 is performed in the digital domain.
  • timing errors should only be measured when first data correlator 1008 is actually sampling an impulse. Timing errors should not be measured when first data correlator 1008 is not sampling an impulse, but second data correlator is sampling an impulse. This is because the lock loop is arranged such that lock loop correlator 1086 should optimally be sampling a zero crossing of received signal 1006 when first data correlator 1008 is synchronously sampling received signal 1006. However, lock loop correlator 1086 will not be sampling a zero crossing when second data correlator is actually sampling an impulse.
  • lock path switch 1048 assures that only the appropriate outputs from lock loop correlator 1086 are used in the lock loop to generate error signal 1052.
  • a second lock loop can be used, if desired, to determine errors based on the sampling by second data correlator 1026.
  • a second lock loop correlator (not shown) would sample received signal 1006 at a point in time that is offset (e.g., delayed) by % of an impulse width from the time that second data correlator 1026 samples received signal 1006.
  • the second lock loop correlator would be provided with a delayed template signal that was generated by delaying template signal 1074 by
  • error signal 1052 is provided to time base 1054.
  • time base 1054 can be implemented as part of precision timing generator 1060.
  • error signal 1052 can be provided directly to precision timing generator 1060.
  • time base 1054 is independent of precision timing generator 1050, error signal 1052 can be provided directly to precision timing generator 1060.
  • error signal 1052 is used to synchronize receiver 1002 with received impulse radio signal 1006 such that data correlators 1008 and 1026 sample received impulse radio signal 1006 at substantially optimal times for data detection.
  • a first position for an impulse waveform (e.g., impulse 902) can be used to represent a first data state (e.g., a binary "0")
  • a second position for an impulse waveform (e.g., e.g., impulse 904) can be used to represent a second data state (e.g., abinary " 1 ").
  • 100 impulses 902 i.e., an impulse train
  • 100 impulses 904 may be transmitted to represent a binary "1”.
  • each impulse of an impulse train (e.g., 100 impulses) may also be adjusted in time based on a code (e.g, code signal 1066).
  • a subcarrier that can be implemented adjusts modulation according to a predetermined pattern at a rate faster than the data rate. This same pattern is then used by a receiver to reverse the process and restore the original data pattern just before detection.
  • This method permits alternating current (AC) coupling of stages, or equivalent signal processing to eliminate direct current (DC) drift and errors from the detection process.
  • AC alternating current
  • DC direct current
  • the subcarrier signal used for subcarrier modulation is internally generated by precision timing generator 1008 (of transmitter 1002) and added to baseband signals (e.g., information signals which may or may not also be coded).
  • FIGS. 13 A and 13B Assume only two transmit states: state A (i.e., impulse 902) associated with data " 0" ; and state B (i.e., impulse 904) associated with data " 1" . Also assume that four impulses are to be transmitted for each data state. As shown in FIG. 13 A, without subcarrier modulation (and assuming no coding), a signal 1302A consisting of AAAA (i.e., four impulses 902) is transmitted to represent a data "0" . As shown in FIG. 13B, without subcarrier modulation (and without coding), a signal 1302B consisting of BBBB (i.e., four impulses 1004) is transmitted to represent a data "1".
  • state A i.e., impulse 902
  • state B i.e., impulse 904
  • FIGS. 13 A and 13B Assume only two transmit states: state A (i.e., impulse 902) associated with data " 0" ; and state B (i.e., impulse
  • An example of a subcarrier modulation scheme is to transmit a signal 1304 A consisting of AB AB to represent a data " 0 " (as shown in FIG. 13 A) and a signal 1304B consisting of BAB A to represent a data " 1 " (as shown in FIG. 13B).
  • Other possibilities include, but are not limited to, transmitting a signal consisting of AABB (not shown) to represent a data "0" and a signal consisting of BB AA (not shown) to represent a data " 1" .
  • an impulse radio receiver When subcarrier modulation is used, an impulse radio receiver must demodulate (i.e., remove) the subcarrier signal to yield an information signal.
  • An impulse radio receiver is typically a direct conversion receiver with a cross correlator front end in which the front end coherently converts an electromagnetic impulse train of monocycle pulses to a baseband signal in a single stage. The receiver uses the same pattern, that was used to produce the subcarrier modulation, to reverse the process and restore the original data pattern just before data detection.
  • subcarrier demodulator 1016 performs any necessary subcarrier demodulation. More specifically, subcarrier demodulator 1016 provides its outputs 1018 and 1038 to the correct accumulators 1020 and 1040 so that max value selector 1027 can correctly determine which data state was represented by a train of impulses. Accordingly, the exact structure and function of subcarrier demodulator 1016 is dependent on the subcarrier modulation pattern that is used by an impulse radio transmitter (e.g., by transmitter 602). Referring back to FIG. 10, in this embodiment, subcarrier demodulator 1016 outputs signals 1018 and 1038, which represent values that correspond to possible data states.
  • signal 1018 corresponds to a binary "0" and signal 1038 corresponds to a binary "1".
  • Signal 1018 is provided to a summing accumulator 1020, and signal 1038 is provided to a summing accumulator 1040.
  • max value selector 1027 compares an output 1024 of accumulator 1020 to an output 1044 of accumulator 1040 to determine, for example, if the data bit (associated with the received impulses) is a "0" or a "1".
  • accumulators 1020 and 1040 are only necessary if more than one impulse (e.g., 4, 8 or 100 impulses) are used to represent each data state (e.g., bit or bits).
  • accumulators 1020 and 1040 will each add 100 values (i.e., accumulator 1020 will sum signals 1018 and accumulator 1040 will sum signals 1038) and provide the summation values (signals 1024 and 1044, respectively) to max value selector 1027, and then add the next 100 values and provide the summation values to max value selector 1027, and so on.
  • each data state e.g., bit or bits
  • output signals 1018 and 1038 are provided directly (i.e., without the need for accumulators 1020 and 1040) to max value selector 1027.
  • Subcarrier demodulator 1016 provides its outputs 1018 and 1038 to the correct accumulators 1020 and 1040 so that max value selector 1027 can correctly determine which data state was represented by a train of impulses.
  • receiver 1002 including two distinct data correlators 1008 and 1026 and one distinct lock loop correlator 1086. It is noted that the functions of these correlators can be combined into one or two correlators.
  • FIG. 14 shows a receiver 1402 that includes a single correlator 1404 that samples received signal 1006 three times during each frame.
  • receiver 1402 is shown as including multiplexers 1412 and 1414. Multiplexer 1414 is used to provide the appropriate reference signal 1074, 1078 or 1082 to correlator 1404.
  • correlator 1404 samples received signal 1006 at a first precise time that is controlled by a phase lock loop, at a second precise time slightly delayed from the first time (e.g., by 0.125 nsec) and used in the phase lock loop, and at a third precise time delayed from the first time by the offset used in the modulation scheme (e.g., by 5.0 nsec).
  • Outputs 1406 of correlator 1404 are provided to A/D converter 1408 which converts outputs 1406 to digital values 1410.
  • Multiplexer 1412 separates values
  • receiver 1402 into three paths 1014, 1036 and 1092.
  • the remaining elements of receiver 1402 function the same as they do in receiver 1002, discussed above.
  • FIG. 16 shows an exemplary impulse radio receiver 1602 for receiving one-of-four positions modulated signals.
  • An example of a one-of-four-positions modulation scheme is described in connection with FIG. 15.
  • an impulse waveform 1502 (or a plurality of impulse waveforms 1502) is used to represent a first data state (e.g., bits "00")
  • an impulse waveform 1504 (or a plurality of impulse waveforms 1504) is used to represent a second data state ( e -g- 3 bits "01 ")
  • an impulse waveform 1506 or a plurality of impulse waveforms
  • an impulse waveform 1508 (or a plurality of impulse waveforms 1508) is used to represent a fourth data state (e.g., bits "11").
  • Impulses 1502, 1504, 1506 and 1508 are exemplary waveforms associated with transmitted signals (e.g., signals transmitted through the air from a transmitter to areceiver). Once impulses 1502, 1504, 1506 and 1508 are received by an antenna of a receiver, their waveforms typically resemble their first derivatives due to the receive antenna response, as discussed above. Thus the received impulses resemble a "w" (signals 1512, 1514, 1516 and 1518, respectively) and are referred to as “w-pulses” or "triplets”.
  • receiver 1602 is similar to receiver 1002, except receiver 1602 includes four data correlators 1608, 1609, 1626 and 1621, where a template signal 1680 provided to second data correlator 1609 is delayed by 5.0 nsec (i.e., from template signal 1674), atemplate signal 1678 provided to third data correlator 1626 is delayed by 10.0 nsec, and a template signal 1676 provide to the fourth correlator 1621 is delayed by 15.0 nsec.
  • a template signal 1680 provided to second data correlator 1609 is delayed by 5.0 nsec (i.e., from template signal 1674)
  • atemplate signal 1678 provided to third data correlator 1626 is delayed by 10.0 nsec
  • a template signal 1676 provide to the fourth correlator 1621 is delayed by 15.0 nsec.
  • first data correlator 1608 is used to sample impulses 1512
  • second data correlator 1604 is used to sample impulses 1514
  • third data correlator 1606 is used to sample impulses 1516
  • fourth data correlator 1608 is used to sample impulses 1518.
  • Receiver 1602 functions in a similar manner as receiver 1002 explained in detail above, except receiver 1602 is capable of detecting four different positions of received impulses.
  • data detector 1603 can detect at least four different data states (e.g., bits "00", "01", “10” or "1 1). Accordingly, data detector 1603 is shown as having two parallel outputs 1646 and 1648. Data detector 1603 can alternatively have a single serial output.
  • lock loop switch 1648 only provides an output 1696 (of an accumulator 1694) to a lock loop filter 1650 (as value 1649) when data outputs 1646 and 1647 (of a max value selector 1627) indicates that value
  • 1624 is greater than output values 1625 , 1644 and 1645.
  • a second, third and even forth lock loop can be used if desired, to determine errors based on the sampling by second data correlator 1609, third data correlator 1626 and fourth data correlator 1621.
  • the above described embodiment can be modified to support more than four different data states. For example, a one-of-five positions, a one-of-eight positions, or a one-of-N positions receiver can be implemented in a similar manner to that described above. Further, additional lock loops can be added as discussed above.
  • An example of subcarrier modulation, for use with a one-of-four positions modulation scheme, can be illustrated with reference to FIGS. 17 A and 17B.
  • state A impulse 1502/1512
  • state B impulse 1504/ 1514
  • state C impulse 1506/ 1516
  • state D impulse 1508/1518
  • data e.g., bits
  • a signal 1702 A consisting of AAAA i.e., four impulses 1502
  • a signal 1702B consisting of BBBB i.e., four impulses 1504
  • BBBB i.e., four impulses 1504
  • a signal (not shown) consisting of CCCC i.e., four impulses 1506 is transmitted to represent data "10" and a signal (not shown) consisting of DDDD (i.e., four impulses 1508) is transmitted to represent data "11".
  • An example of a subcarrier modulation scheme is to transmit a signal 1704A consisting of ABCD to represent data state "00" (as shown in FIG. 17A), transmit a signal 1704B consisting of BCD A to represent data state "01” (as shown in FIG. 17B), transmit a signal consisting of CDAB to represent data state "10” (not shown), and transmit a signal consisting of DABC to represent data state "11” (not shown).
  • an impulse radio receiver is typically a direct conversion receiver with a cross correlator front end in which the front end coherently converts an electromagnetic impulse train of monocycle pulses to a baseband signal in a single stage. This same pattern is then used to reverse the process and restore the original data pattern just before detection.
  • subcarrier demodulator 1616 of impulse radio receiver 1602 performs any necessary subcarrier demodulation. More specifically, subcarrier demodulator 161 provides its outputs 1618, 1619, 1638 and 1639 to the correct accumulators 1620, 1621, 1640 and 1641 so that max value selector 1627 can correctly determine which data state was represented by a train of impulses. Accordingly, the exact structure and function of subcarrier demodulator 1616 is dependent on the subcarrier modulation pattern that is used by an impulse radio transmitter (e.g., by transmitter 602).
  • A/D converters 1612, 1613, 1634 and 1035, subcarrier demodulator 1616, summing accumulators 1620, 1621, 1640 and 1644 and max value selector 1627 can be thought of as being components of a data detector 1603 (shown by dotted lines).
  • data detector 1603 produces parallel data output signals 1646 and 1647 based on outputs 1610, 1611, 1632 and 1633 of first, second, third, and fourth correlators 1608, 1609, 1626 and 1621.
  • data detector 1603 can output a single serial data output signal.
  • FIG. 18 shows another exemplary impulse radio receiver 1802 for receiving one-of-many positions modulated signals.
  • Receiver 1802 includes an antenna 1804 for receiving a propagated impulse radio signal.
  • Received signal
  • data correlator 1808 also called sampler 1808
  • data correlator 1808 produces a baseband output signal 1810 (also referred to as a correlator output signal 1810, or correlator output 1810).
  • Correlator 1808 ideally comprises a multiplier followed by a short term integrator to sum the multiplied product over the pulse interval.
  • Received signal 1806 is also input to lock loop correlator 1886 that is used in a lock loop that corrects drifts in a receiver time base 1854. It is important to correct drifts in time base 1854 so that data correlator 1808 samples received signal 1806 as the appropriate times.
  • the lock loop function is described in additional detail below.
  • Receiver 1802 also includes a precision timing generator 1860, which receives a periodic timing signal 1856 from receiver time base 1854.
  • Time base receives a periodic timing signal 1856 from receiver time base 1854.
  • Precision timing generator 1860 provides a synchronization signal 1866 to an optional code generator 1864 and receives a code control signal 1862 (also referred to as coding signal 1862) from optional code generator 1864. Precision timing generator 1860 utilizes periodic timing signal 1856 and optional code control signal 1862 to produce a (coded) timing signal 1870. Template generator 1872 (also referred to as a pulse generator 1872) is triggered by (coded) timing signal 1870 and produces a train of template signal pulses 1874 (also referred to as reference signal pulses 1874).
  • time base 1854, precision timing generator 1860, template generator 1872 and delay 1876 can be combined into a single sampling/timing generator that provides the appropriate reference signals to data correlator 1808 and lock loop correlator 1886 at the precise times to synchronously sample received signal 1806.
  • precision timing generator 1860, template generator 1872 and delay 1876 can be combined into a single sampling/timing generator that provides the appropriate reference signals to data correlator 1808 and lock loop correlator 1886 at the precise times to synchronously sample received signal 1806.
  • delay 1876 can be combined into a single sampling/timing generator that provides the appropriate reference signals to data correlator 1808 and lock loop correlator 1886 at the precise times to synchronously sample received signal 1806.
  • these elements are shown as being distinct elements to better explain the present invention.
  • code generator 1864 If code generator 1864 is used, then the code for receiving a given signal is the same code utilized by the originating transmitter (e.g., used by code generator 612 of transmitter 602) to generate the propagated signal.
  • the timing of template pulse train 1874 also referred to as template signal 1874 matches the timing of received signal impulse train 1806, allowing received signal 1806 to be synchronously sampled by correlator 1808.
  • Template signal 1874 is used by data correlator 1808 to sample received signal 1806, as discussed above.
  • the number of impulses per frame (e.g., 100 nsec) in template signal 1874 is dependent upon the modulation scheme used. For example, if one-of-two positions modulation is used, then template signal 1874 consists of two reference pulses per frame. If one-of-four positions modulation is used, then template signal 1874 consists of four reference pulses per frame. More generally, if a one-of-N positions modulation is used, then template signal 1874 consists of N reference pulses per frame.
  • each template reference pulse within a frame is dependent upon the possible positions where impulses (of received signal 1806) may be located. For example, if one-of-four positions modulation is used, as shown in FIG. 15, then template signal 1874 consists of four template reference pulses that are spaced 5.0 nsec apart. The use of coding can place these reference pulses at various positions within each frame, depending on the coding scheme. What is important is that template signal 1874 includes the same number of reference pulses as there are modulation states, and that the position of each reference pulse is dependent on the possible positions of an impulse in received signal 1806. Put in other words, precision timing generator 1860 triggers the sampling of received signal 1806 at each possible position of an impulse within a frame of received signal 1806.
  • Template signal 1874 is also provided to counter 1828.
  • Counter 1828 is incremented by one each time it receives a reference pulse of template signal 1874.
  • Counter 1828 is designed such that it resets after the total number of possible modulations states are counted. For example, if a one-of-four positions modulation scheme is used, then counter 1828 counts up to four, and then resets. Thus, counter 1828 is reset once every frame.
  • a count output 1830 is provided to a latch 1816, which is triggered by a tlireshold output 1814 of a threshold compare 1812, as discussed in more detail below.
  • Data correlator 1808 correlates received signal 1806 with template signal 1874 and outputs a correlator output 1810. In other words, data correlator 1808 samples received signal 1806 based on precision timing generator 1860.
  • template signal 1874 includes a number of reference pulses per frame that is equal to the number of modulation states that was used by the transmitter. For example, if a one-of-four positions modulation scheme is used, then template signal 1874 includes four reference pulses per frame, causing data correlator 1808 to sample received signal 1806 four times per frame. Additionally, as discussed above, the location of each reference pulse of template signal 1874 is dependent on the possible locations where impulses (or received signal 1806) may be located.
  • output 1810 of data correlator 1808 should be zero for all points in time except where the actual impulse is located within a frame of received signal 1806. However, this is not typically the case because received signal 1806 includes multipath reflections and noise.
  • Data correlator output 1810 is provided to a tlireshold comparator 1812 and to a data sample and hold (S/H) 1818.
  • Threshold compare 1812 which compares data correlator output 1810 to a threshold voltage value, provides a trigger signal 1814 to both latch 1816 and data S/H 1818 when threshold compare 1812 receives a data correlator output 1810 that exceeds the threshold value.
  • Data S/H 1818 samples the value of data correlator output 1810 so that if more than one threshold crossing is detected within a frame, the magnitudes of the threshold crossings can be compared (this is explained in more detail below).
  • Latch 1816 stores the value of counter 1828 (in this example, counter value 1828 is one, two, three or four). If the counter is a binary counter, then the values stored in counter 1828 are, for example, "00", "01", "10" or "11".
  • the threshold value used by threshold compare 1812 can be a predetermined value.
  • the threshold value used by threshold compare 1812 can be determined by controller 1830 based on output 1824 of A/D converter 1820, and thus vary over time.
  • the threshold value determined by controller 1830 is slightly greater than one half
  • Output 1824 of data S/H 1818 is provided to A/D converter 1820, which converts the stored value of data correlator output 1810 to a digital value 1824, which is provided to data detector 1803 and to optional controller 1830.
  • An output 1822 of latch 1816 is also provided to data detector 1803.
  • data detector 1803 can match each digital output 1824 of A/D 1820 with in impulse position (based on output 1822 of latch 1816).
  • FIGS. 19 and 20 illustrate example embodiments of data detector 1803.
  • data detector 1803 receives output 1824 of A/D converter 1820 and output 1822 of latch 1816.
  • output 1822 of latch 1816 which is a count value that corresponds to when the threshold is exceeded, is used to select (e.g., using a switch 1902) which summing accumulator 1920, 1921, 1940 and 1941 receives output 1824 of A/D converter 1820.
  • a max value selector 1927 compares an output 1924 of accumulator 1920, an output 1925 of accumulator 1921, an output 1944 of accumulator 1940 and an output 1945 of accumulator 1941 to determine, for example, if the data bits (associated with a plurality of received impulses) are "00", “01 ", " 10" or "11". This determination is also referred to as a demodulation decision.
  • accumulators 1920, 1921, 1940 and 1941 are only necessary if more than one impulse (e.g., 4, 8 or 100 impulses) are used to represent each symbol (also referred to as data state (e.g., bits)).
  • accumulators 1920, 1921 , 1940 and 1941 will each provide a summation value (signals 1924, 1925, 1944 and 1945) to max value selector once every 100 frames.
  • each data state e.g., bits
  • the outputs of switch 1902 are provided directly (i.e., without the need for accumulators 1920, 1921, 1940 and 1941) to max value selector 1927.
  • data detector 1803 is used for demodulating one- of-four positions modulated signals. Accordingly, data detector 1803 is shown as having two parallel data outputs 1846 and 1847. The number of parallel outputs is dependent on the modulation scheme used.
  • data detector 1603 should have three parallel data outputs (i.e., because three bits are required to represent eight different states) .
  • Data detector 1803 can alternatively have a single serial output.
  • FIG. 19 if more than one threshold crossing is detected during one frame, then more than one of accumulators 1920, 1921, 1940, 1941 will receive a value 1824 from A/D converter 1820.
  • a per frame max value detector 2002 determines which ofthe values (causing the more than one threshold crossings) is greatest in magnitude.
  • the per frame max value detector 2002 will provide the value 2024 that is greatest in magnitude to the accumulator 1920, 1921 , 1940 or 1941 based on the count value 1822 (provided by latch 1816) that corresponds to that value (i.e., of greatest magnitude), using a selector signal 2022.
  • This embodiment should have a greater signal to noise ratio than the embodiment of FIG. 19, because the probability is reduce of providing values consisting purely of noise and delayed multipath reflections to one ofthe accumulators 1920, 1922, 1040 or 1941.
  • a lock loop (also referred to as a control loop) is used to generate an error signal 1852 that corrects drifts in time base 1854. More specifically, a control loop including lock loop filter 1850, time base 1854, precision timing generator 1860, template generator 1872, delay 1876, lock loop correlator 1886, a lock S/H 1890, an A/D converter 1894 and a lock path switch 1848, is used to generate error signal 1852. Error signal 1852 provides adjustments to the adjustable time base 1854 to time position periodic timing signal 1856 in relation to the position of received signal 1806. The function ofthe lock loop is described in more detail, below.
  • delay 1876 is precisely selected such that lock loop correlator 1886 samples an impulse of received signal 1806 at a zero crossing when received signal 1806 and non-delayed template signal 1874 are synchronized.
  • delay 1876 delays template signal 1874 by a quarter of an impulse width.
  • the width of each received impulse is 0.5 nsec, then delay 1876 delays template signal 1874 by 0.125 nsec
  • Data correlator 1808 which receives template signal 1874, samples received signal 1806 at each position where an impulse may be located within a frame.
  • lock loop correlator 1886 which receives delayed template signal 1874, samples received signal at precise positions where each impulse may be crossing zero. For example, if receiver 1802 receives signals that are modulated according to the one-of-four positions modulation scheme discussed above, then data correlator 1808 samples received signal 1806 four times per frame (preferably near the center of each possible impulse position), and lock loop correlator 1886 also samples received signal 1806 four time per frame.
  • Lock loop S/H 1890 samples the value of lock loop correlator output 1888 when it is triggered by signal 1814.
  • An output 1892 of lock loop S/H 1890 is converted to a digital value 1896 by A/D converter 1896.
  • Digital value 1896 is provided to lock loop switch 1848, which also receives output 1822 of latch 1816.
  • lock loop switch 1848 can match each digital value 1896 with an impulse position (i.e., based on output 1822 of latch 1816).
  • Lock loop switch 1848 also receives data output 1846 (and possibly additional data outputs, such as 1847, depending on the number of data states and depending on whether parallel outputs are used or a serial output is used).
  • lock loop switch 1848 can determine which ofthe values (causing the more than one tlireshold crossings) is greatest in magnitude, and then use the corresponding digital value 1896 in the lock loop. In other words, if lock loop switch 1848 receives more than one digital value 1896 during a single frame, lock loop switch 1848 determines which digital value 1896 to provide to lock loop filter 1850 via a path 1849.
  • FIGS. 21 and 22 can be used to further explain the above discussed embodiment of receiver 1802.
  • the first possible position of an impulse begins 5.0 nsec into a 100 nsec frame (where the impulse width if 0.5 nsec); the second possible position of an impulse begin 10.0 nsec into the 100 nsec frame; the third possible position of an impulse begins 15.0 nsec into the 100 nsec frame; and the fourth possible position of an impulse begins 20.0 nsec into the 100 nsec frame.
  • FIG. 21 also shows an example template signal 1874 that is used by data correlator 1808 to sample received signal 1806.
  • template pulses 2112, 2114, 2116 and 2118 are preferably centered about the center of each possible impulse position.
  • Exemplary reference pulses 2112, 2114, 2116 and 2118 are shown as being less than a half the width ofthe possible received impulses. More specifically, pulses 2112, 2114, 2116 and 2118 are shown as being 0.15 nsec wide, where the received impulses are approximately 0.5 nsec wide.
  • FIG. 22 shows an example of correlator output 1810 over a frame interval (e.g., 100 nsec). Notice, in this example, correlator output 1810 exceeds a tlireshold value (designated by dotted line 2206) at a first point in time 2202 and a second point in time 2204. As discussed above, in theory, output 1810 of data correlator 1808 should be zero for all points in time except for where the actual impulse is located within a frame of received signal 1806. However, this is not the case, as shown in FIG.22, because received signal 1806 includes noise and/or delayed multipath reflections.
  • a frame interval e.g. 100 nsec
  • modulation is accomplished by placing impulses at distinct positions within a frame.
  • a one-of-four positions modulation scheme four distinct positions separated by 5 nsec, exists within each frame (e.g., a 100 nsec frame).
  • modulation can be accomplished by placing an impulse at one ofthe four positions.
  • an impulse in the first position can represent bits "00”
  • an impulse in the second position can represent bits "01”
  • an impulse in the third position can represent bits "10”
  • an impulse in the fourth bin can represent bits "11 ".
  • Such an example modulation scheme is discussed above with connection to FIG. 15. Referring to the received signals 1512, 1514, 1516 and 1518 of FIG.
  • impulse radios are typically resistant to the effects of multipath effects because delayed multipath reflections typically arrive outside the correlation time and thus have generally been ignored. However, this is not necessarily the case when receiving impulses that have been modulated using a one-of-many positions modulation scheme. Rather, in a one-of-many positions modulation scheme, it is very probable that a delayed multipath reflection associated with an impulse placed in a first location will arrive during the correlation times (also referred to as sampling times) of downstream correlations (also referred to as downstream samples).
  • ringing also referred to as downstream artifacts
  • ringing is defined as those signal attributes associated with an impulse that are located later in time than (i.e., downstream from) the intended (or expected) waveform of a received impulse.
  • those signal attributes located later in time than 2302 are downstream artifacts.
  • ringing can be caused by a number of other things.
  • ringing can also be caused by components within an impulse radio transmitter and/or by components within an impulse radio receiver.
  • This ringing can cause demodulation decision errors if the ringing plus noise is greater than the signal (i.e., impulse) plus noise.
  • a receiver used in a one-of-four positions modulation scheme samples a received signal at least four times per frame in an attempt to determine which data state was received. If the sample value (i.e., correlation output) associated with a downsteam artifact plus noise (e.g., taken at the second position of the four positions) is greater than the sample value ofthe actual impulse plus noise (e.g., taken at the first position), then the receiver can make a wrong demodulation decision regarding which data state (also referred to as, symbol) is associated with the frame ofthe receive signal.
  • the sample value i.e., correlation output
  • the actual impulse plus noise e.g., taken at the first position
  • a feature ofthe present invention is the use these downstream artifacts to increase the confidence of demodulation decisions. Another feature of the present invention is to adjust the downstream positions (e.,g., the second, third and fourth positions) used during transmission of impulses and to adjust the downstream sampling positions during reception of impulses, so that the disruptive effects of downstream artifacts are reduced. A further feature ofthe present invention is to combine the above features such that downstream positions are adjusted to maximize the confidence of a demodulation decision that includes consideration of downstream artifact measurements. These aspects ofthe invention can be illustrated using FIGS. 23 A - 23D.
  • FIG. 23 A when an impulse 2302 is in the first position it can cause ringing in the following three positions. As shown in FIG.23B, when an impulse 2304 is in the second position, it can cause ringing in the third and fourth positions. As shown in FIG. 23 C, when an impulse 2306 is in the third position, it causes ringing in the fourth position. When an impulse 2308 is in the fourth position, it causes no ringing in any of the other three positions.
  • a receiver is trained so that the artifacts received at downstream positions can be used to assist in malting demodulation decisions. More specifically, assuming a one-of-four positions modulation scheme, a training sequence is sent from a transmitter to the receiver.
  • the training sequence consists of a plurality of frames (e.g., 100 frames) with an impulse in the first position of each frame, followed by a plurality of frames with an impulse in the second position of each frame and then followed by a plurality of frames with an impulse in the third position of each frame.
  • This training sequence can occur periodically (e.g., between each packet, or more likely between each of a plurality of packets) so that the receiver's knowledge of downstream artifacts can still be useful even if the receiver and/or transmitter are moving with respect to one another, if the noise pattern is varying and/or if the surfaces causing multipath reflections are moving.
  • the plurality of impulses in the second position are received and the second correlator locks onto the impulses in the second position, the third correlator samples the ringing at the third position, and the fourth correlator samples the ringing at the fourth position.
  • This information is also stored in the artifact table.
  • the plurality of impulses in the third position are received, the third correlator locks onto the impulses in the third position, and the fourth correlator samples the ringing the fourth position. This information is also stored in the artifact table.
  • values corresponding to the samples by the first correlator when the impulse is in the first position can also be stored in the artifact table and used during demodulation decisions. This is discussed below in connection with FIG. 24.
  • the receiver can also predict what the third correlator will see when the impulse is in the third position. Thus, by measuring the downstream artifacts, the confidence in decisions can be increased.
  • An example of an artifact table 2402 for use in a receiver that receives one-of-four positions modulated signals is shown in FIG. 24. In table 2402, "A" corresponds to the first position, "B” corresponds to the second position, "C” corresponds to the third position and "D" corresponds to the fourth position.
  • a value A A associated with the first correlator is stored in column 2412
  • a downstream artifact value B A associated with the second correlator is stored in column 2414
  • a downstream artifact value C A associated with the third correlator is stored in column 2416
  • a downstream artifact value D A associated with the fourth correlator is stored in column 2418.
  • the full scale letters i.e., A, B, C and D
  • the subscript letters i.e., A , B and c
  • the value A A associated with the first correlator can be the output 1624 of first accumulator 1620
  • the value B A associated with the second correlator can be the output 1625 of second accumulator 1621
  • the value C A associated with the third correlator can be output 1644 of third accumulator 1640
  • the value D A associated with the fourth correlator can be output 1645 of fourth accumulator 1641. Since the values stored in row 2404 are associated with frames where the impulse is located in the first position, values B A , C A and D A are referred to as downstream artifact values.
  • a value B B associated with the second correlator is stored in column 2414
  • a downstream artifact value C B associated with the third correlator is stored in column 2416
  • a downstream artifact value D B associated with the fourth correlator is stored in column 2418. Since the values stored in row 2406 are associated with frames where the impulse is located in the second position, values C B and D B are referred to as downstream artifact values.
  • a value C c associated with the third correlator is stored in column 2416 and a downstream artifact value D c associated with the fourth correlator is stored in column 2418. Since the values stored in row 2408 are associated with frames where the impulse is located in the third position, value D B is referred to as a downstream artifact value.
  • downstream artifact values e.g., B A , C A , D A , C B , D B and D c
  • downstream artifact values e.g., B A , C A , D A , C B , D B and D c
  • C c can also be used during demodulation decisions. This is especially useful where a downstream artifact value exceeds the value associated with the output ofthe correlator actually sampling an impulse (e.g., if B A >A A )
  • downstream positions e.,g., the second, third and fourth positions
  • downstream sampling positions used during reception of impulses are correspondingly adjusted, so that the disruptive effects of downstream artifacts are reduced.
  • scanning correlators are used to fill an artifact table, which has more entries than the artifact table 2402 discussed in connection with FIG. 24. Additional details of scanning correlators are disclosed in commonly owned U.S. Patent Application No. 09/537,264, filed March 29, 2000, entitled “System and Method Utilizing Multiple Correlator Receivers in an Impulse Radio System,” which is incorporated herein by reference in its entirety.
  • a plurality of frames having the impulse in the first position are sent to the receiver.
  • the receiver receives an impulse radio signal and a first correlator ofthe receiver locks onto the impulses in the first position. While this first correlator remains locked onto the impulses in the first position, a one or more scanning correlators are used to sample multiple points (e.g., with each of the points separated by approximately 1/4 of the width of each impulse) surrounding the remaining positions (i.e., the second, third and fourth positions) in order to populate an artifact table.
  • the receiver can determine points near the second, third, and fourth positions where the ringing causes a max positive peak (e.g., point 2310), a null (e.g., point 2314) and a max negative peak (e.g., point 2312).
  • Some or all ofthe information in the artifact table can then be provided (i.e., transmitted) to the transmitter so that the transmitter can adjust the locations of the second, third, and fourth positions, to thereby increase the probability of making correct decisions.
  • the transmitter adjusts the second, third and fourth positions such that they are located at nulls ofthe downstream artifacts.
  • the transmitter When the transmitter changes the locations of these positions, the transmitter must inform the receiver of the changed locations so that the correlators of the receiver sample received signals at the appropriate points in time.
  • the receiver determines how the transmitter should adjust the positions of impulses (based on the artifact table) and transmits information relating to the new positions back to the transmitter.
  • the transmitter transmits impulses at the nulls near the second third and fourth positions. This can increase the confidence that a data detection decision is correct because ringing should not as significantly corrupt the downstream samples ofthe received signal made by the second, third, and fourth correlators. However, if the downstream artifact values are also used to make a decision (as described above, under the heading "Use of Artifacts to Increase Confidence of a Decision"), then it may not be optimal to transmit impulses at such nulls.
  • the downstream locations are shifted with respect to the first location.
  • all ofthe locations can be changing on a frame by frame basis due to coding, which is discussed above.
  • the shifting that is referring to in this embodiment is shifting in addition to any moving of impulse positions due to coding.
  • FIGS. 25 A and 25B This embodiment ofthe present invention can be further explained with reference to FIGS. 25 A and 25B.
  • a first frame e.g., a 100 nsec frame
  • the second, third and fourth ofthe four possible positions of an impulse are each shifted by 1 nsec as compared to their original positions and with respect to the first position.
  • the shift is greater than the width of each impulse (for this example, greater than 0.5 nsec).
  • the second, third and fourth possible positions of an impulse can be the same as in FIG. 25 A.
  • each ofthe second, third and fourth positions can be shifted to yet another location.
  • the receiver is adjusting the locations within a frame where its second, third, and fourth correlators are sampling frames of a received signal. Referring to FIG. 16, this can be accomplished, for example, by appropriately adjusting delays 1679, 1677 and 1675.
  • an impulse is placed within one of a plurality of possible positions within each time frame of an impulse radio signal. For example, if two possible positions exist within a time frame, then each position can represent one of two data states (e.g., a 0 bit, or a 1 bit). If four possible positions exist within a time frame, then four data states can be represented (e.g., each position can represent two bits, i.e., 00, 01,
  • each position can represent one of eight data states (e.g., bits 000, 001, 010, 011, 100, 101, 110 , or 111), and so on. Collectively, these embodiments have been referred to as "one-of-many" positions modulation or "one-of-N" positions modulation.
  • impulses can be placed in more than one position within each time frame.
  • impulses in a "two- of-four" positions modulation scheme, impulses can be placed in the first and second positions, in the first and third positions, in the first and fourth positions, in the second and third positions, in the second and fourth positions, or the third and fourth positions.
  • five different data states can be represented. This is one additional data state than in the "one-of-four" positions modulations scheme discussed above.
  • 28 different data states can be represented. This is 20 additional data states than in the "one-of-eight" positions modulation scheme discussed above.
  • an "M-of-N" positions modulation scheme also referred to as an "M-of-many” positions modulation scheme can be used to significantly increase the data throughput of an impulse radio communications system.
  • each impulse in addition to placing each impulse at one-of-N widely separated positions within each time frame, each impulse can also be dithered by less than Vz the width of each impulse, thereby doubling the number of data states.
  • Vz the width of each impulse
  • each impulse in addition to placing each impulse in one-of-N positions within each frame, each impulse can also be flipped (i.e., inverted), thereby doubling the number of data states.
  • a non-inverted impulse in a one-of-four positions with shift modulation scheme, can be located in one of four possible positions or an inverted impulse can be located in one of the four possible postions, providing for eight data states.
  • Flip modulation was described in U.S. Patent Application No. 09/537,629, filed
  • the amplitude of each impulse in addition to placing each impulse in one-of-N positions within each frame, can also be varied to create additional data states. For example, if each impulse can have one of two different amplitudes in a one-of-four positions modulation scheme, then eight data states exist. If each impulse can have one of three different amplitudes in a one-of-four positions modulation scheme, then twelve data states exist.
  • the various embodiment of the present invention can be combined to further increase the number of different data states that can be represented in a frame, and thus to increase the data throughput in an impulse radio communications system.
  • M-of-N positions modulation can be combined with flip and/or amplitude modulation.
  • one-of-N positions with shift modulation can be combined with flip modulation.
  • the present invention relates to the transmission and reception of signals that are modulated using what has been referred to as "one-of-many positions" modulation.
  • what has been referred to as “one-of-four-positions” modulation is used.
  • a first data state corresponds to an impulse located at a first position within a time frame
  • a second data state corresponds to an impulse located at a second position within the time frame
  • a third data state corresponds to an impulse located at a third position within the time frame
  • a fourth data state corresponds to an impulse located at a fourth position within the time frame.
  • the teachings of the present invention can be used to develop modulation schemes that include even more data states, while still being within the spirit and scope ofthe present invention.
  • the teachings ofthe present invention can be used to create modulations schemes with six, eight, or more different data states. Accordingly, the intention is for the present invention to encompass such additional modulation schemes and the apparatus, methods, and systems associated with them.
  • the present invention also includes the combination of one-of-many positions modulation with other modulation techniques, such as, flip and amplitude modulation.
  • control processor which in effect comprises a computer system.
  • a computer system includes, for example, one or more processors that are connected to a communication bus.
  • processors that are connected to a communication bus.
  • telecommunication-specific hardware can be used to implement the present invention, the following description of a general purpose type computer system is provided for completeness.
  • the computer system can also include a main memory, preferably a random access memory (RAM), and can also include a secondary memory.
  • the secondary memory can include, for example, a hard disk drive and/or a removable storage drive.
  • the removable storage drive reads from and/or writes to a removable storage unit in a well known manner.
  • the removable storage unit represents a floppy disk, magnetic tape, optical disk, and the like, which is read by and written to by the removable storage drive.
  • the removable storage unit includes a computer usable storage medium having stored therein computer software and/or data.
  • the secondary memory can include other similar means for allowing computer programs or other instructions to be loaded into the computer system. Such means can include, for example, a removable storage unit and an interface.
  • Examples of such can include a program cartridge and cartridge interface (such as that found in video game devices), a removable memory chip (such as an EPROM, or PROM) and associated socket, and other removable storage units and interfaces which allow software and data to be transferred from the removable storage unit to the computer system.
  • a program cartridge and cartridge interface such as that found in video game devices
  • a removable memory chip such as an EPROM, or PROM
  • PROM PROM
  • other removable storage units and interfaces which allow software and data to be transferred from the removable storage unit to the computer system.
  • the computer system can also include a communications interface.
  • the communications interface allows software and data to be transferred between the computer system and external devices. Examples of communications interfaces include, but are not limited to a modem, a network interface (such as an Ethernet card), a communications port, a PCMCIA slot and card, etc.
  • Software and data transferred via the communications interface are in the form of signals that can be electronic, electromagnetic, optical or other signals capable of being received by the communications interface. These signals are provided to the communications interface via a channel that can be implemented using wire or cable, fiber optics, a phone line, a cellular phone link, an RF link, and the like.
  • computer program medium and “computer usable medium” are used to generally refer to media such as removable storage device, a removable memory chip (such as an EPROM, or PROM) within a transceiver, and signals.
  • Computer program products are means for providing software to the computer system.
  • the software can be stored in a computer program product and loaded into the computer system using the removable storage drive, the memory chips or the communications interface.
  • the control logic when executed by a control processor, causes the control processor to perform certain functions ofthe invention as described herein.
  • features of the invention are implemented primarily in hardware using, for example, hardware components such as application specific integrated circuits (ASICs).
  • ASICs application specific integrated circuits
  • features ofthe invention can be implemented using a combination of both hardware and software.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
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  • Spectroscopy & Molecular Physics (AREA)
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  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)

Abstract

L'invention concerne un appareil, des systèmes et procédés d'émission et de réception de signaux radio à impulsions modulés d'une parmi plusieurs positions. Un récepteur radio à impulsions, destiné à démoduler un signal radio à impulsions reçu modulé selon une logique de modulation d'une de N positions, N étant le nombre de positions différentes possibles où une impulsion peut être située à l'intérieur de chaque cadre temporel du signal radio à impulsions, comprend un générateur de synchronisation, au moins un échantillonneur et un détecteur de données. Le générateur de synchronisation produit N signaux de synchronisation, chacun de ces signaux étant temporellement séparé par au moins ½ de la largeur des impulsions du signal radio à impulsions radio reçu. Les échantillonneurs sont déclenchés pour échantillonner le signal radio à impulsions reçu selon N signaux de synchronisation et pour produire une première à une Nième sorties d'échantillonneur. Le détecteur de données produit des décisions de démodulation fondées sur la première à la Nième sorties de l'échantillonneur.
PCT/US2001/018332 2000-06-07 2001-06-07 Appareil, systeme et procede de modulation a une position parmi plusieurs dans un systeme de communications radio a impulsions WO2001095508A2 (fr)

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2006073371A1 (fr) * 2005-01-03 2006-07-13 Matsushita Electric Industrial Co., Ltd. Procede et appareil de mappage d'impulsions dans la modulation radio a impulsions ('impulse radio' ou ir)
US10704386B2 (en) 2015-01-12 2020-07-07 Halliburton Energy Services, Inc. Wave reflection suppression in pulse modulation telemetry

Family Cites Families (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5677927A (en) * 1994-09-20 1997-10-14 Pulson Communications Corporation Ultrawide-band communication system and method
JP3307527B2 (ja) * 1995-09-29 2002-07-24 シャープ株式会社 Ｐｐｍ復調装置

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2006073371A1 (fr) * 2005-01-03 2006-07-13 Matsushita Electric Industrial Co., Ltd. Procede et appareil de mappage d'impulsions dans la modulation radio a impulsions ('impulse radio' ou ir)
US10704386B2 (en) 2015-01-12 2020-07-07 Halliburton Energy Services, Inc. Wave reflection suppression in pulse modulation telemetry

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AU2001265381A1 (en) 2001-12-17

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